Multiple Bosonic Mode Coupling in Electron Self-Energy of (La_2-xSr_x)CuO_4

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示波器(CRT)的操作和应用说明书

示波器(CRT)的操作和应用说明书

The Oscilloscope: Operation andApplications1.The OscilloscopeOscilloscope Operation (X vs Y mode)An oscilloscope can be used to measure voltage. It does this by measuring the voltage dropacross a resistor and in the process draws a small current. The voltage drop is amplified andused to deflect an electron beam in either the X (horizontal) or Y (vertical) axis using anelectric field. The electron beam creates a bright dot on the face of the Cathode Ray Tube(CRT) where it hits the phosphorous. The deflection, due to an applied voltage, can bemeasured with the aid of the calibrated lines on the graticule.First we will consider the circuitry that amplifies and conditions the voltage to be measured (the “Amp” block in figure 1).Figure 1. X vs. Y Deflection Block Diagram of the CRT The deflection of the oscilloscope beam is proportional to the input voltage (after ac or dccoupling). The amount of deflection (Volts/Division) depends upon the setting of theAMPL/DIV control for that channel (see figure 2).The input signal can be ac or dc coupled. Ac coupling involves adding a series capacitor. This has the effect of blocking (removing) the dc bias and low frequency components of a signal.Dc coupling does not have this problem and therefore allows you to measure voltages rightdown to 0 Hz. Ac coupling is useful when you are trying to measure a small ac voltage that is “on-top” of a large dc voltage. A typical example is trying to measure the noise of a dc power supply.Figure 2. Amplifier Block DiagramAmplifier FeaturesAMPL/DIV - This abbreviated name varies but it is generally some short form of amplitude per division. The control is a simple voltage divider (attenuator) which is used to change the sensitivity of the oscilloscope. At a 1 volt/DIV setting, a deflection of one major division on the graticule represents a one volt change at the oscilloscope input.Calibrated voltage measurementsThe small knob within the AMPL/DIV control must be rotated clockwise into its detente position for the amplifiers to be calibrated. Otherwise the voltage/division will be some unknown value greater than what the dial indicates.INV - There is almost always a control which lets you invert one channel. This can be used along with the ADD function to subtract two voltages. This is necessary because the common input (black lead of the oscilloscope cable) can only be connected to a 0V node. If channel A has V1 + V2 and channel B has voltage V1 then the reading of channel A + (-channel B) = (V1 + V2) + (-V1) = V2Position - For each axis there is a control which lets you shift the electron beam. With this you can set the zero voltage point to anywhere that is convenient for you.Oscilloscope InputsThe input of the oscilloscope can usually be modelled as a resistance and a parallel capacitance (see figure 3). The resistance is usually 1M S but it and the capacitance can vary greatly. The total or effective capacitance includes the oscilloscope circuitry (approx. 30 pF), cables (approx. 30 pF/m) and stray capacitance. The resistance will draw current from the circuit while the capacitance will add an RC time constant with its associated time delay, frequency response and distortion of some waveforms.The common connection (black lead or shield) at the input of the oscilloscope goes to the metal case as the symbol by the input connector shows. Because of this, the common input can only be connected to a 0V point in the circuit. Since the common inputs for both the A and B channels are connected to the case, they are effectively shorted together.Figure 3. DC-coupled Oscilloscope Input Circuitand Frequency ResponseFigure 4. AC-coupled Oscilloscope Input Circuit andFrequency ResponseFrequency response is calculated or measured by applying a pure sinusoidal waveform to a circuit. The circuit response is the output voltage divided by the input voltage. This is a complex number that can also be expressed as a magnitude (gain) and phase.Due to limitations in the amplifiers, the oscilloscope's frequency response is limited. Themanufacturer simply lists the half-power point for the oscilloscope without any external effects.Half power is also called the -3dB point. At this point, the voltage has decreased to 70.7% of its maximum. This means that only one-half of the maximum power would be dissipated in aresistive load. Keep in mind that an oscilloscope that is rated at 20 MHz is usually only accurate to 4 MHz for non-sinusoidal waveforms before distortion becomes a problem.With ac coupling (figure 4), an oscilloscope has another series RC circuit. It acts like a high pass filter (HPF). If you are viewing low frequency signals when ac coupled, not only will you not be able to measure any dc offset, but you will also be removing some low frequency information.Oscilloscope Operation (Voltage vs Time)The main function of an oscilloscope is to show voltage vs time. This is done by applying a ramp (or sawtooth) waveform into the X-axis amplifier as shown in figure 5. During the rising edge of the ramp, the electron beam scans across the screen. When the voltage drops back to 0V, the beam is turned off and quickly goes back to its starting point. This is signified by a thick line when the beam is on and a thin one when it is off (blanked).To obtain a stable picture on the CRT screen, the ramp waveform has to be in phase with the signal that you want to observe. This is done with a triggering circuit. The triggering circuit allows the oscilloscope to draw repeatedly the same waveform over and over by identifying the same point on a repetitive waveform.Figure 5.A Ramp-driven X-axis inputFigure 6. A Triggering ExampleFigure 7. Several Triggering CyclesThe triggering circuit allows you to select a voltage (an analog value) and an edge or slope (positive or negative) for the triggering circuit to compare to the input waveform. When the two are equal, the circuit puts out a pulse. This pulse triggers the ramp waveform generator to do one cycle of its rising and falling edges. Once the ramp has started a cycle of increasing voltage, it can not be retriggered until it has completed the full ramp and returned to 0V. This is illustrated in figure 6 for a single cycle and in figure 7 for multiple cycles.Not only do you have control over the starting point of the ramp, but the amount of time that the ramp takes to reach its maximum voltage (the right hand side of the CRT screen) can be adjusted with the timebase control. In essence, you have a “window”. You can move the window to any point on a waveform with the triggering circuit and you can change the size of the window with the timebase.The time-base control allows you to set the time / division that the beams takes to scan acrossthe screen. Just like the voltage selector, there is a calibration knob in the middle of the control. Unless the vernier (calibration knob) is 'clicked' in to its most clockwise position, the time per division is unknown.When set to AUTO (automatic) triggering, the oscilloscope will always show a trace. However, when you use a manual triggering mode (DC, AC), many strange things can happen. For example, if the triggering voltage or level is set to +10V and the waveform never exceeds +5V, the triggering circuit will never trigger and the screen will stay blank.You may think that in a condition of no triggering, you would still have a bright dot on the screen because the electron beam would go to its 'home' or undeflected position. Since the oscilloscope is designed to work with a moving electron beam, a stationary beam can very quickly 'burn' a hole in the phosphorous coating of the screen. To prevent this, there is a ' blanking' circuit which turns off the electron beam. Blanking occurs when there is no triggering or when the electron beam is sweeping from the right edge back to the left side of the screen. Time measurements are done the same way as voltage measurements. As long as the timebase is calibrated you multiply the number of divisions by the number of seconds per division to get the total time difference. Phase measurements are done by comparing the measured time to the period of the waveform.Oscilloscope Two Channel OperationYou can view two voltage waveforms at once by using two Y-axis (vertical) input channels. The individual channels are sometimes labelled as '1' and '2' or as 'A' and 'B'. Since there is only one electron beam, you have to share its drawing time between both waveforms. This may be accomplished using either the chop or alternate modes.When in the chop mode (figure 8), the oscilloscope displays a little bit of channel A, then a little bit of B, then A, then B ....during a single sweep of the electron beam. If you increase the timebase to about 1:s/division, you can start to see the individual pieces as it chops between one channel and the other channel.Figure 8. Chop ModeIn the alternate mode (figure 9), the oscilloscope will sweep the electron beam twice across the screen. The first time it will draw the signal from channel A and the next time from channel B. At very low timebase settings, you can see it draw one channel and then the other in successive passes.Note: When you use the alternate function, the two waveforms that you see are from different points in time and the triggering circuit has to trigger twice.Figure 9.Alternate ModeFigure 10. A Simple Oscilloscope Block DiagramThe reason that you can see a non-flickering image on the screen is because the phosphorous coating on the CRT has persistence. In essence, the phosphorous acts like a low pass filter and averages several images that are drawn on the screen.By viewing two signals at a time, you can measure relative time differences. By combining a voltage and phase measurement (relative to the appropriate reference), you can measure a phasor value.With a two channel oscilloscope, you have the ability to trigger on each waveform andelectronically switch (chop or alt) between them as well. A block diagram of a oscilloscope has now become as shown in figure 10.Some oscilloscopes offer a way to alternately trigger as depicted in figure 11. When combined with alternate displaying, you can stably display two waveforms of any frequency by alternately showing each channel and triggering on the channel that is being drawn. This way, the oscilloscope is acting like a two beam scope with both waveforms triggering at the same voltage and slope. However, there is no way to know what the relationship is between one waveform and the other when using alternate triggering.Figure 11. Stable Triggering of Two Different FrequenciesIf you have two waveforms that do not have the same frequency, it is still possible to show them as two stable waveforms on a normal oscilloscope. In figure 11, you will notice that if the triggering occurs at the 'X', both waveforms are in phase (ie. at the same phase each time the timebase triggers). The condition for a stable display is not that two waveforms have to be of exactly the same frequency, but that when they are triggered, they have to be in phase.Or n A f1 = m A f2 where n, m are integers. That is not necessary, but it is sufficient. There are many other ways to achieve a stable trace when you consider that the trigger circuit will wait for the next triggering point. There is also a control on some oscilloscopes, called 'hold off', which allows you to add a delay between the end of the trace being drawn and the time when the triggering circuit starts to look for the next triggering point. That can be used to stabilize the display under some circumstances.Remember that all of this applies only for repetitive waveforms that are properly triggered. If the triggering is not stable, or the waveform is not repetitive, you will see a constantly moving image or several images offset and superimposed.A slightly more complicated block diagram of an oscilloscope, with the typical functions found in the laboratory, is illustrated in figure 12.AccuracyThere are many factors affecting the accuracy of oscilloscope measurements.There are errors due to the input channel voltage divider, timebase control, the use of magnifiers, the accuracy to which the CRT deflection can be read, beam thickness, temperature etc. The voltage divider error will be the same for all readings that are done on the same timebase and voltage range, but may be different each time the range is changed. Measurements over only two divisions can incur two to three times the error of those made over the centre eight divisions.If the phase angle is used in a trigonometric function, this error can be multiplied by the slope of the function. Consider that the tangent of a 1% phase error entered at 85 degrees is much worse (20%) than the same 1% error on a sine function (0.2%) at the same angle. To get a feel for this look at the Taylor expansion of the trigonometric functions.It is wise to consult the user manual for a particular instrument’s accuracy specifications.Figure 12. Oscilloscope Block Diagram2.Iwatsu Model SS-5702 OscilloscopeAccuracyThe Iwatsu SS-5702 oscilloscope is used in this laboratory. All measurements on the graticule should be made over as many divisions as possible. For simplicity, assume the Iwatsu SS-5702 oscilloscope’s error is ±5% for a measurement on either the vertical or horizontal scales over eight divisions.Front Panel ControlsThe front panel controls, shown in figure 13, will be described in the remainder of this section.Figure 13. The Iwastu SS-5702 OscilloscopeFront Panel ControlsThe power, trace rotation, intensity, focus, and scale illumination controls are located at the bottom centre.The vertical controls and input selection are on the left side.The horizontal controls and horizontal input selection are on the right side.The triggering selection section is at the bottom right.The power on/off, trace rotation, intensity, focus and scale illuminationTrace rotation has to be checked with just a straight line across the screen of the oscilloscope. The intensity should be adjusted to a mid point (or more clockwise).The focus of the beam can be done while observing the straight line display.The scale illumination can be adjusted to the operator’s preference.Vertical InputsVertical channels are used for measuring voltage.The beam is deflected vertically as a result of the signal being applied to the vertical input of the channel (CH).CH-1 and CH-2 are the labels for the two vertical inputs.Each channel has a position control, a range selection switch, a pull “x5” knob, a coupling selector switch (AC/GND/DC), and an input connector.Each channel Range switch (VOLTS/DIV) has a smaller knob in the middle of the Range Selector Switch. And there is an arrow showing that the Range Selector Switch is in the “CAL” position when rotated fully clockwise.In the centre of the two channel sections is the channel selection switch. You can choose to have CH-1, CH-2, both (DUAL), or ADD.In addition, CH-2 has a “Polarity” switch. You push the polarity switch in to “INVERT” the polarity of the signal being displayed. The “NORM” or out position is the normal position of the polarity switch.Horizontal InputsThe “EXT” (HORIZONTAL IN) can be supplied a voltage directly via the connector at the bottom right of the panel, or the Horizontal can be driven by an internal timebase circuit which generates the voltage.TimebaseIn the timebase mode, the horizontal signal is from an internal source which changes linearly with respect to time. Hence, the beam is deflected to give us a calibration of time for the horizontal scale.The position control, the pull “x5” magnifier switch, the time range selection switch, and the range “CAL” knob all affect the X-axis of the display.Trigger SourcesThe TRIGGER SOURCE may be selected from one of three sources, CH-1, CH-2, or EXT. Look at the bottom right of the control panel.Calibration SourceThe Calibration Source is an internal source, available on the oscilloscope.Look at the bottom right side of the front panel. The output is labelled 0.3 V.This is a 1000 Hz, square-wave ( the 50 % duty cycle is not accurate ).Figure 14.Differential Measurement Example3.Oscilloscope ApplicationsVoltage and Time MeasurementsNote:The oscilloscope measures divisions of deflection not voltage or time. From the divisions of deflection you can calculate the time or voltage.Differential MeasurementsAn important application of the oscilloscope is differential measurements. Such measurements are necessary because both vertical channels have one terminal connected to the chassis common (ie single ended). To measure a floating (off ground) voltage you have to use the “invert and add” feature of the oscilloscope. For example, in figure 14, to measure V1:Channel A measures (V1 + V2) relative to ground while channel B measures V2 relative to ground. By pushing the invert button you negate the voltage displayed on channel B. Then you can add the channels together with the “ADD” display mode. The waveform now displayed is (V1 + V2) + (- V2) = V1.Bandwidth (-3 dB) MeasurementThis measurement is easily done by first finding the maximum gain (max. VOUT / VIN atfrequency To) and adjusting the oscilloscope so that the sinewave fills seven divisions peak to peak. A -3dB point can be found by increasing and/or decreasing the frequency until the gain is reduced to /2 . If the input voltage has remained constant this will occur when the output voltage is five divisions peak to peak.The frequency is then simply read with a frequency counter or the oscilloscope. Not by reading the dial of the signal generator.RisetimeThe risetime indicates how quickly a circuit responds. The risetime is the time it takes a waveform to go from 10% of the voltage range to 90% of the voltage range. This is in response to a square wave and the output voltage must settle to a steady-state voltage (0% and 100%). Most oscilloscopes have dotted lines on the graticule marking the 10% and 90% points to aid in this measurement. Usually these dotted lines assume that 0% is the lowest line of the graticule and 100% is the highest line. The measurement, as shown in figure 15, also includes the risetime of the oscilloscope and the squarewave source.Figure.15 A Risetime and Phase MeasurementPhasePhase is most accurately measured when the waveform is as large as possible and the difference is measured at the zero crossings. Typically the timebase is uncalibrated so that a 180 degree section of the waveform is expanded to the full 10 divisions of the graticule. Then the sign of the phase can be determined by observing more than one period of both waveforms. Both waveforms must be symmetrical about the centre line of the graticule. The angle is determined by: phase = # of divisions * 180 degrees / 10 divisions.。

多层平板屏蔽体的多目标优化设计

多层平板屏蔽体的多目标优化设计
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一种基于相邻耦合误差的多电机同步控制策略

一种基于相邻耦合误差的多电机同步控制策略

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如果对于所有的轴 ε i (t ) = 0 ,则性能指标式(2)即可 实现。 基于上述跟踪误差和同步误差的定义,n 台电 机的同步控制问题可以描述为:设计控制器,使其 能 够控制转矩,使 速度跟踪误差 e i( t) 和同步误差 ε i (t ) 收、敛趋于零。 对于多轴(n>2)同步控制系统, 如果每一轴的控 制仅仅考虑另外一轴的状态,则在本质上成为一种 主从结构控制,这种控制结构存在固有的缺陷即跟 踪性能差,易受负载扰动的影响。因此,对于多于 两台电机的同步控制系统,每一轴的控制应该至少 附加考虑其他两个轴的状态。这里将相邻两轴称为 最小相关轴。 为 此 提出 如下 基于最小相关轴数目的同步控 制策略:每个轴的控制转矩应能使这一轴的跟踪误 差和这一轴与相邻两轴的同步误差稳定收敛,即第 i 轴的控制转 矩 应能 够使 ei (t ) → 0 的同 时 , 使 第 (i −1) 、 从而使同步误差 ε i−1 (t ) i 、(i +1) 轴保持同步, 和 ε i (t ) 收敛于零。 根据该思想,在对每一轴实施控制时,仅仅考 虑其它两个轴的状态,这将会大大方便了每一轴的 控制。在该同步控制基本思想的基础上,本文提出 了基于相邻耦合误差的同步控制算法。
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中 国 电 机 工 程 学 报
第 27 卷
1
基于最小相关轴数目的同步控制思想
对于具有 n 台电机(轴)的控制系统,定义第 i 轴的跟踪误差为 ei (t ) = xid (t ) − xi (t )
基金项目:国家自然科学基金项目(50477042);高等学校博士学科点 专项科研基金项目(20040422052);山东省自然科学基金项目(Z2004G04)。 Project Supported by National Natural Science Foundation of China(50477042).

三星MITSUBOSHI皮带选型手册

三星MITSUBOSHI皮带选型手册

三星MITSUBOSHI皮带选型手册V601-CRIndustrial Power Transmission ProductsConveyor Belts Engineering PlasticsT e c h n o l o g y N a t u r e S o c i e t ySafety PrecautionsPlease read all the warnings!●Please take all necessary precautions when using our products. Also, Please review relevant productcatalog and design documents, etc.Significances of safety precautions are categorized as follows:Plastic Conveyor BeltsEngineering Plastic ProductsCouplingsI N D E XCONTENTSSafety Precautions.....................................................P1,2 1. Power Transmission Beltsand Related Products................................P4~682. Conveyor Belts...........................................P69~773. Engineering Plastic Products...............P78~85 Global Factories & Sales Offices...............................P88 34Round Tooth Timing BeltBelt Type, Dimensions & Product Code . . . . . . . . . . . . . . . . P13SUPER TORQUE Timing Belt GN . . . . . . . . . . . . . . . . . . P14, 15SUPER TORQUE Timing Belt G . . . . . . . . . . . . . . . . . . . . . . . . . P16SUPER TORQUE Timing Pulley . . . . . . . . . . . . . . . . . . . . . . . . . . P17MEGA TORQUE Timing Belt G&U . . . . . . . . . . . . . . . . . . P18, 19H Series . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P20green eco ?Series . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P27Application Examples (5)6Lineup of Belts and Related Products (7)8Industrial Power Transmission Product Selection Chart ......P9,10Dimensions .......................................................................... .........P11Trapezoidal T ooth Timing BeltBelt Type, Dimensions & Product Code . . . . . . . . . . . . . . . . P21Timing Belt G ......................................................P22 24Timing Belt U .......................................................P25, 26green eco ?Series ......................................................P27Timing Pulley ........................................................P28, 29Other type of Timing BeltLONG-SPAN Timing Belt ............................................P301Synchronous Power TransmissionClassical V-Belt / Red label V-Belt .............................P32ClassicalV-Belt / Red label V-Beltfor DIN2215 / ISO4184.........................................P33 36MAXSTAR WEDGE V-Belt ............................................P37MAXSTAR WEDGE V-Belt for RMA / MPTA ...............P38Narrow V-Belt for DIN7753 / ISO4184........................P39SUPER VS ?Belt (Variable Speed Belt)......................P40MAXSTAR WEDGE Bushing Pulley .....................P41, 42e-Power ?Belt .............................................................P43RIBSTAR Belt G (Rubber V-Ribbed Belt)...................P44RIBSTAR Belt U (Polyurethane V-Ribbed Belt) ........P45RIBSTAR Pulley (V-Ribbed Pulley) .............................P46FLEXSTAR ?Belt ............................................. ..............P47SUPERFLEXSTAR ?Belt ..............................................P48FLEXSTAR ?Belt J .......................................................P49POLYMAX Belt ..............................................................P50MB Belt .........................................................................P51STARROPE ?, SUPER STARROPE ?............................P52PRENE V-ROPE, PRENE HEXAGONAL-ROPE ..........P52Flat Belt .........................................................................P532Frictional Forced Power TransmissionChemi-Chan ?(High Performance Miniature Coupling). . . . . . . . . . . . . . P55HYPERFLEX ?Coupling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P56TSCHAN ?Coupling NOR-MEX ?. . . . . . . . . . . . . . . . . . P57, 58TSCHAN ?Coupling S . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P59, 603Direct Connected Power TransmissionSLEEVE ROLL(Polyurethane moldedproduct)................................P62STARLOCK ?(Shaft Fastener)..............................P63, 64For Timing Belt .........................................................P67For V-Belt ...................................................................P685Troubleshooting for Power Transmission Products1. Power Transmission Belts and Related Products4Other Related Products5S y n c h r o n o u s P o w e r T r a n s m i s s i o n &D i r e c t C o n n e c t e d P o w e r T r a n s m i s s i o nApplicationExamplePolisher (SUPER TORQUE Timing Belt)Large Knitting Machine (SUPER TORQUE Timing Belt)Food Processor (RIBSTAR Belt)Heavy Duty Polisher (MAXSTAR WEDGE V-Belt)Combine (V-Belt for Agricultural Machines)Copier (Rubber Timing Belt)F r i c t i o n a l F o r c e d P o w e r T r a n s m i s s i o n6Marine Engine (TSCHAN ?Coupling S )Packaging Machine(Polyurethane Double Timing Belt)Plastic Card Vender (Polyurethane Timing Belt)Actuator (Chemi-Chan ?)Dryer (V-Belt)NC Lathe (Multi-POLYMAX Belt)NC Lathe (RIBSTAR Belt)To show the application examples clearly, safety covers are removed in the pictures on this page.Always put on the safety covers when in use.7Lineup of Belts & Related ProductsProduct Name & ApplicationTiming PulleyTSCHAN ?Coupling NOR-MEX ?TSCHAN ?Coupling S Chemi-Chan ?(High Performance Miniature Coupling)Product Name & ApplicationSUPER TORQUE Timing Belt (General use)MEGA TORQUE Timing Belt G&U (Heavy-duty use)LONG-SPAN Timing Belt (Endless & Open-End)Multi V-Belt(General use : Joined V-Belt)Less Thickness Wrapped V-Belt LA, LB, LCAgricultural useBARE-BACK ?V-Belt(Agricultural use, Washer and Dryer )MAXSTAR WEDGE V-Belt (Heavy-duty use)()green eco ?SeriesPerforated V-ROPEe-POWER ?Belt (Notched type)V-Belt with lugs (Agricultural use)HEXAGONAL V-BeltT i m i n g B e l tT i m i n g P u l l e yC o u p l i n gW r a p p e d B e l tTiming Belt G(General use : Rubber)Timing Belt U(General use : Polyurethane)Classical V-Belt (General use)Multi MAXSTAR WEDGE V-Belt (Joined V-Belt for Heavy-duty) 8Product Name & Application Product Name & Application e-POWER ?Belt (Cogged type)MB BUSHINGSLEEVE ROLLSUPER GOLD 1000Raw Edge COGGED V-Belt (Agricultural use)Raw Edge MULTI-PLY V-Belt (Automotive use)Raw Edge COGGED V-Belt (Automotive use)Raw Edge COGGED Variable Speed BeltPOLYMAX Belt(Wide-angle Polyurethane V-Belt)MB Belt(For Sewing Machine)PRENE V-ROPE & HEXAGONAL-ROPE (Polyurethane Rope) RIBSTAR Belt G (Rubber)RIBSTAR Belt U (Polyurethane)FLEXSTAR ?SUPER FLEXSTAR ?(Heavy-duty use)FLEXSTAR ?Belt J (For Conveyor)R a w E d g e B e l tP u l l e yB u s h i n gP o l y u r e t h a n e M o l d e d R i b b e d B e l tP o l y u r e t h a n e B e l t F l a t B e l tOrange labelRaw Edge COGGED V-Belt (Agricultural use)Pulley and Bushing for WEDGE V-Belt Pulley and Bushing for RIBSTAR Belt STARROPE ?& SUPER STARROPE ?(Polyurethane Round Rope)Flat Belt (For Lifter)Flat Belt (General use)■ Industrial Power Transmission Product Selection Chart9A wide range of products are available at Mitsuboshi to suite your needs.Refer to the following property chart when selecting belts.10■ DimensionsThe conditions of use such as pulley diameter and speed affect belt durability.(Use the above table as a reference when selecting belts)Note 2For V-Belt and MAXSTAR WEDGE V-Belt, values are nominal.111112Round Tooth Timing BeltBelt Type, Dimensions & Product Code .......P13SUPER TORQUE Timing Belt GN ............P14,15SUPER TORQUE Timing Belt G ...................P16SUPER TORQUE Timing Pulley .....................P17MEGA TORQUE Timing Belt G&U ...........P18,19H Series ..........................................................P20green eco ?Series ...........................................P27Trapezoidal T ooth Timing BeltBelt Type, Dimensions & Product Code ..........P21Timing Belt G ..........................................P22 24Timing Belt U ...........................................P25, 26green eco ?Series ..........................................P27Timing Pulley ...........................................P28, 29Other Timing BeltLONG-SPAN Timing Belt ................................P301Synchronous PowerTransmission1313For S1.5M, S2M and S3M, polyurethane (U) and rubber (G) models are available. Choose the appropriate material for your application.Sizes not indicated above are also available under limitedconditions.14For S1.5M, S2M, and S3M, polyurethane (U) and rubber (G) models are available. Choose the appropriate material for your application. 15151617 17Shape is a 100mm long round cylinder product.22 teeth and below sizes of S3M are available only in rod shape and C type. Also, 24 teeth and above sizes are available only in A or B type.Standard SizesMaterialsA andB types are made from steel whereas W type is made from cast metal.we revised the standard item lists in April, 2006.18MEGA TORQUE Timing Belt UMEGA TORQUE Timing Belt G。

多通道低相噪同步频率源设计

多通道低相噪同步频率源设计

电路与系统多通道低相噪同步频率源设计胥伟,潘明海,张艳睛(南京航空航天大学电子信息工程学院,江苏南京211106)摘要:针对数字射频存储器(Digital Radio Frequency Memory,DRFM)系统在进行对外部输入信号采集时,对高稳频率源需求问题,提出了一种基于两级锁相环的多通道低相噪同步频率源设计方法,实现了6路在2.26〜2600MHz 范围内任意频率信号输出遥通过线性叠加的方法,理论分析了锁相环中相位噪声的模型,并根据相位噪声的来源进行优化设计遥最后对频率源电路杂散和相位噪声进行测试,测试结果表明该频率源电路输出1.25GHz频率时的杂散抑制优于-60dBc,相位噪声抑制优于-104.91dBc/Hz@500kHz遥关键词:频率源;锁相环;相位噪声;杂散中图分类号:TN95文献标识码:A DOI:10.16157/j.issn.0258-7998.200921中文引用格式:胥伟,潘明海,张艳睛.多通道低相噪同步频率源设计[J].电子技术应用,2021,47(3):97-101,114.英文引用格式:Xu Wei,Pan Minghai,Zhang Yanjing.Design of multi-channel low phase noise synchronous frequency source[J]. Application of Electronic Technique,2021,47(3):97-101,114.Design of multi-channel low phase noise synchronous frequency sourceXu Wei,Pan Minghai,Zhang Yanjing(School of Electronic Information Engineering,Nanjing University of Aeronautics and Astronautics,Nanjing211106,China) Abstract:In order to meet the requirement of high stability frequency source when DRFM(Digital Radio Frequency Memory)systemcollects external input signals,a design method of multi-channel,low phase noise synchronous frequency source based on two-stage PLL is proposed in this paper.Six channels of arbitrary frequency signal output in the range of2.26〜2600MHz are realized.Through the method of linear superposition,the phase noise model of PLL is analyzed theoretically,and the optimal design is carried out ac­cording to the source of phase noise.Finally,the spurious and phase noise of the frequency source circuit are tested.The test re­sults show that the spurious suppression is better than-60dBc and the phase noise suppression is better than-104.91dBc/ Hz@500kHz when the frequency source circuit outputs 1.25GHz frequency.Key words:frequency source;phase locked loop;phase noise;spurious0引言DRFM系统在产生雷达欺骗干扰回波时,需要一个高稳定度的频率源信号用于对外部输入信号的采集与重构。

超特克(Supertex)MD1812和MD1813芯片应用说明书

超特克(Supertex)MD1812和MD1813芯片应用说明书

Supertex inc.AN-H56Application NoteIntroductionThe MD1812 and the MD1813 are two unique composite return-to-zero (RTZ) pulser drivers for ultrasound applications. The ICs have built-in level shifters that provide negative P-MOS gate DC bias and fast AC coupled gate drivesignals. They enable the fast damping functions necessary to generate return-to-zero bipolar pulses, and are also able to keep the zero-state to as long as needed, even to infinity. These kinds of fast return-to zero and DC coupled features are very useful for medical ultrasound imaging equipment, piezoelectric transducer drivers, material flaw detection, ultrasonic NDT detection, and sonar ranger applications, especially for those that need to launch ultrasound in pseudo-random codes. Designing a Pulser with the MD1812/13This a pplication n ote d escribes h ow t o u se M D1812 o r M D1813 to design the basic channel of an ultrasound transmitter with the RTZ feature. The circuit is a single channel ultrasound transmitter using the MD1812 or MD1813 to drive TC6320 & TC2320 MOSFETs. It can generate fast return to zero waveforms. The output of high voltage to transducer has ±2A source and sink current capability. A CPLD programmable logic circuit and on-board 40MHz crystal oscillator generate a fast logic signal to control the pulse circuit. The CPLD hasa six-pin JTAG connection for Xilinx’s USB or a convenient parallel-port programming link cable. The circuit consists of one MD1812K6 or MD1813K6 in a 16-lead 4x4x0.9mm QFN package, driving TC6320FGs and TC2320FGs, two complementary high-voltage P and N- channel MOSFETs in one single SO-8 package. The input stage of the MD1812/13 is a high-speed level translator that is able to operate with logic input signals of 1.2V to 5.0V amplitude. In this circuit, the CPLD output logic is typically 3.3V. An adaptive threshold circuit is used with the OE pininside of the MD1812 to set the level translator threshold to the middle of the input logic 0 and logic 1 levels. The OE pin serves a dual purpose. First, its logic 1 level is used to compute the threshold voltage level for the channel input level translators. Second, when OE is low, the outputs are disabled, with the A and C outputs high and the B and Doutputs low (for MD1812 only). This assists in properly pre-charging the coupling capacitors that may be used in series in the gate drive circuit of external PMOS and NMOS FETs. The MD1812/13 level translator uses a proprietary composite drive circuit, which provides DC coupling, together with high-speed operation. The output pin, OUT C , is designed to drive the return-to-zero PMOS FET through a capacitor as fast as an AC coupling gate driver, and OUT G provides delayed DC coupling negative biased gate control to the same PMOS FET. The OUT C swings between V H and V L voltages, while OUT G is within V SS or V NEG levels. Note that the OUT C and OUT G pins of one chip are designed to drive together forone PMOS FET, and that the PMOS FET source is typicallyconnected to the same potential of the MD1812/13 V SS voltage. Each of the output stages of OUT A , OUT B , OUT C & OUT D of MD1812/13 are capable of peak currents of up to ±2.0A, depending on the supply voltages used and load capacitance. But a 2kΩ resistor, R36, must be between OUT G and the gate of the PMOS FET, which is driven by the OUT C through a capacitor. This configuration provides the optimal series resistance value of the gate DC bias driver circuit.The output stage of the MD1812/13 has separate power connections enabling the output signal high and low levels to be chosen independently from the driver supply voltages. As an example, the input logic levels may be 0V and 1.8V, the control logic may be powered by +5V and –5V, and the output high and low levels may be varied anywhere over the range of +5V to -5V. In this design example, MD1812/13’s V DD and V H are both powered by +10V, V SS and V L are grounded, and V NEG is –10V. The source pin of the RTZ PMOS FET driven by the OUT C and OUT G pins is connected to ground.PCB Layout TechniquesIt is very important that the slab at the bottom of the IC package, which is the IC substrate “pin”, be externally connected to the V NEG pin to make sure it always has the lowest potential in any condition.Designing An Ultrasound Pulser with MD1812/MD1813 Composite DriversBy Ching Chu, Sr. Application EngineerUse high-speed PCB trace design practices that are compatible with the circuit’s operating speed. The internal circuitry of the MD1812/13 can operate at up to 100MHz, with the primary speed limitation being due to load capacitance. Because of this high speed and the high transient currents that result when driving capacitive loads, the supply voltage bypass capacitors should be as close to the supply pins as possible. The V SS and V L pins should have low inductance feed-through connections that are connected directly to a solid ground plane. If these voltages are not zero, they will require bypass capacitors similar to the positive power supplies. The V DD and V H supplies determine the output logic levels. These two pins can draw fast transient currents of up to 2.0A, so they should be provided with a low-impedance bypass capacitor at the chip’s pins. A ceramic capacitor of up to 1.0µF may be appropriate. Minimize the trace length to the ground plane, and insert a ferrite bead in the power supply lead to the capacitor to prevent resonance in the power supply lines. A common voltage source and local decoupling capacitor may be used for the V DD and V H pins, which should always have the same DC level applied to them. For applications that are sensitive to jitter and noise, insert another ferrite bead between V DD and V H and decouple each pin separately.Pay particular attention to minimizing trace lengths and using sufficient trace width to reduce inductance. Surfacemount components are highly recommended. Since the output impedance of this driver is very low, in some cases it may be desirable to add a small value resistor in series with the output to obtain better waveform integrity at the load terminals. This will, of course, reduce the output voltage slew rate at the terminals of a capacitive load. Pay particular attention to the parasitic coupling from the driver’s output to the input signal terminals. This feedback may causeoscillations or spurious waveform shapes on the edges of signal transitions. Since the input operates with signals down to 1.2V, even small coupling voltages may cause problems. Use of a solid ground plane and good power and signal layout practices will prevent this problem. Also ensure that the circulating ground return current from a capacitive load cannot react with common inductance to create noise voltages in the input logic circuitry.Testing the Ultrasound Pulser The MD1812 RTZ pulser design example is tested with the following power supply voltage and current limiting: V PP 0 to +100V 5mA, V NN 0 to -100V 5mA, V DD = +10V 50mA, V NEG -10V 5mA, V CC +3.3V, 90mA.The HV OUT signal appears at the SMA connector J6. There is a 5:1 attenuation of the signal, due to the value of resistor R11. When driving a real transducer load, the value of this resistor should be reduced in value to match the load impedance.The HV OUT signal passes through jumper J5, which can be used to terminate the HV OUT signal in a dummy load, comprising a 220pF capacitor in parallel with a 1kΩ resistor. When an external load is connected, the dummy load is not required, and J5 can be configured to pass the signals straight through to the output connector J6.All the on-board test points are designed to work with an active oscilloscope probe, such as the Tektronix P6243 1MΩ active probe. Because TP7 is connected to the HV OUT , where potentially damaging voltages could be present, make sure that V PP /V NN does not exceed the probe limit. If using another type of high impedance oscilloscope probe for the test points, ensure that the ground lead connections to the circuit board ground plane are as short as possible.There are multiple frequency and waveform combinations that can be selected as bipolar pulses, PW or CW waveforms. An external clock input can be used if the on-board 40MHz-oscillator is disabled. The external trigger input can be used to synchronize the output waveforms. There are five push buttons for selecting demo waveform, frequency, phase, and MD1812 chip enable functions. Color LEDs indicate the demo selection states. The CH1 output allows the monitoring of one of the 5 inputs (IN A , IN B , IN C , IN D or O E ) of the MD1812/13 via the select button. The MD1812 and the MD1813 are very similar in function. The only differences between them are the control of the OE (MD1812) vs VLL (MD1813) pin and their logic functions. Please read their data sheets for the details. In this design example, the CPLD program is using an on-board solder jumper, R34, to sense the difference and works accordingly. The example MD1812/13 pulser circuit schematic, detailsignals definitions, and some measured waveforms areshown below.Waveform C, 20MHz, 8 cycles Load: 220pF//1kPulser Circuit SchematicWaveform AWaveform CWaveform BWaveform DOE INA INB INC INDHV OUTV PPV NNOE INA INB INC INDHV OUTV PPV NNOE INAINBINC INDHV OUTV PPV NNOEINAINBINC IND HV OUTV PPV NNNote: The duty cycle of the PW burst is set about 0.2% for limitedpower dissipationNote: The duty cycle of the PW burst is set about 25% at ≤5.0MHz forlimited power dissipation.AN-H56MD1812/13 Reference DesignJ 6X D C RJ E X = L oFig. 1 Waveform of 2.5MHz Fig. 2 Waveform of 5MHzFig. 3 Waveform of 10MHz Fig. 4 Waveform of 10MHz InvertingFig. 5 Waveform of 20MHz 8 Cycles Fig. 6 Waveform of 5mHz & Delay ReadingsFig. 7 Waveform of 10MHz(at IN C , OUT C , OUT G , and P- Gate, V DD = 12V, V NEG = -10V)Fig. 8 Waveform of 5MHz(at IN C , OUT C , OUT G , and P- Gate, V DD = 5V, V NEG = -10V)Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//)©2013 Supertex inc.All rights reserved. Unauthorized use or reproduction is prohibited.Supertex inc.。

可实现低频输电系统不对称故障穿越的M3C_电容电压均衡控制策略

可实现低频输电系统不对称故障穿越的M3C_电容电压均衡控制策略

第51卷第23期电力系统保护与控制Vol.51 No.23 2023年12月1日Power System Protection and Control Dec. 1, 2023 DOI: 10.19783/ki.pspc.230464可实现低频输电系统不对称故障穿越的M3C电容电压均衡控制策略郑 涛1,康 恒1,宋伟男2(1.新能源电力系统国家重点实验室(华北电力大学),北京 102206;2.国网辽宁省电力有限公司大连供电公司,辽宁 大连 116000)摘要:低频输电作为一种新型输电技术,在海上风电送出、新能源场站送出等多个场景具有良好的应用前景。

但在不对称故障下,故障侧功率不对称将严重影响模块化多电平矩阵变换器(modular multilevel matrix converter, M3C)的电容电压均衡,对低频输电系统安全稳定运行产生不利影响。

为此,提出了一种可实现低频输电系统不对称故障穿越的M3C电容电压均衡控制策略。

首先,介绍M3C的系统结构及双αβ0数学模型,并分析不对称故障下电容电压不均衡的原因。

然后,基于双αβ0数学模型针对输电线路不对称故障情况计算桥臂功率不均衡分量的表达式,通过M3C功率平衡关系引入电流补偿分量,消除桥臂功率的不均衡,并得到适用于不对称故障的环流控制目标,进而通过环流控制实现故障下M3C电容电压的均衡。

最后,搭建基于M3C的低频输电系统仿真模型验证所提控制方案的可行性和有效性。

关键词:低频输电;模块化多电平矩阵变换器;环流控制;电容电压均衡Asymmetric fault ride-through control strategy for low-frequency transmission systems realizing the capacitor voltage balance of modular multilevel matrix convertersZHENG Tao1, KANG Heng1, SONG Weinan2(1. State Key Laboratory of Alternate Electrical Power System with Renewable Energy Sources (North China ElectricPower University), Beijing 102206, China; 2. Dalian Power Supply Company, State Grid LiaoningElectric Power Company, Dalian 116000, China)Abstract: As a new transmission technology, low frequency transmission has good application prospects in several scenarios such as offshore wind power transmission and new energy field station transmission. However, with asymmetric faults, the power asymmetry on the fault side will affect the capacitor voltage balance of the M3C and affect the stable operation of the low frequency transmission system. Therefore, an M3C capacitor voltage balance control strategy that can realize asymmetric fault ride-through in low frequency transmission systems is proposed. First, the system structure and dual αβ0 mathematical model of an M3C are introduced, and the causes of capacitor voltage imbalance under asymmetric faults are analyzed. Second, based on the dual αβ0 mathematical model for the transmission line asymmetric fault case, the expression for the bridge arm power imbalance component is calculated. Then the M3C power balance relationship introduces the current compensation to eliminate the power imbalance. It also obtains the circulating currents control objective applicable to asymmetric faults. Then through the loop current control it achieves the M3C capacitor voltage balance under faults. Finally, a simulation model of the M3C-based low frequency transmission system is built to verify the feasibility and effectiveness of the proposed control scheme.This work is supported by the Joint Fund of National Natural Science Foundation of China (No. U2166205).Key words: low frequency transmission system; modular multilevel matrix converter; circulating currents control;capacitor voltage balance基金项目:国家自然科学基金联合基金项目资助(U2166205)郑涛,等可实现低频输电系统不对称故障穿越的M3C电容电压均衡控制策略- 131 -0 引言低频输电技术,也被称为分频输电技术,通过AC/AC变频器将传统工频交流电变换为低频交流电进行输送,或者直接对新能源场站低频交流电进行送出[1-3]。

基于双派克变换的新型三相锁相环技术

基于双派克变换的新型三相锁相环技术

基于双派克变换的新型三相锁相环技术李研达【摘要】Aiming at the problem that the double frequency harmonic perturbation exists in the output of three phase locked loop (PLL) when the power grid voltage has unbalanced fault, a novel PLL based on the double Park transformation was proposed.The positive and negative sequence Park transformation for the voltage signal was carried out, respectively.In addition, the DC component in the negative sequence Park transformation was multiplied by the transformation matrix, and the double frequency component in the positive sequence Park transformation could be eliminated through the cross decoupling.The fundamental positive DC component was effectively extracted, and the accurate tracking of phase and frequency could be realized under the unbalanced grid voltage condition.A simulation model was established in theMatlab/Simulink environment, and the steady state response of PLL under the unbalanced grid voltage was analyzed.The results show that compared with the traditional PLL, the proposed PLL can detect the voltage phase and frequency information more accurately under the grid fault.%针对电网电压不平衡故障时,三相锁相环输出存在二倍频谐波扰动问题,提出了一种基于双派克变换的新型锁相环.对电压信号分别进行正序派克变换和负序派克变换,并将负序派克变换中的直流量乘以变换矩阵,通过交叉解耦可消除正序派克变换中的二倍频分量,有效提取了基波正序直流分量,可实现电网电压不平衡条件下的相位和频率准确跟踪.在Matlab/Simulink环境中建立了仿真模型,分析了锁相环在电网不平衡下的稳态响应.结果表明,本文提出的锁相环较传统的锁相环而言,在电网故障下更能精确检测电压相位和频率信息.【期刊名称】《沈阳工业大学学报》【年(卷),期】2017(039)003【总页数】5页(P253-257)【关键词】锁相环;三相不平衡;二倍频;双派克;正负序分量;直流量;交叉解耦【作者】李研达【作者单位】安阳师范学院物理与电气工程学院河南安阳 455000;河南省光伏并网发电及储能技术工程实验室, 河南安阳 455000【正文语种】中文【中图分类】TM615锁相环能够有效检测电网电压信号的幅值、相位和频率等信息,目前被广泛应用于新能源并网变流器的控制算法中,以实现新能源变流器的输出电流与电网电压同步[1-2].在并网变流器控制中,通常将三相电流信号转换成正交的两直流信号进行控制,以达到PI无静差控制,其中从三相转换到两相的参考相位都来自于锁相环,因此,锁相环的输出精度对电流信号转换起到了至关重要的作用[3].为提高新能源并网的稳定性和可靠性,对锁相环功能提出了更高的要求,需要锁相环在各类扰动下均能有效地跟踪电网电压正序分量的幅值、相位和频率,同时具有较高的检测精度和响应时间.鉴于锁相环在新能源并网控制中的重要性,国内外学者展开了一系列研究.文献[4-6]介绍了基于过零鉴相器的锁相环,通过检测电网电压的过零点和周期实现电网电压相位的跟踪,但是当电网电压处于扰动情况下会对过零点的检测产生影响,使结果存在一定的偏差;文献[5-7]提出了一种基于乘法鉴相器的锁相环,将输出信号和输入信号相乘,并通过滤波再经PI控制器即可得到输出相位信号,以实现锁相功能,此方法简单易于实现,但环路滤波器的带宽需设计的很窄,不利于响应速度;文献[8-9]提出了一种基于同步坐标系的锁相环,根据派克变换实现三相到两相同步旋转坐标系的转换,通过控制无功分量即可实现锁相.该锁相环在电网电压平衡时,能够取得较好的锁相效果,然而在电网电压出现扰动导致三相不平衡时,输出因含负序分量而导致出现大量的二次谐波分量,从而影响系统的稳定性.上述研究在理想电网情况下都具有较好的稳态和动态特性,但是当电网处于不平衡扰动时,如何保证输出的稳定性和精确性,给锁相环的设计带来了一定的挑战.本文提出一种基于双派克变换的新型三相锁相环,对电网电压信号分别进行正序派克变换和负序派克变换,提取了基波正序直流分量,从而准确跟踪电网电压不平衡条件下的相位和频率信息,并用Matlab/Simulink软件进行了仿真验证.实际的电网电压会因各种故障或扰动而导致不平衡,而不平衡的电网电压会导致锁相环的输出产生各种误差,从而影响新能源并网系统的稳定性[10].电网电压在不平衡情况下的表达式为式中:ea、eb、ec为电网三相实际电压;Em为电网电压正常时的幅值;θ为a相电压的相位;χ和γ分别为不平衡常数.式(1)在两相坐标系中的表达式为式中,eα、eβ为电网电压在静止坐标系下的电压.由式(2)可以看出,当电网电压处于三相不平衡时,两相静止坐标系下的分量不是完全正交的.根据派克变换,式(2)在两相旋转坐标系下q轴表达式为eq= Emsin式中,为锁相环的实际输出相位角.一般而言,锁相环的误差信号较小,则可近似假设=2θ,则式(3)可以简化为eq=Emsin从式(4)可以看出,锁相环输出的无功直流分量因电网电压不平衡而产生了电网电压两倍频率的误差信号,该误差信号的幅值和电网电压基波信号幅值相等,会给锁相环的输出精度带来较大的误差.针对二倍频误差信号,可利用低通滤波器将其滤除,但截止频率设计过低会影响锁相环输出的动态性能.为了解决上文描述中的二倍频问题,本文提出了一种基于双派克变换的三相锁相环.根据电路原理中三相稳态电路的基本思想,可将任意三相信号分解为对称的三相正序分量、三相对称的负序分量以及三相对称的零序分量[11-12].对于零序分量而言,其不影响输出结果,可不考虑,故只需对分解的正序分量和负序分量分别进行正序派克变换和负序派克变换,将其分解为直流分量和二倍频交流分量.对输入信号进行两次坐标变换,即对输入信号的正负序分量分别进行派克变换可得式中分别为正序和负序派克变换,;分别为静止坐标系下的正序分量和负序分量;分别为正负序轴对应下的电压幅值;ω为电网电压的角频率;φ为相位变化角度.电压矢量经正序派克变换和负序派克变换时,其与旋转参考坐标轴的关系如图1所示.根据假设条件及坐标关系,可得到近似关系式,即将式(9)代入式(6)和式(8)可得由式(10)和式(11)可以看出,正序派克变换中的二倍频分量是由负序派克变换的幅值分量乘以变换矩阵得到的,负序派克变换中的二倍频分量是由正序派克变换的幅值分量乘以变换矩阵得到.因此,可以将上述二倍频分量通过交叉解耦进行消除,解耦后静止坐标系下的正序分量则为所以只需要控制旋转坐标系下的q轴分量为零,即可实现电网电压在不平衡状况下的相位跟踪.基于上述分析,本文提出的双派克变换锁相环结构图如图2所示,图2中,Tαβ为三相到静止两相的坐标变换矩阵,LPF为低通滤波器,VCO为锁相环的压控振荡环节.通过双派克变换的自解耦模型提取电网电压不平衡条件下的基波正序分量,同时实现了二倍频的抑制,进一步使有效的基波无功正序分量经过环路滤波器和压控振荡器,达到电网电压不平衡条件下的跟踪.根据上述理论分析,在Matlab/Simulink中搭建了基于双派克变换的三相锁相环仿真模型,并与传统的三相锁相环进行了对比分析.假设电网电压为三相不平衡,各相表达式分别为图3为输入的三相不平衡电网电压波形,图4为锁相环输出的无功直流分量,也即是锁相环环路滤波器的输出信号,从图4中可以看出,针对传统的三相锁相环而言,当电网电压处于不平衡时,会造成环路滤波器的输出信号含有大量的二次谐波分量,而采用本文提出的双派克变化自解耦方法能够有效抑制二倍频谐波分量,输出无脉动的直流信号.图5和图6分别为锁相环输出频率和正弦信号的对比分析结果,仿真对比说明,传统锁相环在电网不平衡时,输出频率因二倍频分量存在,波动较大,使得输出的正弦信号也发生畸变,而本文提出的基于双派克变换的锁相环具有更好的二倍频谐波抑制和跟踪效果,能够实现锁相环在电网电压不平衡条件下的跟踪.图7为提出算法锁相环的输出相位稳态波形,并将其与幅值缩小的a相电压波形进行了对比分析(其中仿真图中a相电压幅值缩小至了6.2 V).从图7中可以看出,锁相环输出的相位与输入信号保持了完全的一致,说明提出的锁相环结构能够很好地跟踪输入信号的相位,稳态误差较小.针对传统锁相环受电网电压不平衡影响较大的问题,本文提出一种基于双派克变换的新型锁相环技术,锁相环能够在电网电压不平衡时准确地跟踪电网电压正序基波分量的相位和频率,与传统方法的仿真对比结果验证了所提新型锁相环的可行性和有效性,具有一定的借鉴价值.【相关文献】[1]王宝归,李泽泉,林勇,等.并网型电力电子装置数字锁相环研究[J].大功率变流技术,2012(4):39-42.(WANG Bao-gui,LI Ze-quan,LIN Yong,et al.Study on digital phase-locked loop forgrid-connected power electronic device [J].High Power Converter Techno-logy,2012(4):39-42.)[2]王湘明,肇文婷.风电并入微网逆变器合成谐波阻抗谐波抑制 [J].沈阳工业大学学报,2014,36(2):143-148.(WANG Xiang-ming,ZHAO Wen-ting.Synthesis harmonic impedance for harmonic suppression of wind power micro-grid inverter [J].Journal of Shenyang University of Technology,2014,36(2):143-148.)[3]温华生,谢潮.基于dq变换的三相自适应锁相环技术 [J].上海电力学院学报,2016,32(2):151-155.(WEN Hua-sheng,XIE Chao.Three-phase adaptive phase-locked loop based on dq transformation [J].Journal of Shanghai University of Electric Power,2016,32(2):151-155.)[4]戴永辉,洪巧文,蔡逢煌,等.一种基于多过零鉴相器的数字锁相环 [J].电源学报,2012(5):58-62.(DAI Yong-hui,HONG Qiao-wen,CAI Feng-huang,et al.A DPLL based on multi-zero-crossing phase detector [J].Journal of Power Supply,2012(5):58-62.)[5]曹小丽.光伏并网软件锁相技术的研究 [D].南昌:南昌航空大学,2013:1-76.(CAO Xiao-li.The research on software phase locked loop of PV grid-connected generation system [D].Nanchang:Nanchang Hangkong University,2013:1-76.)[6]袁庆庆,戴鹏,符晓,等.单相电力锁相环技术综述 [J].变频器世界,2010(7):43-46. (YUAN Qing-qing,DAI Peng,FU Xiao,et al.An overview of single-phase power phase-locked loop tech-nique [J].The World of Inverters,2010(7):43-46.)[7]胡为兵,熊杰.一种新颖的锁相环的研究 [J].电气技术,2008(1):69-71.(HU Wei-bing,XIONG Jie.Research on a novel phase-locked loop system [J].Electrical Engineering,2008(1):69-71.)[8]洪小圆,吕征宇.基于同步参考坐标系的三相数字锁相环 [J].电工技术学报,2012(11):203-210.(HONG Xiao-yuan,LÜ Zheng-yu.Research on a novel phase-locked loop system[J].Transactions of China Electrotechnical Society,2012(11):203-210.)[9]胡应占,郭素娜.适用于电网不平衡时的广义积分器锁相环设计 [J].电力系统保护与控制,2014(11):148-154.(HU Ying-zhan,GUO Su-na.Design of generalized integrator phase locked loop for unbalanced grid [J].Power System Protection and Control,2014(11):148-154.)[10]闫斌斌,贾焦心.电压不平衡且畸变下基于平均值环节锁相环的研究 [J].黑龙江电力,2015,37(6):483-486.(YAN Bin-bin,JIA Jiao-xin.Research on phase locked loop based on mean value links under unbalanced and distorted voltage [J].Heilongjiang Electric Power,2015,37(6):483-486.)[11]陈国栋,朱淼,蔡旭.一种软件锁相环和电压跌落检测新算法 [J].中国电机工程学报,2014,34(25):5385-5394.(CHEN Guo-dong,ZHU Miao,CAI Xu.A new algorithm for software phase locked-loop and voltage sag detection [J].Proceedings of the CSEE,2014,34(25):5385-5394.) [12]江燕兴,潘逸菎,窦伟.一种用于光伏并网逆变器的高性能锁相环设计 [J].电工电能新技术,2016,35(7):75-80.(JIANG Yan-xing,PAN Yi-kun,DOU Wei.Design of high-performance phase locked loop used ingrid-connected inverter [J].Advanced Technology of Electrical Engineering and Energy,2016,35(7):75-80.)。

双界面耦合增强效应 锂金属

双界面耦合增强效应 锂金属

双界面耦合增强效应锂金属双界面耦合增强效应是指在锂金属电池中,通过合理设计和优化电池结构,使得锂金属电池的性能得到提升的一种机制。

锂金属作为一种理想的负极材料,具有高比容量、低电压平台和丰富的储能资源等优点,被广泛应用于锂离子电池、锂硫电池等领域。

然而,锂金属电池在实际应用中面临着诸多挑战,如锂枝晶的生长、锂金属表面的不稳定性等问题,这些问题限制了锂金属电池的电化学性能和循环寿命。

为了解决锂金属电池中的问题,研究人员提出了双界面耦合增强效应的概念。

双界面耦合增强效应是指通过在锂金属电池中引入第二个界面,通过这两个界面之间的相互作用来增强电池性能。

具体而言,双界面耦合增强效应可以通过以下几种方式实现:1. 电解液界面优化:在锂金属电池中,电解液是锂离子传输的媒介,也是锂枝晶生成的主要原因之一。

通过优化电解液的配方和添加功能性添加剂,可以抑制锂枝晶的生长,减少电池内部的电化学反应,从而提高电池的循环寿命和安全性能。

2. 电极表面改性:锂金属电池中,锂金属负极表面的不稳定性是限制电池循环寿命的关键问题之一。

通过在锂金属表面引入稳定的锂盐膜、聚合物膜等材料,可以有效地抑制锂金属表面的氧化和枝晶生长,提高电池的循环寿命和安全性能。

3. 电池结构设计:通过合理设计电池的结构,如引入多孔结构、导电添加剂等,可以提高电池的电解液扩散速率和锂离子传输速率,减少电池内部的电阻,提高电池的功率密度和循环寿命。

4. 温度控制:锂金属电池的工作温度对电池性能具有重要影响。

通过合理控制电池的工作温度,可以减缓锂金属表面的氧化反应和锂枝晶的生长,提高电池的循环寿命和安全性能。

双界面耦合增强效应的研究对于提高锂金属电池的性能具有重要意义。

通过合理设计和优化电池结构,可以有效地解决锂金属电池中的问题,提高电池的循环寿命、安全性能和能量密度。

然而,双界面耦合增强效应的机制还需要进一步的研究和探索,以实现锂金属电池的商业化应用。

双界面耦合增强效应是一种通过优化电解液界面、电极表面改性、电池结构设计和温度控制等方式来提高锂金属电池性能的机制。

半导体器件机理 英文

半导体器件机理 英文

半导体器件机理英文Semiconductor Device Mechanisms.Semiconductors are materials that have electrical conductivity between that of a conductor and an insulator. This unique property makes them essential for a wide range of electronic devices, including transistors, diodes, and solar cells.The electrical properties of semiconductors are determined by their electronic band structure. In a semiconductor, the valence band is the highest energy band that is occupied by electrons, while the conduction band is the lowest energy band that is unoccupied. The band gap is the energy difference between the valence band and the conduction band.At room temperature, most semiconductors have a relatively large band gap, which means that there are very few electrons in the conduction band. This makessemiconductors poor conductors of electricity. However, the electrical conductivity of a semiconductor can be increased by doping it with impurities.Donor impurities are atoms that have one more valence electron than the semiconductor atoms they replace. When a donor impurity is added to a semiconductor, the extra electron is donated to the conduction band, increasing the number of charge carriers and the electrical conductivityof the semiconductor.Acceptor impurities are atoms that have one lessvalence electron than the semiconductor atoms they replace. When an acceptor impurity is added to a semiconductor, the missing electron creates a hole in the valence band. Holes are positively charged, and they can move through the semiconductor by accepting electrons from neighboring atoms. This also increases the electrical conductivity of the semiconductor.The type of impurity that is added to a semiconductor determines whether it becomes an n-type semiconductor (witha majority of electrons as charge carriers) or a p-type semiconductor (with a majority of holes as charge carriers).The combination of n-type and p-type semiconductors is used to create a wide range of electronic devices,including transistors, diodes, and solar cells.Transistors.Transistors are three-terminal devices that can be used to amplify or switch electronic signals. The threeterminals are the emitter, the base, and the collector.In a bipolar junction transistor (BJT), the emitter is an n-type semiconductor, the base is a p-type semiconductor, and the collector is another n-type semiconductor. When a small current is applied to the base, it causes a large current to flow between the emitter and the collector. This makes BJTs ideal for use as amplifiers.In a field-effect transistor (FET), the gate is a metal electrode that is insulated from the channel. When avoltage is applied to the gate, it creates an electricfield that attracts or repels electrons in the channel. This changes the conductivity of the channel, which in turn controls the flow of current between the source and the drain. FETs are ideal for use as switches.Diodes.Diodes are two-terminal devices that allow current to flow in only one direction. The two terminals are the anode and the cathode.In a p-n diode, the anode is a p-type semiconductor and the cathode is an n-type semiconductor. When a voltage is applied to the diode, it causes electrons to flow from the n-type semiconductor to the p-type semiconductor, but not vice versa. This makes diodes ideal for use as rectifiers, which convert alternating current (AC) to direct current (DC).Solar Cells.Solar cells are devices that convert light energy into electrical energy. They are made of a semiconductor material, such as silicon, that has a p-n junction.When light strikes the solar cell, it creates electron-hole pairs in the semiconductor. The electrons areattracted to the n-type semiconductor, while the holes are attracted to the p-type semiconductor. This creates a voltage difference between the two semiconductors, which causes current to flow.Solar cells are used to power a wide range of devices, including calculators, watches, and satellites. They are also used to generate electricity for homes and businesses.Conclusion.Semiconductors are essential for a wide range of electronic devices. Their unique electrical properties make them ideal for use in transistors, diodes, and solar cells. As semiconductor technology continues to develop, we canexpect to see even more innovative and efficient electronic devices in the future.。

新型双凸极电机及转矩脉动抑制技术

新型双凸极电机及转矩脉动抑制技术

新型双凸极电机及转矩脉动抑制技术英文回答:New Dual Convex Pole Motor and Torque Ripple Suppression Technology.Introduction:I am excited to discuss the new dual convex pole motor and its torque ripple suppression technology. This innovative motor design offers improved performance and reduced torque ripple, making it suitable for a wide range of applications. In this article, I will explain the principles behind the motor design and discuss the techniques used to suppress torque ripple.Motor Design:The dual convex pole motor is a unique design that features two convex-shaped poles instead of the traditionalflat poles found in most motors. This design allows for a more efficient magnetic field distribution, resulting in improved torque production and reduced cogging torque.The convex shape of the poles enhances the magneticflux density at the air gap, resulting in a higher torque output. Additionally, the unique pole shape helps to minimize torque ripple by reducing the variation in magnetic flux density during motor operation.Torque Ripple Suppression:Torque ripple refers to the variation in torque output during motor operation. It is an undesirable characteristic that can cause vibration, noise, and reduced motor efficiency. The new dual convex pole motor incorporates advanced torque ripple suppression technology to mitigate these issues.One technique used to suppress torque ripple is the implementation of a feedback control system. By continuously monitoring the motor's performance andadjusting the current waveform, the control system can effectively minimize torque ripple. This technique is particularly useful in applications that require precise torque control, such as robotics or electric vehicles.Another technique used to reduce torque ripple is the use of advanced motor control algorithms. These algorithms take into account the motor's characteristics and dynamically adjust the current waveform to minimize torque ripple. For example, the control algorithm may modify the current waveform during specific motor operating conditions to achieve smoother torque output.Example:To illustrate the benefits of the new dual convex pole motor and torque ripple suppression technology, let's consider an electric vehicle application. In this scenario, precise torque control is crucial for smooth acceleration and deceleration.With the traditional motor design, the electric vehiclemay experience noticeable torque ripple during acceleration, resulting in a jerky ride experience for the driver and passengers. However, with the new dual convex pole motorand torque ripple suppression technology, the torque output can be significantly smoother, providing a more comfortable and seamless acceleration.中文回答:新型双凸极电机及转矩脉动抑制技术。

高频下分流器集肤效虚及邻近效应的有限元分析

高频下分流器集肤效虚及邻近效应的有限元分析

高频下分流器集肤效虚及邻近效应的有限元分析【摘要】本文建立了分流器的二维有限元模型,分析了分流器电阻以及电感随频率变化的关系。

计算结果表明:在高频条件下,集肤效应和邻近效应是增大交流电阻和减小电感的主要原因。

【关键词】分流器集肤效应临近效应分流器是电路中用来测量电流的一种常用器件。

在高频下,其交流电阻和电感对能否准确测量电流至关重要。

因此,准确计算分流器的交流电阻以及电感就显得非常有实际意义。

对分流器电阻与电感产生影响的因素包括温度、集肤效应和邻近效应等,其中集肤效应和邻近效应是主要因素。

为探讨集肤效应和邻近效应对电阻与电感的影响,本文建立一个二维有限元模型,很方便地得到了交流电阻和电感随频率的变化关系,避免采用Bessel函数直接计算法在高频下得不到可靠结果的缺点。

一、模型假设(一)略端部效应的影响由于分流器的长度比分流器的半径大得多,为了简化模型,同时减小有限元模型与计算量,暂且忽略端部效应的影响,从而可以把三维模型简化为二维模型。

同时考虑到分流器的对称性,只需要建立分流器截面的1/6即可。

(二)简化为准静态模型在本文中,不考虑位移电流以及波动效应,认为该问题是一个准静态过程。

(三)不考虑热效应由于该模型是一个准静态模型,而分流器是铜锰材料制成,具有很好的散热能力,可认为导体温度稳定在室温,即293K。

二、模型建立描述集肤效应和临近效应的电磁场方程如下式所示:其中a为铜锰导体电导率,W为电流角频率,A为向量磁位,A V为导体两端的电势差,/为导体棒长度。

本文中,用到的常数有z\V=lKV,盯=l/(-1.3×10J+9.6×10-1l×293),为铜在293K下的电导率。

(一)分流器结构:分流器是由90根长为240mm,直径为6mm的铜锰导体棒分两排圆形排列而成,如图1所示:(a)单根示意图(b)整体示意图图1分流器结构图(二)单根铜锰棒分析集肤效应是由场量在导体内部的衰减形成的,而场量在导体内的衰减快慢又可用导体的透入深度8表示,其一般电流密度.,随着深度6的增加按指数规律衰减:,其中K为导体表面的电流密度,d为从导体表面算起的深度。

半导体一些术语的中英文对照

半导体一些术语的中英文对照

半导体一些术语的中英文对照离子注入机ion implanterLSS 理论Lindhand Scharff and Schiott theory 又称“林汉德-斯卡夫-斯高特理论”。

沟道效应channeling effect 射程分布range distribution 深度分布depth distribution 投影射程projected range 阻止距离stopping distance 阻止本领stopping power 标准阻止截面standard stopping cross section 退火annealing 激活能activation energy 等温退火isothermal annealing 激光退火laser annealing 应力感生缺陷stress-induced defect 择优取向preferred orientation 制版工艺mask-making technology 图形畸变pattern distortion 初缩first minification 精缩final minification 母版master mask 铭版chromium plate 干版dry plate 孚L胶版emulsion plate 透明版see-through plate高分辨率版high resolution plate, HRP超微粒干版plate for ultra-microminiaturization 掩模mask掩模对准mask alignment 对准精度alignment precision 光刻胶photoresist又称“光致抗蚀剂”。

负性光刻胶negative photoresist 正性光刻胶positive photoresist 无机光刻胶inorganic resist 多层光刻胶multilevel resist 电子束光刻胶electron beam resist X射线光刻胶X-ray resist 刷洗scrubbing 甩胶spinning 涂胶photoresist coating 后烘postbaking 光刻photolithographyX 射线光刻X-ray lithography电子束光刻U electron beam lithography离子束光刻ion beam lithography深紫外光刻deep-UV lithography光刻机mask aligner投影光刻机projection mask aligner曝光exposure接触式曝光法contact exposure method接近式曝光法proximity exposure method光学投影曝光法optical projection exposure method 电子束曝光系统electron beam exposure system 分步重复系统step-and-repeat system显影development线宽linewidth去胶stripping of photoresist氧化去胶removing of photoresist by oxidation等离子[体]去胶removing of photoresist by plasma亥U蚀etching干法刻蚀dry etching反应离子刻蚀reactive ion etching, RIE各向同性刻蚀isotropic etching各向异性刻蚀anisotropic etching反应溅射刻蚀reactive sputter etching离子铳ion beam milling又称“离子磨削”。

Multi-mode synchronous memory device and methods o

Multi-mode synchronous memory device and methods o

专利名称:Multi-mode synchronous memory deviceand methods of operating and testing same发明人:Brian Johnson,Brent Keeth,Jeffrey W.Janzen,Troy A. Manning,Chris G. Martin申请号:US11001231申请日:20041201公开号:US07057967B2公开日:20060606专利内容由知识产权出版社提供专利附图:摘要:A synchronous semiconductor memory device is operable in a normal mode and an alternative mode. The semiconductor device has a command bus for receiving aplurality of synchronously captured input signals, and a plurality of asynchronous input terminals for receiving a plurality of asynchronous input signals. The device further has a clock input for receiving an external clock signal thereon, with the device being specified by the manufacturer to be operated in the normal mode using an external clock signal having a frequency no less than a predetermined minimum frequency. An internal delay locked loop (DLL) clocking circuit is coupled to the clock input terminal and is responsive in normal operating mode to be responsive to the external clock signal to generate at least one internal clock signal. control circuitry in the device is responsive to a predetermined sequence of asynchronous signals applied to the device's asynchronous input terminals to place the device in an alternative mode of operation in which the internal clocking circuit is disabled, such that the device may be operated in the alternative mode using an external clock signal having a frequency less than the predetermined minimum frequency. The alternative mode of operation facilitates testing of the device at a speed less than the minimum frequency specified for the normal mode of operation.申请人:Brian Johnson,Brent Keeth,Jeffrey W. Janzen,Troy A. Manning,Chris G. Martin 地址:5977 E. Gateway Dr. Boise ID 83716 US,5077 N. Fifeshire Pl. Boise ID 83713 US,8000 S. Federal Way MS 607 Meridian ID 83707-0066 US,8153 S. Obudah La. Meridian ID 83642 US,3669 E. Alta Ridge Ct. Boise ID 83716 US国籍:US,US,US,US,US代理机构:Wong, Cabello, Lutsch, Rutherford, & Brucculeri更多信息请下载全文后查看。

锂离子电池多场耦合

锂离子电池多场耦合

锂离子电池多场耦合英文回答:Multi-Field Coupling in Lithium-Ion Batteries.Lithium-ion batteries (LIBs) are electrochemical energy storage devices that play a crucial role in powering portable electronics, electric vehicles, and grid storage systems. The performance and lifespan of LIBs are governed by complex physical and chemical processes that occur at multiple length and time scales. Multi-field coupling models provide a comprehensive framework to capture these complex interactions and predict the behavior of LIBs under various operating conditions.Multi-field coupling in LIBs involves the simultaneous consideration of multiple physical fields, including:Electrochemical field: Captures the electrochemical reactions, charge transport, and potential distributionwithin the battery.Thermal field: Models heat generation, transfer, and dissipation due to electrochemical reactions and external factors.Mechanical field: Accounts for mechanical stresses, strains, and deformations caused by volume changes, thermal expansion, and external loads.Transport field: Describes the transport of ions, electrons, and heat through the battery components.The coupling between these fields creates intricate feedback loops that significantly influence the performance and durability of LIBs. For instance, electrochemical reactions generate heat, which affects the battery's temperature distribution and can lead to thermal runaway. Conversely, temperature variations can alter the electrochemical kinetics and transport properties of the battery materials. Mechanical stresses and deformations can impact the electrical and thermal contact between batterycomponents, affecting the overall performance.Multi-field coupling models are essential for understanding and predicting the behavior of LIBs under various operating conditions, including charging, discharging, cycling, and extreme environments. These models are used to optimize battery design, improve performance, and ensure safety by:Predicting the temperature distribution and mitigating thermal runaway risks.Estimating mechanical stresses and deformations to prevent structural failure.Optimizing charge and discharge rates to maximize battery life and performance.Investigating the effects of aging, cycling, and external factors on battery health.Advanced numerical simulation techniques, such asfinite element analysis (FEA) and computational fluid dynamics (CFD), are employed to solve the multi-field coupling equations. These simulations provide detailed insights into the internal behavior of LIBs, enabling researchers and engineers to make informed decisions regarding battery design and operation.中文回答:锂离子电池多场耦合。

SWITCHING BETWEEN MULTIPLE COUPLING MODES

SWITCHING BETWEEN MULTIPLE COUPLING MODES

专利名称:SWITCHING BETWEEN MULTIPLE COUPLING MODES发明人:KLABUNDE, Karin,CORROY, Steven,BALDUS, Heribert,GALLO, Francesco申请号:IB2008055385申请日:20081217公开号:WO09/081337P1公开日:20090702专利内容由知识产权出版社提供摘要:A device for transmitting in multiple coupling modes has a transmission module (11), at least one periphery module (18), and an antenna (16, 17) for each of the multiple coupling modes. Further, in a method of switching between multiple coupling modes, switching is conducted between at least polling and listening phases of first and second coupling modes. The device and the method enable a seamless switching between e.g. a near field communication and a body coupled communication. Such a coupling or switching is particularly useful for performing secure transactions whereby through body coupled communication a body-worn tag is interrogated which provides a secure code for a transaction initiated through near field communication with a transaction terminal.申请人:KLABUNDE, Karin,CORROY, Steven,BALDUS, Heribert,GALLO, Francesco地址:Steindamn 94 20099 Hamburg DE,Groenewoudseweg 1 NL-5621 BA Eindhoven NL,c/o PHILIPS IP & S - NL High Tech Campus 44 NL-5656AE Eindhoven NL,c/o PHILIPS IP & S - NL High Tech Campus 44 NL-5656AE Eindhoven NL,c/o PHILIPS IP & S - NL High Tech Campus 44 NL-5656AE Eindhoven NL,c/o PHILIPS IP & S - NL High Tech Campus 44 NL-5656AE Eindhoven NL国籍:DE,NL,NL,NL,NL,NL代理机构:VAN VELZEN, Maaike 更多信息请下载全文后查看。

基于改进型偏差耦合结构的多电机同步控制

基于改进型偏差耦合结构的多电机同步控制

基于改进型偏差耦合结构的多电机同步控制作者:彭晓燕刘威张强来源:《湖南大学学报·自然科学版》2013年第11期摘要:针对现有的多电机同步控制方案难以满足高精度控制的要求和不能实现比例同步控制的局限性,提出一种带PI补偿控制的改进型偏差耦合控制结构,可适用于多电机完全同步和比例同步控制.针对永磁同步电动机非线性和强耦合特性,设计了自适应模糊滑模变结构控制器来实现永磁同步电动机的跟踪控制.建立了4台永磁同步电动机的同步控制仿真模型,仿真实验表明,所提出的多电机同步控制结构相对于带固定增益补偿的控制结构具有更高的同步控制精度.与PID和常规滑模控制算法相比,自适应模糊滑模控制策略具有更高的同步稳定性和更强的鲁棒性.关键词:多电机同步控制;偏差耦合控制;PI补偿器;滑模控制中图分类号:TH165.3 文献标识码:A目前,多电机同步控制技术已广泛应用于造纸、印染、纺织以及机械加工等工业制造领域.多电机同步控制结构和控制算法不仅是实现高精度同步控制的关键,同时也直接影响着系统的可靠性和产品制造的质量.同步控制根据控制参数不同可分为速度同步、位置同步和相位同步等;按照控制参数数值不同可分为完全同步控制和比例同步控制,最常见的应用是速度完全同步控制,如大型龙门吊车,需要控制两台电机以相同的速度驱动负载.而在有些场合,需要使各电机的速度保持一定的比例关系来驱动负载,如超高速卷接机,为了保证各鼓轮在单位时间内烟支的传输量相等,各交接处鼓轮的线速度应相等,因此要求各鼓轮的转速严格保持一定的比例关系.本文重点研究具有速度完全同步和比例同步要求的多电机同步控制结构和控制策略.常用的多电机同步控制结构有主从式、交叉耦合式和偏差耦合式等.相邻耦合[1]控制结构仅考虑相邻两电机的状态,当某台电机受到扰动产生速度波动时,只能通过其相邻电机逐个传递给其他电机,这将导致一定的控制时延从而造成较大的同步误差,因此在实际工程应用中受到了很大的限制.通过对交叉耦合控制结构的改进,PerezPinal等[2]提出了适用于多电机的偏差耦合控制结构,其控制效果有了质的变化,克服了其他控制策略的缺点,具有很好的同步性能.但是如何根据各个电机拖动子系统的负载变化以及干扰等因素实时调节速度补偿值是实现偏差耦合控制的关键.文献[3-5] 采用固定增益速度补偿器实现偏差耦合控制,该算法虽结构简单,但因其补偿器只考虑了电机的转动惯量对同步性能的影响,当负载变化大时,系统波动较大,甚至会导致系统不稳定.为此,本文提出基于PI补偿器的偏差耦合控制结构可综合考虑电机参数、负载波动以及扰动等因素造成的速度偏差,实时修正速度补偿值,从而获得更高的同步控制精度.永磁同步电动机(PMSM)具有动态响应快、效率高、可靠性高等优点,非常适合应用于高性能伺服系统,在多电机同步控制系统中也被广泛采用,但是PMSM伺服系统是一个多变量、强耦合、非线性时变系统,一般的控制算法难以达到令人满意的效果.文献[6]采用模糊PID控制算法来实现多电机系统的同步控制,虽对多输入多输出、时变及滞后等复杂系统都能进行控制,但模糊控制规则太多且过于依赖操作者的经验,参数整定困难,无自学习能力,其应用范围受到较大的局限.文献[7]采用神经网络控制算法实现多电机的同步控制,神经网络控制虽具有自学习和自适应能力,但计算复杂且实效性较差.针对不确定非线性控制系统,滑模变结构控制是一种较为有效的跟踪控制方法[8],它具有响应速度快、控制精度高、鲁棒性强、算法简单、易于在线实现等许多优点.本文从实际工程应用角度出发,结合滑模变结构控制和模糊控制方法,设计了基于切换增益自适应调节的模糊滑模控制器,并采用带PI补偿控制的偏差耦合控制结构,对4台永磁同步电动机进行同步控制,仿真结果证明了该控制结构和控制策略的可行性和有效性.1改进型偏差耦合控制结构设计偏差耦合控制是利用各个电机系统之间的阻尼系数关系在速度反馈信号中添加各电机的相对速度信号,根据每台电机的工作状态动态地在各电机之间分配速度补偿信号,从而达到很好的同步性能.传统的偏差耦合控制结构只能实现多电机的完全同步控制,而不能实现多电机的比例同步控制.针对这一问题,本文对传统的偏差耦合控制结构做了改进,以实现多电机的比例同步控制,以3台电机为例,改进型的偏差耦合控制结构示意图如图1所示.固定增益速度补偿器只考虑了电机的转动惯量,当负载变化较大时,因电机特性参数和机电时间常速的不同,导致速度波动大,且消除速度波动的时间长,这也造成多电机间速度不同步,即出现同步误差.对于每一台电机而言,其他任意一台电机速度的波动都是一种干扰,而这种干扰是可测且经常变化的,故可通过引入前馈作用实时消除干扰对各电机驱动子系统输出的不良影响,很好地提高系统的控制品质.本文采用PI控制器代替固定增益,实现电机的前馈控制,其结构如图3所示.干扰一旦出现,在被控制量发生变化前,PI调节器就产生控制作用,即直接根据检测到的其他电机的速度按一定规律快速消除电机之间的跟随误差,使其稳定收敛于零,从而保证系统具有优良的同步性能的同时,使系统获得更好的动态和静态性能.2模糊滑模控制器设计由于永磁同步电动机是一个多变量、强耦合和非线性的复杂系统,一般的控制方法难以达到令人满意的调速性能,而滑模控制(SMC)能够克服复杂系统的不确定性,对干扰和未建模动态有很强的鲁棒性,非常适合作为一种不确定系统的鲁棒控制器.但是普通的滑模控制会存在抖振现象,所以必须采取降抖振措拖才能应用于实际.本文将模糊控制和变结构控制相结合,设计模糊滑模控制器,采用模糊控制自适应调节引起滑模抖振的增益系数以削弱系统抖振,用滑模控制确保系统的稳定性,它不仅保持了模糊控制不依赖系统模型的优点,同时还可在保证控制精度的前提下减小滑模控制的抖振,实现对永磁同步电机速度的鲁棒控制.2.1基于比例切换函数的SMC控制器设计文献[9]提出了基于比例切换函数的滑模控制方法,对永磁同步电动机实现了较好的速度控制.本文将该方法应用到多永磁同步电动机的速度同步控制中,并将其与所设计的模糊滑模控制方法进行仿真对比.3.4仿真实验结果分析1)由图5可知,2种补偿器均能实现多电机的完全同步控制,而与固定增益补偿器相比,PI补偿控制下4台电机能够更快地到达稳定状态.当4台电机发生负载突变时,PI补偿控制使系统能重新快速准确地跟踪目标,速度波动小,调节时间短,消除系统误差速度快.由表2可以看出,PI补偿控制下4台电机之间的最大同步误差率更小,到达稳态时同步精度更高,体现了较好的抗干扰性.2)由图6可知,2种补偿器均能实现多电机的比例同步控制,而与固定增益补偿器相比,PI补偿控制下,4台电机在初始时刻能够快速、稳定地到达稳定状态,当转速下降时,同样能够快速稳定地到达稳态,当发生负载突变时,系统能够快速消除误差,体现了较好的动态性能.由表2可以看出,PI补偿控制下,无论是在降速过程中还是在负载发生突变情况下,4台电机之间的最大同步误差率更小,到达稳态时同步精度更高,体现了较好的鲁棒性.3)由图7可知,与传统的PID控制和文献[9]提出的基于比例切换函数的滑模控制(SMC)相比,无论是电机1还是电机2,FuzzySMC控制下系统响应更快,速度波动更小.由表3可以看出,在FuzzySMC控制下,系统的同步精度更高,鲁棒性更强.4结论本文从多电机同步控制结构和控制策略2个方面进行了仿真实验研究,主要结论如下:1)对传统的偏差耦合控制结构进行了改进,实现了多电机的比例同步控制,为多电机的比例同步控制提供了一种参考模型.2)提出了一种带PI补偿的速度补偿器,与传统的固定增益速度补偿器相比,它能使系统更快地到达稳态,当负载突变时,系统消除速度波动的时间更短,控制精度更高,性能更优越.3)从实际工程应用角度出发,设计了一种自适应模糊滑模控制器,与PID控制和普通滑模控制算法相比,该控制器能使系统的动态性能和静态性能更好,为多电机的同步控制提供了一种简单实用的控制策略.综上所述,本文设计的改进型偏差耦合控制结构和自适应模糊滑模控制器均为有效的、可靠的多电机同步控制方案,具有较高的实际应用价值.参考文献[1]SHIH Y T, CHEN C S, LEE A C. A novel crosscoupling control design for biaxis motion [J].International Journal of Machine Tool & Manufacture, 2002, 42(14): 1539-1548.[2]PEREZPINAL F J,NUNEZ C,ALVAREZ passion of Multimotor synchronization techniques[C]//The 30th Annual Conference of me IEEE Industrial Electronics Society,Busan,Korea,2004,10:2-6.[3]苗新刚,汪苏,韩凌攀,等.基于偏差耦合的多电机单神经元同步控制[J].微电机,2011,44(2):44-47.[4]曹玲芝,王红卫,李春文,等.基于偏差耦合的起重机起升机构同步控制[J].计算机工程与应用,2008,44(25):233-235.[5]PEREZPINAL F J,CALDERON G.Relative coupling strategy[J].IEEE,2003,2(6):1162-1166.[6]许宏,李乐宝,张怡,等.变摩擦负载下双电机同步控制系统设计与实验[J].中国机械工程,2011,22(24):2908-2913.[7]戴先中,刘国海.两变频调速电机系统的神经网络逆同步控制[J].自动化学报,2005,31(6):890-900.[8]刘金琨.滑模变结构控制MATLAB仿真[M]. 北京:清华大学出版社,2005:10-14.[9]方斯琛,周波,黄佳佳,等.滑模控制永磁同步电动机调速系统[J].电工技术学报,2008,23(8):29-35.[10]高为炳.变结构控制的理论及设计方法[M].北京:科学出版社,1996.。

新型双层半插入式交替极永磁电机的解析建模

新型双层半插入式交替极永磁电机的解析建模

第27卷㊀第9期2023年9月㊀电㊀机㊀与㊀控㊀制㊀学㊀报Electri c ㊀Machines ㊀and ㊀Control㊀Vol.27No.9Sep.2023㊀㊀㊀㊀㊀㊀新型双层半插入式交替极永磁电机的解析建模倪有源,㊀张亮,㊀钱威(合肥工业大学电气与自动化工程学院,安徽合肥230009)摘㊀要:为了提高永磁电机的电磁转矩,提出一种新型双层半插入式交替极永磁电机模型㊂内层为交替极磁极,外层为表贴式永磁体,内外层永磁均为径向磁化㊂采用二维子域模型法,将求解域划分为定子槽㊁定子槽口㊁气隙㊁外层永磁和内层永磁共5个子域㊂根据拉普拉斯和泊松方程,通过分离变量法获得各子域矢量磁位的通解,再根据边界条件并求解矩阵方程,获得各子域矢量磁位的直流分量和谐波系数㊂以一台8极9槽新型电机为例,计算了电机的气隙磁密㊁空载感应电动势和电磁转矩,结果表明,在永磁用量相同的前提下,与传统的表面插入式电机相比,新型电机具有更高的电磁转矩㊂最后,用有限元法验证了解析模型的正确性㊂关键词:双层半插入交替极永磁电机;子域模型;矢量磁位;解析模型;有限元法;电磁转矩DOI :10.15938/j.emc.2023.09.005中图分类号:TM351文献标志码:A文章编号:1007-449X(2023)09-0042-11㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀收稿日期:2022-08-04基金项目:安徽省自然科学基金(2008085ME179);高等学校学科创新引智计划资助(BP0719039)作者简介:倪有源(1976 ),男,博士,副教授,研究方向为特种电机设计及控制;张㊀亮(1996 ),男,硕士,研究方向为永磁电机设计;钱㊀威(1997 ),男,硕士,研究方向为永磁电机设计㊂通信作者:倪有源Analytical modeling of novel double-layer semi-inset consequent-polepermanent magnet machineNI Youyuan,㊀ZHANG Liang,㊀QIAN Wei(School of Electrical Engineering &Automation,Hefei University of Technology,Hefei 230009,China)Abstract :In order to improve the electromagnetic torque of permanent magnet machines,a novel double-layer semi-inset consequent-pole permanent magnet machine model was proposed.On the basis of an in-ner layer with consequent-pole magnets,an outer layer with surface-mounted magnets was added,and all the magnets were radially ing a two-dimensional accurate subdomain model method,the solution domains were divided into the stator slot,the stator slot,the air-gap,the outer layer magnet and the inner layer magnet subdomains.According to the Laplace /Poisson equations,the general solution of the vector magnetic potential in each subdomain was derived by a variable separation method.According to the boundary and interface conditions,the matrix equation was solved and the DC component and har-monic coefficients of the vector magnetic potential in each subdomain were obtained.An 8-pole /9-slot machine was investigated,the air-gap flux density,the no-load back EMF and the electromagnetic torque were calculated.The results show that with the same amount of magnets,the novel double-layer semi-insetconsequent-pole permanent magnet machine has a higher electromagnetic torque than the traditional sur-face-inset machine.Finally,correctness of the analytical model was verified by a finite element method.Keywords :double-layer semi-inset consequent-pole permanent magnet machine;subdomain model;vec-tor magnetic potential;analytical model;finite element method;electromagnetic torque0㊀引㊀言无刷直流/交流永磁电机因结构简单㊁功率密度高及效率高等优点,在各个领域应用十分广泛[1-2]㊂与传统的表面插入式永磁电机相比,交替极永磁电机在保证较高功率密度的同时,可以节省永磁材料,降低制造成本,具有广泛的应用前景[3]㊂国内外学者对交替极永磁电机进行了广泛的研究㊂文献[4]建立了外转子交替极电机的通用解析模型,并计算了其电磁性能㊂文献[5]分析了交替极磁通反向永磁电机,与传统电机相比,该电机可减少漏磁,从而提高了感应电动势和电磁转矩㊂文献[6]分析比较了传统内置式永磁同步电机与内置式交替极永磁同步电机的电磁性能㊂文献[7]建立了开口槽交替极电机空载磁场㊁负载磁场和电枢反应磁场的解析模型,由于不对称性,气隙磁场中的偶次谐波含量较高㊂由于交替极电机具有非对称气隙磁场,文献[8]研究了不同极槽配合下偶次谐波对空载感应电动势㊁齿槽转矩和电磁转矩波动的影响㊂对永磁电机的精确建模是获得其电磁性能及进行优化设计的前提㊂文献[9-10]采用等效磁路法分别研究了内置式永磁电机的电感参数和磁化特性㊂文献[11-12]采用等效磁网络法对直线永磁电机进行了建模分析㊂文献[13-14]基于数值法研究了不同转子磁路结构内置式永磁电机的转矩特性㊂文献[15]采用保角变换法建立了开槽永磁电机的解析模型,计算了气隙比磁导㊂相比以上方法,子域模型法不仅计算精度高,而且计算时间少,非常适合分析电机电磁场㊂文献[16-18]建立了表贴式开口槽电机空载磁场解析模型,计算了气隙磁密㊁反电动势和电磁转矩,但未考虑实际电机定子齿尖的影响㊂半开口槽电机比开口槽电机多了槽口子域,使得方程的求解变得更为复杂㊂文献[19]采用子域模型法研究了表贴式开口槽永磁电机的电磁性能,并与有限元的结果进行了对比㊂文献[20-21]建立了表面插入式半开口槽电机的二维子域模型,可以计算任意极槽比㊁不同磁化方式的永磁电机的磁场分布㊂文献[22]基于子域模型法研究了一种表面插入式半开口槽电机的磁场分布和运行参数㊂为降低电磁转矩脉动和永磁体涡流损耗,文献[23]提出一种表贴式分段Halbach永磁结构,利用二维解析建模,获得了两种不同磁极结构的电磁性能㊂由于交替极永磁电机具有不对称性,导致气隙磁场和感应电动势的波形中谐波含量较高㊂与表贴式永磁电机相比,交替极永磁电机的电磁转矩脉动较大㊂因此,本文在交替极电机的基础上,提出了一种新型双层半插入式交替极永磁结构,该结构的内层永磁和外层永磁分别为交替极和表贴式,削弱了交替极永磁电机不对称性对电磁性能的影响㊂在电机的分析方法中,子域模型法不仅计算精度高㊁计算速度快,而且每个变量的物理意义明确,非常适合对该电机进行解析建模㊂本文首先给出双层半插入式交替极永磁电机的结构㊂然后采用二维子域模型法对其进行建模,利用微分方程和边界条件建立矩阵方程,并求解获得磁场分布以及电磁性能㊂计算结果表明,与传统的表面插入式电机相比,新型电机具有更高的电磁转矩㊂最后,利用有限元仿真结果验证解析模型的有效性㊂1㊀新型电机结构传统表插式转子结构如图1(a)所示㊂新型双层半插入式交替极电机的转子结构如图1(b)所示㊂采用双层磁极结构,内层为交替极磁极,外层为表贴式磁极,内外层永磁都采用径向充磁㊂图1㊀两种电机转子结构图Fig.1㊀Rotor structures of two machines新型电机的定转子结构图如图2所示㊂为方便求解,新型双层半插入式交替极电机定子槽为理想的带齿尖的扇形结构㊂图中:ξ1为扇形定子槽对应圆心角的弧度;ξ2为扇形定子槽口对应圆心角的弧度;ξ3为转子槽对应圆心角的弧度㊂为便于计算,将图中的定转子位置设为初始位置㊂此时转子槽中心线与扇形槽中心线重合,定义该转子槽和定子槽为第1个转子槽和第1个定子槽㊂转子槽序号以逆时针为正方向,依次为第2个转子槽, ,第s个转子槽;定子槽序号以逆时针为正方向,依次为第2个定子槽, ,第i个定子槽;R1为内层磁极内半径,R2为内层磁极外半径,R3为外层磁极外半径,R4为定子内半径,R5为定子齿尖外半径,R6为定子槽外半径;θi为第i个定子槽中心线与第1个定子槽中心线之间的夹角;θs为第s个转子槽中心线与第1个转子槽34第9期倪有源等:新型双层半插入式交替极永磁电机的解析建模中心线之间的夹角㊂θi 和θs 的表达式分别为:θi =2πN s(i -1),i =1,2,3, ,N s ;θs =2πN r(s -1)+θ0,s =1,2,3, ,N r ㊂üþýïïïï(1)式中:N r 和N s 分别为电机转子槽和定子槽的数量;θ0为转子相对于初始位置的位置角㊂图2㊀新型电机定转子结构图Fig.2㊀Stator and rotor of proposed novel machine2㊀解析建模分析以新型双层半插入式交替极永磁电机为对象,基于子域模型法进行磁场分析,需要如下假设:a)定子转子的铁心磁导率为无穷大;b)永磁体的相对磁导率μr =1.05;c)只考虑二维模型;d)定子槽和定子槽口均为理想的径向扇形㊂将新型电机的求解域分为:定子槽㊁定子槽口㊁气隙㊁外层永磁和内层永磁共5个子域㊂采用子域模型法,需对各子域矢量磁位A 的z 分量建立偏微分方程㊂在二维极坐标系中,磁密的径向分量㊁切向分量与矢量磁位的关系可以表示为:B ρ=1ρ A (ρ,θ) θ;B θ=- A (ρ,θ) ρ㊂üþýïïïï(2)文献[7]定义了一类方程,该方程对偏微分方程通解的表达式进行了谐波分量系数的缩放,从而能够化简运算㊂该方程定义为:P (x ,y ,z )=(y z )x +(zy)x ;E (x ,y ,z )=(y z )x -(z y )x ㊂üþýïïïï(3)接下来对新型电机各子域的方程进行求解㊂2.1㊀槽子域磁场分析对于有N s 个定子槽的新型电机,共有N s 个槽子域㊂设二维极坐标下i A 1(ρ,θ)为第i 个槽内矢量磁位A 的z 分量,其中i =1,2,3, ,N s ㊂当线圈中电流为0时,第i 个槽子域内拉普拉斯方程的通解为iA 1(ρ,θ)=iW 1+iX 1ln ρ+ðɕm =1,2,3,(i m Yρτm+i m Z 1ρ-τm )ˑcos[τm (θ+ξ12-θi )]㊂(4)可以对系数i m Y 1和i m Z 1进行缩放,可得iA 1(ρ,θ)=i W 1+i X 1ln ρ+ðɕm =1,2,3,im Y 1(ρR 6)τmcos[τm (θ+ξ12-θi )]+ðɕm =1,2,3,im Z 1(ρR 5)-τmcos[τm (θ+ξ12-θi )]㊂(5)式中:i W 1㊁i X 1㊁i m Y 1和i m Z 1分别为定子槽域矢量磁位的直流分量系数和谐波分量系数;τm 的表达式为τm =m πξ1㊂(6)式中m 为定子槽域矢量磁位的谐波阶数㊂2.2㊀槽口子域磁场分析由新型电机的结构图可得,对于有N s 个槽的新型电机,其槽口子域共有N s 个㊂设二维极坐标下jA 2(ρ,θ)为第j 个槽口子域内的矢量磁位A 的z 分量,其中j =1,2,3, ,N s ,那么在线圈中电流为0时,对第j 个定子槽口子域内拉普拉斯方程的通解使用分离变量法求解,同时对各系数进行缩放,可得槽口子域矢量磁位的通解为jA 2(ρ,θ)=j W 2+j X 2ln ρ+ðɕn =1,2,3,jn Y 2E (τn ,ρ,R 5)E (τn ,R 4,R 5)cos[τn (θ+ξ22-θj )]+ðɕn =1,2,3,jn Z 2E (τn ,ρ,R 4)E (τn,R 4,R 5)cos[τn (θ+ξ22-θj )]㊂(7)式中:j W 2㊁j X 2㊁j n Y 2和j n Z 2分别为定子槽口域矢量磁位的直流分量系数和谐波分量系数;n 为定子槽口域矢量磁位的谐波阶数;θj 是第j 个槽口中心线所处位置;θj 和τn 分别为:θj =2πN s(j -1),j =1,2,3, ,N s ;(8)τn =n πξ2㊂(9)44电㊀机㊀与㊀控㊀制㊀学㊀报㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀第27卷㊀2.3㊀气隙子域磁场分析由图2可得,该求解域是位于R3与R4之间的环形区域㊂设二维极坐标下A3(ρ,θ)为气隙子域中矢量磁位A的z分量,通过求解该子域内的拉普拉斯方程,可得其通解为A3(ρ,θ)=a3+b3lnρ+ðɕk=1,2,3, (k W3ρk+k X3ρ-k)cos(kθ)+ðɕk=1,2,3, (k Y3ρk+k Z3ρ-k)sin(kθ)㊂(10)与槽子域和槽口子域不同,气隙子域为一连通的环形区域㊂根据安培环路定律可得,该通解不含有直流分量㊂于是将式(10)进行化简,其解的最终形式为A3(ρ,θ)=ðɕk=1,2,3, [k W3R3k P(k,ρ,R4)E(k,R3,R4)]cos(kθ)+ðɕk=1,2,3, [k X3R4k P(k,ρ,R3)E(k,R4,R3)]cos(kθ)+ðɕk=1,2,3, [k Y3R3k P(k,ρ,R4)E(k,R3,R4)]sin(kθ)+ðɕk=1,2,3, [k Y3R3k P(k,ρ,R4)E(k,R3,R4)]sin(kθ)㊂(11)式中:k W3㊁k X3㊁k Y3和k Z3分别为气隙子域矢量磁位的直流分量系数和谐波分量系数;k为气隙域矢量磁位的谐波阶数㊂2.4㊀外层永磁子域磁场分析由图2可得,该求解域是位于R2与R3之间的外层磁极区域㊂设二维极坐标下A4(ρ,θ)为外层永磁子域矢量磁位的z分量,其偏微分方程和域范围为:2A4(ρ,θ)ρ2+1ρA4(ρ,θ)ρ+1ρ22A4(ρ,θ)θ= -μ0ρ(Mθ- Mρ θ);R2ɤρɤR3,0ɤθɤ2π㊂üþýïïïïïï(12)式中Mρ和Mθ分别为永磁体磁化强度的径向分量与切向分量㊂对于径向磁化的永磁体,其磁化强度的径向分量与切向分量为:Mρ=ðɕu=4,12,20, Mρu cos[u(θ-θ0)]; Mθ=ðɕu=4,12,20, Mθu sin[u(θ-θ0)]㊂üþýïïï(13)式中:Mρu=4pB ruπμ0sin(uπ2p);Mθu=0㊂}(14)利用非齐次偏微分方程解的性质,由安培环路定律,采用分离变量法,可得其解为A4(ρ,θ)=ðɕu=1,2,3, (u W4ρu+u X4ρ-u)cos(uθ)+ðɕu=1,2,3, (u Y4ρu+u Z4ρ-u)sin(uθ)+ðɕu=1,2,3, [F u(ρ)sin(uθ0)]cos(uθ)+ðɕu=1,2,3, [F u(ρ)sin(uθ0)]sin(uθ)㊂(15)对该解进行化简,可得其解的最终形式为㊀A4(ρ,θ)=ðɕu=1,2,3, [u W4R2u P(u,ρ,R3)E(u,R2,R3)+u X4R3uP(u,ρ,R2)E(u,R3,R2)]cos(uθ)+ðɕu=1,2,3, [u Y4R2u P(u,ρ,R3)E(u,R2,R3)+u Z4R3uP(u,ρ,R2)E(u,R3,R2)]sin(uθ)+ðɕu=1,2,3, [F u(ρ)sin(uθ0)]cos(uθ)+ðɕu=1,2,3, [F u(ρ)cos(uθ0)]sin(uθ)㊂(16)式中:u W4㊁u X4㊁u Y4和u Z4分别为外层磁极子域矢量磁位的直流分量系数和谐波分量系数;u为外层磁极子域矢量磁位的谐波阶数;函数F u(ρ)为:F u(ρ)=μ0ρuMρu+Mθuu2-1,u/p=1,2,3, ;-μ0ρln(ρ)Mρu+Mθu2,up=1;0,其他㊂ìîíïïïïïï(17)2.5㊀内层永磁子域磁场分析由新型电机的结构图可得,对于有N r个转子槽的新型电机,其内层永磁子域共有N r个㊂设二维极坐标下s A5(ρ,θ)为第s个内层永磁子域内的矢量磁位的z分量,其中s=1,2,3, ,N r㊂当线圈电流为0时,在该子域内s A5(ρ,θ)的偏微分方程和域范围为:54第9期倪有源等:新型双层半插入式交替极永磁电机的解析建模2[s A 5(ρ,θ)] ρ2+1ρ [sA 5(ρ,θ)]ρ+㊀㊀1ρ22[s A 5(ρ,θ)] θ=-μ0ρ[s M θ-∂(sM ρ) θ];R 1ɤρɤR 2;θs -ξ32ɤθɤθs +ξ32㊂üþýïïïïïïïïï(18)式中s M ρ和s M θ分别为永磁体磁化强度的径向分量与切向分量㊂对于径向磁化㊁极弧系数为1,且置于转子槽内的永磁体,sM ρ和sM θ为:sM ρ=Cðɕv =1,2,3,M ρv cos[τv (θ+ξ32-θs )];sM θ=C ðɕv =1,2,3,M θv sin[τv (θ+ξ32-θs )]㊂üþýïïïï(19)式中:C =1,交替极㊁磁极均为N 极;-1,交替极㊁磁极均为S 极;(-1)v -1,传统表面插入式㊂ìîíïïï(20)M ρv =4B r πvμ0sin 2(v π2);M θv =0㊂}(21)使用分离变量法,考虑非齐次偏微分方程的解的性质以及系数缩放,可得其解的最终形式为sA 5(ρ,θ)=s W 5+s X 5ln ρ+ðɕv =1,2,3,F v (ρ)ˑcos[τv (θ+ξ32-θs )]+ðɕv =1,2,3, [s v Y 5P (τv ,ρ,R 1)P (τv,R 2,R 1)+svZ 5P (τv ,ρ,R 2)P (τv ,R 2,R 1)]ˑcos[τv (θ+ξ32-θs )]㊂(22)式中:s v W 5㊁s v X 5㊁s v Y 5和s v Z 5分别为内层永磁子域矢量磁位的直流分量系数和谐波分量系数;τv 和F v (ρ)分别为:τv =v πξ3;(23)F v (ρ)=C -μ0ρτv M ρv -M θvτ2v -1,τv ʂ1;μ0ρln ρM ρv -M θv 2,τv =1㊂ìîíïïïï(24)式中v 为内层永磁子域矢量磁位的谐波阶数㊂综上所述,已建立5个子域中含待定系数的矢量磁位㊂其中i W 1㊁i X 1㊁i m Y 1㊁i m Z 1㊁j W 2㊁j X 2㊁j n Y 2㊁j n Z 2㊁k W 3㊁k X 3㊁k Y 3㊁k Z 3㊁u W 4㊁u X 4㊁u Y 4㊁u Z 4㊁s W 5㊁s v X 5㊁s v Y 5和s vZ 5共20组的系数(包括直流分量系数和谐波分量系数),将由边界条件确定㊂3㊀谐波系数求解利用在ρ=R 1㊁ρ=R 2㊁ρ=R 3㊁ρ=R 4㊁ρ=R 5以及ρ=R 6共6个位置处的边界条件建立矩阵方程,求解各子域矢量磁位中的各谐波分量系数㊂考虑到在ρ=R 1和ρ=R 6处,可对矢量磁位进行化简,因此,首先对这两处的边界条件进行求解,然后再求解其他位置㊂3.1㊀在ρ=R 1处的边界条件分析由图2可得,ρ=R 1为内层永磁子域与转子铁心的交界面,在此处有[s A 5(ρ,θ)]ρρ=R 1=0㊂(25)将式(22)代入式(25),可以解得s vX 5=0;(26)s vZ 5=-F ᶄv (R 1)P (τv ,R 2,R 1)R 1E (τv ,R 1,R 2)τv㊂(27)将式(26)代入式(22),得内层永磁子域的矢量磁位为sA 5(ρ,θ)=sW 5+ðɕv =1,2,3,F v (ρ)cos[τv (θ+ξ32-θs )]+ðɕv =1,2,3, [s v Y 5P (τv ,ρ,R 1)P (τv ,R 2,R 1)+s vZ 5P (τv ,ρ,R 2)P (τv ,R 2,R 1)]cos[τv (θ+ξ32-θs )]㊂(28)3.2㊀在ρ=R 6处的边界条件分析由图2可得,ρ=R 6为定子槽子域与定子槽底的交界面,此处满足[i A 1(ρ,θ)]ρρ=R 6=0㊂(29)将式(5)代入式(29),可以解得:iX 1=0;(30)imY 1=i m Z 1(R 6R 5)-τm㊂(31)将式(30)和式(31)代入式(5),可得槽子域的矢量磁位为64电㊀机㊀与㊀控㊀制㊀学㊀报㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀第27卷㊀iA 1(ρ,θ)=iW 1+ðတm =1,2,3,[i mZ 1(R 6R 5)-τm (ρR 6)τm ]ˑcos[τm (θ+ξ12-θi )]+ðɕm =1,2,3,[i m Z 1(ρR 5)-τm]ˑcos[τm (θ+ξ12-θi )]㊂(32)3.3㊀在ρ=R 5处的边界条件分析由图2可得,ρ=R 5为定子槽子域与定子槽口的交界面,根据径向气隙磁密连续的边界条件,可得:iA 1(R 5,θ)=j A 2(R 5,θ);θj -ξ22ɤθɤθj +ξ22㊂}(33)显然,定子槽口与定子槽为一一对应关系,因此在计算过程中,始终有i =j ㊂根据傅里叶级数的性质和边界条件式(33),在槽口子域矢量磁位的直流分量系数和谐波分量系数分别为:jW 2+jX 2ln R 5=1ξ2ʏθj +ξ2/2θj-ξ2/2jA 2(R 5,θ)d θ=1ξ2ʏθj +ξ2/2θj-ξ2/2iA 1(R 5,θ)d θ=iW 1+ðɕm =1,2,3,[i mZ 1(R5R 6)2τm +i m Z 1]ˑF 1(m );(34)-j n Z 2=2ξ2ʏθj +ξ2/2θj-ξ2/2j A 2(R 5,θ)cos[τm (θ+ξ22-θj )]d θ=2ξ2ʏθj +ξ2/2θj-ξ2/2iA 1(R 5,θ)cos[τm (θ+ξ22-θj )]d θ=ðɕm =1,2,3,[i mZ 1(R 5R 6)2τm +i m Z 1]2ξ2F 2(m ,n )㊂(35)式中:F 1(m )=ʏθj +ξ2/2θj -ξ2/2cos[τm (θ+ξ12-θi )]d θ;(36)F 2(m ,n )=ʏθj +ξ2/2θj -ξ2/2cos[τm (θ+ξ12-θi )]cos[τn (θ+ξ22-θj )]d θ㊂(37)在ρ=R 5处的另一边界条件为切向磁场强度连续,可表示为[i A 1(ρ,θ)]ρρ=R 5=[j A 2(ρ,θ)] ρρ=R 5,θj -ξ22ɤθɤθj +ξ22;0,其他㊂ìîíïïï(38)由式(38),根据傅里叶级数的直流分量系数和谐波分量系数的性质,可得:jX 3=0;(39)im Z 1(τm (R 5R 6)τm -1R 6(R 6R 5)τm-τm R 5)=ðɕn =1,2,3,[j n Y 22τnR 5E (τn,R 4,R 5)]F 2(m ,n )+ðɕn =1,2,3,[j n Z 2τn(R 5R 4)τn -1R 4+R 4R -25τn (R5R 4)τn -1E (τn ,R 4,R 5)E (τn ,R 4,R 5)]F 2(m ,n )㊂(40)于是可得,式(34)㊁式(35)㊁式(39)以及式(40)为谐波分量系数㊂由于后续各边界条件的分析求解过程类似,仅给出边界条件和得到的谐波分量系数方程,求解过程将不再赘述㊂3.4㊀在ρ=R 4处的边界条件分析由图2可得,ρ=R 4为定子槽口与气隙的交界面,满足的两个边界条件为:A 3(R 4,θ)=j A 2(R 4,θ),θj -ξ22ɤθɤθj +ξ22; A 3(ρ,θ) ρρ=R 4=㊀㊀ [jA 2(ρ,θ)] ρρ=R 4,θj -ξ22ɤθɤθj +ξ22;0,其他㊂ìîíïïïüþýïïïïïïïïïï(41)同样对矢量磁位函数进行傅里叶分解,可得其直流分量系数和谐波分量系数分别为:jW 2=ðɕk =1,2,3,[k W 3R 3k 2E (k ,R 3,R 4)+kX 3R 4k P (k ,R 4,R 3)E (k ,R 4,R 3)]F 3(k ,j ,ξ2)ξ2+ðɕk =1,2,3,[k Y 3R 3k 2E (k ,R 3,R 4)+kZ 3R 4k P (k ,R 4,R 3)E (k ,R 4,R 3)]F 4(k ,j ,ξ2)ξ2;(42)74第9期倪有源等:新型双层半插入式交替极永磁电机的解析建模j nY 2=ðɕk =1,2,3,[k W 3R 3k 2E (k ,R 3,R 4)]2F 5(n ,k ,j ,ξ2)ξ2+ðɕk =1,2,3,[k Z 3R 4k P (k ,R 4,R 3)E (k ,R 4,R 3)]2F 5(n ,k ,j ,ξ2)ξ2+ðɕk =1,2,3,[k Y 3R 3k 2E (k ,R 3,R 4)]2F 6(n ,k ,j ,ξ2)ξ2+ðɕk =1,2,3,[k Z 3R 4k P (k ,R 4,R 3)E (k ,R 4,R 3)]2F 6(n ,k ,j ,ξ2)ξ2;(43)-k X 3=ðN s j ðɕn =1,2,3, [j n Y 2τn (R τn -14R τn -25+Rτn 5R τn -34)E (τn ,R 4,R 5)]F 5(n ,k ,j ,ξ2)π+ðN sj ðɕn =1,2,3,[j n Z 22τnR 4E (τn,R 4,R 5)]F 5(n ,k ,j ,ξ2)π;(44)-k Z 3=ðN s j ðɕn =1,2,3,[j n Y 2τn (R τn -14R τn -25+R τn 5R τn -34)E (τn ,R 4,R 5)]F 6(n ,k ,j ,ξ2)π+ðN sj ðɕn =1,2,3,[j n Z 22τnR 4E (τn,R 4,R 5)]F 6(n ,k ,j ,ξ2)π㊂(45)式中:F 3(k ,j ,ξ2)=ʏθj +ξ2/2θj -ξ2/2cos(kθ)d θ;(46)F 4(k ,j ,ξ2)=ʏθj +ξ2/2θj -ξ2/2sin(kθ)d θ;(47)㊀㊀F 5(n ,k ,j ,ξ2)=ʏθj +ξ2/2θj -ξ2/2cos(kθ)cos[τn (θ+ξ22-θj )]d θ;(48)㊀㊀F 6(n ,k ,j ,ξ2)=ʏθj +ξ2/2θj -ξ2/2sin(kθ)cos[τn (θ+ξ22-θj )]d θ㊂(49)3.5㊀在ρ=R 3处的边界条件分析由图2可得,ρ=R 3为气隙区域和外层磁极区域的交界面㊂由于两区域均为环形区域,无需进行傅里叶分解,只需将对应的各阶谐波分量系数代入边界条件运算即可㊂在计算中,始终有k =u ,在此处的边界条件为:A 3(R 3,θ)=A 4(R 3,θ); [A 3(ρ,θ)]ρρ=R 3=1μr [A 4(ρ,θ)] ρρ=R 3㊂}(50)将矢量磁位的表达式代入式(50),可得:[k W 3R 3k P (k ,R 3,R 4)E (k ,R 3,R 4)+k X 3R 4k 2E (k ,R 4,R 3)]=[u W 4R 2u 2E (u ,R 2,R 3)+u X 4R 3u P (u ,R 3,R 2)E (u ,R 3,R 2)]+F u (R 3)sin(uθ0);(51)[k Y 3R 3k P (k ,R 3,R 4)E (k ,R 3,R 4)+k Z 3R 4k 2E (k ,R 4,R 3)]=[u Y 4R 2u 2E (u ,R 2,R 3)+u Z 4R 3u P (u ,R 3,R 2)E (u ,R 3,R 2)]-F u (R 3)cos(uθ0);(52)kW 3=-1μr [u X 4R 3u R k 3R k -22-R k2R k -43E (u ,R 3,R 2)-F ᶄu (R 3)sin(uθ0)];(53)kY 3=-1μr [u Z 4R 3u R k 3R k -22-R k 2R k -43E (u ,R 3,R 2)+F ᶄu (R 3)cos(uθ0)]㊂(54)3.6㊀在ρ=R 2处的边界条件分析由图2可得,ρ=R 2为外层磁极区域和转子槽的交界面㊂其边界条件为:A 4(R 2,θ)=j A 5(R 2,θ),θj -ξ22ɤθɤθj +ξ22; A 4(ρ,θ)ρρ=R 2=㊀㊀ [jA 5(ρ,θ)] ρρ=R 2,θj -ξ22ɤθɤθj +ξ22;0,其他㊂ìîíïïïüþýïïïïïïïïïï(55)同样对矢量磁位函数进行傅里叶分解,可得其直流分量和谐波分量系数的方程为:sW 5=ðɕu =1,2,3,[u W 4R 2u P (u ,R 2,R 3)E (u ,R 2,R 3)]F 3(u ,s ,ξ3)+ðɕu =1,2,3,[u X 4R 3u 2E (u ,R 3,R 2)+F u (R 2)sin(uθ0)]F 3(u ,s ,ξ3)+ðɕu =1,2,3,[u Y 4R 2u P (u ,R 2,R 3)E (u ,R 2,R 3)]F 4(u ,s ,ξ3)+ðɕu =1,2,3,[u Z 4R 3u 2E (u ,R 3,R 2)-F u (R 2)cos(uθ0)]F 4(u ,s ,ξ3);(56)84电㊀机㊀与㊀控㊀制㊀学㊀报㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀第27卷㊀㊀s vY 5P (τv ,R 2,R 1)P (τv ,R 2,R 1)+s v Z 52P (τv ,R 2,R 1)+F v (R 2)=ðɕu =1,2,3,[uW 4R 2u P (u ,R 2,R 3)E (u ,R 2,R 3)]F 5(v ,u ,s ,ξ3)+ðɕu =1,2,3,[uX 4R 3u 2E (u ,R 3,R 2)+F u (R 2)ˑsin(uθ0)]F 5(v ,u ,s ,ξ3)+ðɕu =1,2,3,[uY 4R 2u P (u ,R 2,R 3)E (u ,R 2,R 3)]F 6(v ,u ,s ,ξ3)+ðɕu =1,2,3,[u Z 4R 3u 2E (u ,R 3,R 2)-F u (R 2)cos(uθ0)]F 6(v ,u ,s ,ξ3);(57)uW 4=ðN r s ðɕv =1,2,3,[s v Y 5τv (R τv -12R τv 1-R τv 1R τv +11)P (τv ,R 2,R 1)]F 5(v ,u ,s ,ξ3)μr π+ðN rs ðɕv =1,2,3,[F ᶄv (R 2)]F 5(v ,u ,s ,ξ3)μr π-F ᶄu (R 2)sin(uθ0);(58)uY 4=ðN r s ðɕv =1,2,3,[s v Y 5τv (R τv -12R τv 1-R τv1R τv +11)P (τv ,R 2,R 1)]F 6(v ,u ,s ,ξ3)μr π+ðN r s ðɕv =1,2,3, [F ᶄv (R 2)]F 6(v ,u ,s ,ξ3)μr π+F ᶄu(R 2)cos(uθ0)㊂(59)联立式(26)㊁式(27)㊁式(30)㊁式(31)㊁式(34)㊁式(35)㊁式(39)㊁式(40)㊁式(42)~式(45)㊁式(51)~式(54)以及式(56)~式(59)共计20组方程,即可求解获得各区域中矢量磁位的直流分量系数和谐波分量系数㊂然后将气隙区域中的矢量磁位A 3(ρ,θ)代入式(2),可求解得出新型电机的气隙磁场㊂4㊀电磁性能比较和有限元验证新型电机定子采用集中绕组,即绕组的节距为1㊂对于任意的转子位置,每个线圈交链的磁链可以由线圈两个有效边的平均矢量磁位之差计算,然后由磁链的微分可以获得空载感应电动势㊂一个线圈的磁链可以表示为:ψx =ψx +-ψx -;(60)ψx +=LN c S ʏθi +ξ12θi ʏR 6R 5[1A 1(ρ,θ)]d ρd θ;ψx -=LN c Sʏθiθi -ξ12ʏR 6R5[2A 1(ρ,θ)]d ρd θ㊂üþýïïïï(61)式中:L 为轴向长度;N c 为线圈匝数;S 为每个槽的面积㊂每相绕组有N λ个线圈串联构成,每相串联匝数为N =N c N λ匝,那么该相总磁链为ψλ=ðN λx ψx ㊂(62)于是,该相的空载感应电动势为E =ωd ψλd θ0㊂(63)式中ω为电机的角速度㊂三相电机的电磁转矩表达式为T e =(E A i A +E B i B +E C i C )ω㊂(64)式中:E A ,E B ,E C 是三相感应电动势;i A ,i B 和i C 是三相对称的电枢电流㊂以下比较定子结构完全相同㊁8极9槽两种电机的电磁性能㊂并且这两种电机永磁用量㊁永磁外半径R 3及内半径R 1也相等㊂其中,传统表面插入式永磁电机的永磁占比为0.76;新型电机外层永磁极弧系数为1㊁内层永磁占比为0.5㊂表1给出了新型电机的主要结构参数㊂表1㊀新型双层半插入式交替极永磁电机的主要参数Table 1㊀Main parameters of novel double-layer semi-insetconsequent-pole permanent magnet machine图3比较了传统表面插入式永磁电机和新型双层半插入交替极永磁电机的气隙磁密波形和主要谐波分量㊂图4比较了这两种电机的空载感应电动势94第9期倪有源等:新型双层半插入式交替极永磁电机的解析建模的波形和主要谐波分量㊂图5比较了这两种电机的电磁转矩波形㊂图3㊀两种电机的气隙磁密波形Fig.3㊀Air gap flux density waveforms of twomachines图4㊀两种电机的空载感应电动势波形Fig.4㊀No-load back-EMF waveforms of twomachines图5㊀两种电机的电磁转矩波形Fig.5㊀Electromagnetic torque waveforms of twomachines表2比较了这两种电机的电磁性能㊂可以看出,两种电机气隙磁密和感应电动势所含的偶次谐波幅值较小,奇次谐波幅值较大㊂新型电机虽然气隙磁密和感应电动势的谐波含量较高,但气隙磁密基波幅值和感应电动势基波幅值都较大㊂由于内层交替极磁极具有不对称性,新型电机的径向气隙磁密波形和感应电动势波形中都含有较高的3次谐波㊂在永磁体用量相同的情况下,与传统表面插入式电机相比,新型电机虽然转矩脉动略微增加,但是平均电磁转矩有明显的提升,提高了10.4%㊂表2㊀两种电机的电磁性能比较Table 2㊀Electromagnetic performance comparison oftwo machines㊀㊀电机参数传统电机新型电机气隙磁密基波幅值/T 1.03 1.14气隙磁密THD /%20.0236.21感应电动势基波幅值/V 259.5283.3感应电动势THD /% 1.4313.78电磁转矩平均值/(N㊃m)7.48.17电磁转矩脉动/%3.243.67利用解析法和有限元法,计算得到新型电机的电磁性能,结果如图6所示㊂其中,感应电动势和电磁转矩的解析法结果略大于有限元结果,主要原因是计算各子域矢量磁位的直流分量和谐波系数时,由于受计算机硬件和软件限制,所取矩阵方程的阶数有限㊂但两种方法获得的波形总体一致性较好,验证了解析模型的正确性㊂5电㊀机㊀与㊀控㊀制㊀学㊀报㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀㊀第27卷㊀图6㊀气隙磁密㊁反电动势和电磁转矩波形有限元验证Fig.6㊀Finite element verification of air gap fluxdensity,back-EMF and electromagnetictorque waveforms5㊀结㊀论本文提出了一种新型双层插入式交替极永磁电机模型,在内层为交替极磁极的基础上,外层增加了表贴式永磁体㊂使用二维子域模型法,对其进行解析建模,推导获得了各子域内的矢量磁位和谐波系数㊂以8极9槽新型电机为例,计算得到电机的气隙磁密㊁空载感应电动势和电磁转矩㊂结果表明,在永磁用量相同的前提下,与传统的表面插入式电机相比,提出的双层半插入交替极永磁电机虽然转矩脉动略微增加,但是平均电磁转矩显著提高㊂最后通过有限元法验证了解析模型的正确性㊂参考文献:[1]㊀TONG W,LI S,PAN X,et al.Analytical model for coggingtorque calculation in surface-mounted permanent magnet motors with rotor eccentricity and magnet defects[J].IEEE Transactions on Magnetics,2020,35(4):2191.[2]㊀YOON K,KWON B.Optimal design of a new interior permanentmagnet motor using a flared-shape arrangement of ferrite magnets [J].IEEE Transactions on Magnetics,2016,52(7):1. 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往复走丝电火花线切割机床智能自适应采样控制系统和纳秒级高频电源的研发及应用

往复走丝电火花线切割机床智能自适应采样控制系统和纳秒级高频电源的研发及应用

往复走丝电火花线切割机床智能自适应采样控制系统和纳秒级高频电源的研发及应用潘伯郁(上海磐景自动化技术有限公司,上海200433)摘要:预设工件厚度的往复走丝电火花线切割机床的控制系统有其先天缺陷,会影响加工件精度和切割面粗糙度,这是因为大部分加工件的材质、形状和厚度不同,其加工过程中会因阶梯突变、缓慢渐变等工况使切割厚度发生变化。

研究显示,以Wire-CAXA控制系统模拟人类大脑,自动识别加工件的切割厚度和材质难度,实时动态调整加工状态,科学控制最佳放电频率、最优匹配工作台电机进给速度,是实现高精度加工的最佳解决方案。

关键词:线切割加工;波形;自适应;纳秒;Wire-CAXA控制系统中图分类号:TG661文献标志码:A文章编号:1009-279X(2020)06-0025-04Development and Application of Intelligent Control System Owned Self-adaptionSampling Function and High Frequency Power Supply of Nanosecond PulsedElectro-discharging for Reciprocating WEDMPAN Boyu(Shanghai Panjing Automation Technology Co.,Ltd.,Shanghai200433,China) Abstract:The traditional control system of the reciprocating WEDM with preset workpiece thickness has its inherent defects,which will affect the accuracy of the workpiece and the smoothness of the cutting surface.It is because that the material,shape and thickness of most workpieces are different,and abrupt changes,slow gradual changes and other working conditions will change the cutting thickness during the machining.Research shows that using the Wire-CAXA control system to simulate the human brain,automatically recognizing the cutting thickness and material difficulty of the processed parts,dynamically adjusting the processing status,scientifically controlling the optimal discharge frequency and optimally matching the feed speed of the table motor is the best solution for precision machining.Key words:WEDM;waveform;self-adaption;nanosecond;wire-CAXA control system经过不断的优化改进和长期的实践发展,往复走丝电火花线切割机床实现了多次切割工艺的广泛应用。

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a r X i v :c o n d -m a t /0405130v 3 [c o n d -m a t .s t r -e l ] 29 J u l 2005Multiple Bosonic Mode Coupling in Electron Self-Energy of (La 2−x Sr x )CuO 4X.J.Zhou 1,2,Junren Shi 3,T.Yoshida 1,4,T.Cuk 1,W.L.Yang 1,2,V.Brouet 1,2,J.Nakamura 1,N.Mannella 1,2,Seiki Komiya 5,Yoichi Ando 5,F.Zhou 6,W.X.Ti 6,J.W.Xiong 6,Z.X.Zhao 6,T.Sasagawa 1,7,T.Kakeshita 8,H.Eisaki 1,8,S.Uchida 8,A.Fujimori 4,Zhenyu Zhang 3,9,E.W.Plummer 3,9,ughlin 1,Z.Hussain 2,and Z.-X.Shen 11Dept.of Physics,Applied Physics and Stanford Synchrotron Radiation Laboratory,Stanford University,Stanford,CA 943052Advanced Light Source,Lawrence Berkeley National Lab,Berkeley,CA 947203Condensed Matter Sciences Division,Oak Ridge National Laboratory,Oak Ridge,TN 378314Dept.of Complexity Science and Engineering,University of Tokyo,Kashiwa,Chiba 277-856,Japan5Central Research Institute of Electric Power Industry,Komae,Tokyo 201-8511,Japan6National Lab for Superconductivity,Institute of Physics,Chinese Academy of Sciences,Beijing 100080,China7Department of Advanced Materials Science,University of Tokyo,Japan 8Dept.of Superconductivity,University of Tokyo,Bunkyo −ku,Tokyo 113,Japan 9Department of Physics and Astronomy,University of Tennessee,Knoxville,TN 37996(Dated:February 2,2008)High resolution angle-resolved photoemission spectroscopy data along the (0,0)-(π,π)nodal di-rection with significantly improved statistics reveal fine structure in the electron self-energy of the underdoped (La 2−x Sr x )CuO 4samples in the normal state.Fine structure at energies of (40∼46)meV and (58∼63)meV,and possible fine structure at energies of (23∼29)meV and (75∼85)meV,have been identified.These observations indicate that,in LSCO,more than one bosonic modes are involved in the coupling with electrons.PACS numbers:74.25.Jb,71.18.+y,74.72.Dn,79.60.-iThe recent observation of the electron self-energy renormalization effect in the form of a “kink”in the dispersion has generated considerable interest because it reveals a coupling of the electrons with a collective bo-son mode of the cuprate superconductors[1].However,the nature of the bosons involved remains controversial mainly because the previous experiments can only be used to determine an approximate energy of the mode and this energy is close to both the optical phonon[2,3]and the spin resonance[4].Determining the nature of the mode(s)that couple to the electrons is likely important in understanding the pairing mechanism of superconduc-tivity.In conventional superconductors,identification of the fine structure for the phonon anomalies in the tunnelling spectra has played a decisive role in reaching a con-sensus on the nature of the bosons involved[5].The fine structure provides fingerprints for much more strin-gent comparison with known boson spectra.So far,such fine structure has not been detected in the angle-resolved photoemission spectroscopy (ARPES)data.In this Letter we present significantly improved high resolu-tion ARPES data of (La 2−x Sr x )CuO 4(LSCO)that,for the first time,reveal fine structure in the electron self-energy,demonstrating the involvement of multiple boson modes in the coupling with electrons.The photoemission measurements were carried out on beamline 10.0.1at the ALS,using Scienta 2002and R4000electron energy analyzers.As high energy res-olution and high data statistics are crucial to identify fine structure in the electron self-energy,the experimen-tal conditions were set to compromise between these two conflicting requirements.The measurement is particu-larly challenging for LSCO system because of the ne-cessity to use a relatively high photon energy (55eV).Different energy resolution between 12and 20meV was used for various measurements on different samples,and the angular resolution is 0.3degree.An example of the high quality of the raw data is shown in Figs.1a and 1b.Due to space charge problem,the Fermi level cali-bration has a ±5meV uncertainty.We mainly present our data on the heavily underdoped LSCO x=0.03(non-superconducting),LSCO x=0.063(T c =12K)and LSCO x=0.07(T c =14K)samples.These heavily underdoped LSCO samples are best candidates because they exhibit a stronger band renormalization effect above T c [3];a rel-atively large magnitude of the real self-energy makes the identification of the fine structure easier.The LSCO sin-gle crystals are grown by the travelling solvent floating zone method[6].The samples were cleaved in situ in vac-uum with a base pressure better than 4×10−11Torr.The measurement temperature was ∼20K so all samples were measured in the normal state.Fig.1c shows the energy-momentum dispersion rela-tion along the (0,0)-(π,π)nodal direction extracted by the MDC (momentum distribution curves)method.Be-cause of the larger band-width along the nodal direction,the MDC method can be reliably used to extract high quality data of dispersion in searching for fine structures.It has also been shown theoretically that this approach is reasonable in spite of the momentum-dependent cou-pling if we are only interested in identifying the mode energies[7].As seen in Fig.1c,there is an abrupt slope change (“kink”)in the dispersions for LSCO samples2k - k F (A -1)E - EF (e V )Momentum k (A -1)E - EF (eV)E - E F (e V )FIG.1:(a)Raw data of a two-dimensional image showingthe photoelectron intensity as a function of momentum and energy for LSCO x=0.063sample.The intensity is repre-sented by the false-color.The measurement was taken along the (0,0)-(π,π)nodal direction at a temperature of ∼20K.(b)The photoemission spectra (energy distribution curves,EDCs)for the LSCO x=0.063sample corresponding to Fig.1a.The spectrum at the Fermi momentum k F =(0.44π/a ,0.44π/a )(red curve)shows a sharp peak.(c)The energy-momentum relation determined from MDC method for LSCO x=0.03(c1),x=0.063(c2)and x=0.07(c3)samples.The green dashed lines connecting the two points at the Fermi energy and -0.2eV are examples of a simple selection of the bare band.with different dopings,similar to that reported before[3].However,the new data with improved statistics indicates that the “kink”has fine structure and subtle curvatures in it,as seen for example in the LSCO x=0.03sample (Fig.1c1).Since the bare dispersion is expected to be smooth in such a small energy window,the “kink”and its fine struc-ture represent effects associated with the electron self-energy.The real part of the electron self-energy,ReΣ,can be extracted from the measured dispersion (as in Fig.1c)by subtracting the “bare dispersion”.Following the convention[8,10],within a small energy range near the Fermi level,the “bare dispersion”can be assumed as ǫ0(k)=α1(k-k F )+α2(k-k F )2.As we will describe later,the values of α1and α2are determined so as to yield the best fit of the measured dispersion from the maxi-mum entropy method (MEM);the choice of smooth bare band has little effect on the fine structure that originate from the abrupt change in the dispersion.The “effective”ReΣfor LSCO at various doping levels is shown in Fig.2a.Also note the data were taken at different energy resolutions,using different analyzers and under different measurement conditions,as described in the caption of Fig.2.As seen from Fig.2a,even with the most optimal ex-perimental conditions we can achieve and in these sam-ples that exhibit the strongest self-energy effects,there remains considerable noise in the data statistics.How-ever,by searching for peaks or curvature changes,wecan clearly identify some fine structure in the data.Two clear features are around 40∼46meV and 58∼63meV,as indicated by arrows in Fig.2a,which show up clearly in x=0.03(Fig.2a1)and ∼0.06samples (Fig.2a4),although less clearly in x=0.063and x=0.07samples.There may be another structure near 23∼29meV,which shows up mainly as a shoulder that is visible in x=0.03(Fig.2a1),0.063(Fig.2a2),0.07(Fig.2a3)and pos-sibly in x=∼0.06sample (Fig.2a4).While these fine features are subtle and one may argue about individual curves,the fact that we have invariably observed them near similar energies in many different samples and un-der different measurement conditions make the presence of the fine structure convincing.A natural question is whether the fine structure can be due to instrumental artifact which may be related to detector inhomogeneity and/or system noise.The detec-tor problem can be ruled out because:(1).The data were taken using the “swept mode”of the electron ana-lyzer,i.e.,each energy point was averaged over the en-tire detector range along the energy direction.Therefore,all the energy points for the same angle were taken un-der the same condition.(2).The inhomogeneity along the angle direction is also minimized by normalizing the photoemission spectra with the spectral weight above the Fermi level which comes from the high harmonics of syn-chrotron light and is angle-insensitive.(3).Among differ-ent measurements,the corresponding detector angle with respect to the dispersion bands vary due to slight changes in measuring geometry.The fact that we observed con-sistent results suggest that the results are intrinsic;(4).We have carried out our measurements using different analyzers,from SES2002to the latest R4000,yielding qualitatively similar results.One can also rule out the possibility of noise because they are supposed to be ran-dom.But for many different measurements on different samples and under different experimental conditions,the multiple structures are similar in energy within the un-certainty caused by data statistics[9].The direct inspection in the raw data (Fig.2a)has clearly established the existence of fine structure in the electron self-energy.This is independent of any mod-els that are used in the data analysis,including the MEM method we use below.In order to better quan-tify the characteristic energies of these fine features,we take an approach to fit a smooth curve through the mea-sured ReΣdata and then perform second-order deriva-tive to the fitted curve.Given the statistics level in the data,a spline through the data is difficult because it is somewhat subjective.The MEM procedure,which has been exploited to extract the spectral features of the electron-phonon coupling from ARPES data for the two-dimensional surface state of Be[10],is well suited for this purpose.By incorporating prior knowledge,including positiveness of the bosonic function,and zero value of3the bosonic function at zero frequency and above a max-imum frequency of100meV,the MEM is robust against the random noise in the data,insensitive to thefitting de-tails,and is therefore more objective[10].Since it remains unclear whether the method developed for conventional metal can be extended to cuprate superconductors,as thefirst step,we use it only as a procedure for curve fitting.Thefitted curves are shown in Fig.2a together with the measured ReΣ;the corresponding second-orderderivative of thefitted curves are shown in Fig2b.As expected,two dominant features near40∼46meV and 58∼63meV,and one possible feature near23∼29,have been resolved in Fig.2b.This is consistent with the fine structure identified to a naked eye in the raw data of the electron self-energy(Fig.2a).In addition,this data analysis process also allows us to identify one more pos-sible feature that may exist at high energy near75∼85 meV,which seems to get stronger with increasing dop-ing.We note that,while the agreement is not perfect at this stage,there is sufficient similarity to suggest the detection of multiple modes.Thefine structure in the electron self-energy originates from the underlying bosonic spectral function.The mul-tiple features in Fig.2b show marked difference from the magnetic excitation spectra measured in LSCO which is mostly featureless and doping dependent[11].In compar-ison,the features in Fig.2b show more resemblance to the phonon density-of-states(DOS),measured from neu-tron scattering on LSCO(Fig.2c)[12],in the sense of the number of modes and their positions.This similarity between the extractedfine structure and the measured phonon features favors phonons as the nature of bosons involved in the coupling with electrons.In this case,in addition to the half-breathing mode at70∼80meV that we previously considered strongly coupled to electrons[2], the present results suggest that several lower energy op-tical phonons of oxygens are also actively involved.For conventional metals,the MEM procedure can be used to extract the Eliashberg function which gives spec-tral features of the electron-phonon coupling[10].For strongly correlated cuprate superconductors,a priori it is unclear whether the Eliashberg formalism is applica-ble or not.However,we note that the nodal excitation of LSCO in the normal state may provide a closest case for the procedure to be applicable.Recent transport mea-surements reveal that a quasiparticle picture may be still reasonable for electrons near the nodal direction,even for very low doping[13].This is consistent with the ARPES data which show a well-defined peak in the nodal spec-tra in the lightly doped samples[3].The nodal dispersion does not have pronounced curvature in its bare band, unlike the strong curvature near the saddle point of the antinodal region.The selection of the nodal direction in the normal state also minimizes complications due to the existence of a superconducting gap or pseudogap in theEffectiveReΣ(meV)SecondDerivativeofFittedReΣ(Arb.Unit)E - EF(eV)Phonon Energy (meV)PhononDOS(meV-1)FIG.2:(a).The effective real part of the electron self-energy for LSCO x=0.03(a1),0.063(a2),0.07(a3)and∼0.06(a4) samples.Data(a1-a3)were taken using Scienta2002ana-lyzer,10eV pass energy at an overall energy resolution(con-voluted beamline and analyzer resolution)of∼18meV.Data (a4)were taken using Scienta R4000analyzer,5eV pass en-ergy at an overall energy resolution of∼12meV.For clarity, the error bar is only shown for data(a4)which becomes larger with increasing binding energy.The arrows in thefigure mark possiblefine structures in the self-energy.The data arefitted using the maximum entropy method(solid red lines).The values of(α1,α2)(the unit ofα1andα2are eV·˚A and eV·˚A2, respectively)for bare band are(-4.25,0)for(a1),(-4.25,13) for(a2),(-3.7,7)for(a3)and(-4.3,0)for(a4).(b).The second-order derivative of the calculated ReΣ.The rugged-ness in the curves is due to limited discrete data points.The four shaded areas correspond to energies of(23∼29),(40∼46), (58∼63)and(75∼85)meV where thefine features fall in.(c) The phonon density of state F(ω)for LSCO x=0(red)and x=0.08(blue)measured from neutron scattering[12].extraction process.Given these considerations and the fact that there is no better alternative available,we made an attempt by applying the Eliashberg formalism and the MEM proce-dure to extract the effective bosonic function from the real part of the electron self-energy(Fig.3a).It is clear that the multiple features are rather robust against the choice of the bare band by varyingα1andα2.All other tests as detailed in[10]have been carried out.Thefine structure as obtained from LSCO x=0.03is in agree-ment with that in the second-order derivative shown in Fig.2b.The calculated real part of the electron self-energy is plotted in Fig.3a together with the measured data.Fig.3b shows the MDC width which is directly related to ImΣ=(Γ/2)v0,withΓbeing the MDC width (Full-Width-at-Half-Maximum,FWHM)and v0the bare40.20.150.10.050-0.2-0.15-0.1-0.050-0.1-0.050.5E f f e c t i v e B o s o n i cF u n c t i o n Measured el-ph el-el impurity TotalbM D C W i d t h (A -1)E - EF (eV)E - EF (eV)E f f e c t i v e R e Σ (m e V )FIG.3:(a)Real part of the electron self-energy for LSCO x=0.03as obtained from the dispersion shown in Fig.1c1(solid square)and calculated from the extracted effective bosonic spectral using the MEM procedure (red solid curve)with α1=-4.25and α2=0.Also plotted are the effective bosonic functions obtained by using different bare bands as represented by the different sets of α1and α2values.(b)The MDC width of LSCO x=0.03(open circles).The contri-bution from the electron-phonon coupling (blue line)is cal-culated from the effective bosonic function in Fig.3a with α1=-4.25and α2=0.The “impurity”contribution is assumed to be a constant,0.053˚A −1(dotted black line).The momen-tum resolution here is 0.019˚A −1.After subtracting all the electron-phonon and “impurity”contributions,the residual part is fitted by C ωαwith C ∼0.7and α∼1.5(green line).velocity.While there is a drop near 75meV,there is an overall increase of the MDC width with increas-ing binding energy (Fig.3b),which is different from simple electron-phonon coupling systems such as Be[8].The MEM analysis allows us to calculate the contribu-tion of the electron-phonon coupling which gives rise to the abrupt drop in ImΣ.We note that this calculation has some uncertainty related to the bare band selection (Fig.3a).After subtracting the contributions from the electron-phonon coupling,“impurity”scattering,and an-gular resolution,the residual part is found to be propor-tional to ωα(Fig.3b).This term most likely represents the contribution of the electron-electron interaction.The corresponding electron-electron contribution in the real part of the self-energy is a smooth function and may be absorbed into the “bare dispersion”because we fo-cus only on abrupt structure in ReΣin extracting the bosonic spectral function.Here we also note that while the imaginary part of the electron self-energy is consis-tent with the existence of electron-phonon coupling,it is difficult to identify the fine structure as has been done for the real part because of the larger experimental un-certainty in determining the peak width over the peak position.This analysis also shows that there is an inter-nal consistency in the MEM procedure that connects the real and imaginary parts of the self-energy.In summary,by taking high resolution data on heavily underdoped LSCO samples with high statistics,we havedetected fine structure in the electron self-energy.This indicates multiple bosonic modes are involved in the cou-pling to electrons in the LSCO system.The work at the ALS and SSRL is supported by the DOE’s Office of BES,Division of Material Sci-ence,with contract DE-FG03-01ER45929-A001and DE-AC03-765F00515.The work at Stanford was also sup-ported by NSF grant DMR-0304981and ONR grant N00014-04-1-0048-P00002.EWP is supported by DOE DMS and NSF-DMR-0451163.The work at Oak Ridge National Laboratory was partially supported through DOE under Contract DE-AC05-00OR22725.The work in Japan is supported by a Grant-in-Aid from the Min-istry of Education,Culture,Sports,Science and Tech-nology of Japan and the NEDO.The work in China is supported by NSF of China and Ministry of Science and Technology of China through Project 10174090and Project G1999064601.[1]A.Damascelli,Z.-X.Shen and Z.Hussain,Rev.Mod.Phys.75(2003)473.[2]nzara et al.,Nature (London)412,510(2001).[3]X.J.Zhou et al.,Nature (London)423,398(2003);X.J.Zhou,et al.,Phys.Rev.Lett.92187001(2004);T.Yoshida et al.,Phys.Rev.Lett.91,027001(2003).[4]A.Kaminski et al.,Phys.Rev.Lett.86,1070(2001);P.D.Johnson et al.,Phys.Rev.Lett.87,177007(2001);S.V.Borisenko et al.,Phys.Rev.Lett.90,207001(2003).[5]J.M.Rowell et al.,Phys.Rev.Lett.10,334(1963);D.J.Scalapino et al.,Phys.Rev.148,263(1966).[6]S.Komiya et al.,Phys.Rev.B 65,214535(2002); F.Zhou et al.,Supercon.Sci.Technol.16,L7(2003).[7]A.W.Sandvik et al.,cond-mat/0309171;T.P.Dev-ereaux et al.,Phys.Rev.Lett.93,117004(2004).[8]shell et al.,Phys.Rev.B 61,2371(2000).[9]From the simulations we have done,a finite momentumresolution has little effect on the fine structures in the dispersion extracted from the MDC method.High energy resolution is important to identify the fine structures.[10]J.R.Shi et al.,Phys.Rev.Lett.92,186401(2004).[11]The spin excitation spectrum of LSCO x=0.14shows abroad peak at a lower energy (∼20meV)(S.M.Hay-den et al.,Phys.Rev.Lett.76,1344(1996)).This peak is pushed down to below 5meV in LSCO x=0.07(H.Hiraka et al.,J.Phys.Soc.Jpan 70,853(2001))and in x=0.05(H.Goka Physica C 388-389,239(2003)).A mea-surement from stripe ordered La 1.875Ba 0.125CuO 4and re-lated calculation showed a broad feature centered around 50∼60meV (Tranquada et al.,Nature 429(2004)534).Given that this is a different material at different dop-ing,we do not consider this as relevant to the LSCO x=0.03∼0.07.The rapid change of spin spectra with doping has also been observed in YBa 2Cu 3O 7−δ,S.Chakravarty et al.,Phys.Rev.B 63,094503(2001)).[12]R.J.McQueeney et al.,Phys.Rev.Lett.87,077001(2001);L.Pintschovious and M.Braden,Phys.Rev.B 60,R15039(1999).[13]Y.Ando et al.,Phys.Rev.Lett.87,017001(2001).。

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