Is high amplitude propagated

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High Energy Positrons and the WMAP Haze from Exciting Dark Matter

Recently, a number of diﬀerent experiments have made measurements which are consistent with excesses in positrons and/or electrons that may be indicative of a dark matter signal. Two of these (HEAT and the WMAP haze) are indicative of high energy particles, whereas the results of INTEGRAL/SPI are indicative of low energy particles.
arXiv:0802.2922v1 [astro-ph] 20 Feb 2008
I. INTRODUCTION
An overwhelming amount of evidence has established cold dark matter (CDM) as the standard paradigm for the missing matter of the universe. Beginning with early observations of velocity dispersions of galaxies in clusters [1], and later measurements of galactic rotation curves [2, 3], CDM has been supported by numerous additional observations. Some of these include strong lensing of background galaxies [4], and x-ray emission from galaxy clusters [5], the combination of CMB and type Ia supernovae data [6], measurements of the distributions of galaxies [7, 8], as well as the highly remarkable recent study of the bullet cluster [9].

Proximal colonic

Proximal colonic propagating pressure waves sequences and their relationship with movements of content in the proximal human colonP.G.DINNING,M.M.SZCZESNIAK&I.J.COOKDepartment of Gastroenterology,The St.George Hospital,University of New South Wales,Sydney,AustraliaAbstract Abnormal colonic motor patterns have been implicated in the pathogenesis of severe constipation. Yet in health,the mechanical link between movement of colonic content and regional pressures have only been partially deﬁned.This is largely due to current methodological limitations.Utilizing a combination of simultaneous colonic manometry,high-resolution scintigraphy and a quantitative technique for detect-ing discrete episodicﬂow,our aim was to examine the propulsive properties of colonic propagating sequences (PS)in the healthy colon.In six healthy volunteers a nasocolonic manometry catheter was positioned to record colonic pressures at7.5cm intervals from ter-minal ileum to the splenicﬂexure.With subjects positioned under a gamma camera,30MBq of99m Tc sulfur colloid was instilled into the terminal ileum, 22.5cm proximal to the ileocolonic junction.Isotopic images were recorded(10s/frame)and synchronized with the manometric trace.In the proximal colon we identiﬁed137antegrade PSs,of which93%were deemed to be associated temporally with movements of luminal content.Low amplitude PSs,with compo-nent pressure waves between2mmHg and5mmHg, were as likely to be associated with colonic move-ments as higher amplitude PSs.As such there was no correlation between the amplitude of the PS and the temporal relationship with colonic movements. Within the proximal colon,24retrograde PSs were identiﬁed,23of which were associated with retro-grade movements of colonic content.We conclude that proximal colonic PSs are highly propulsive and are a major determinant of proximal colonicﬂow.Keywords automated analysis,colon,propagating sequences,scintigraphy,transit.INTRODUCTIONColonic propagating pressure waves or propagating sequences(PS)are reported to be an important deter-minant of luminal propulsion and defecation.1–9 Altered colonic PS characteristics may represent important markers of colonic dysfunction.10–13How-ever as manometry can be an insensitive measure of wall motion,particularly in large calibre chambers such as the colon where non-lumen occluding contractions are common,14the clinical signiﬁcance that can attributed to manometricﬁndings remains undetermined.Animal studies have shown that colonic PSs have a close temporal relationship with actual movements of content.15–18Human studies on the other hand have either failed to demonstrate a relationship,16shown a relatively poor relationship1or been unable to establish a real time correlation between all pressure events and ﬂow.4–7Unlike animal studies that can utilize pro-longed,continuous radiology,human studies are con-strained by the radiation exposure and therefore rely on scintiscanning that can be limited by relatively slow frame,capture rates.4–7Other factors,which may hinder human studies,include a reliance upon the relatively insensitive method of visual analysis of isotopicﬂow,1and in all studies a substantial inter-sidehole distance yields a relativelyÔlow resolutionÕmanometric recording which inevitably underscores the frequency of propagating events.19Combined,Address for correspondenceDr.Philip Dinning,Department of Gastroenterology,The StGeorge Hospital,Kogarah,NSW,2217Australia.Tel:+61291132208;fax:+61291133993;e-mail:P.dinning@.auReceived:10September2007Accepted for publication:24October2007A section of this work has been presented in abstract form atthe International Motility Society meeting in Barcelona Spain(October5–8,2003).Dinning PG,Szczesniak M,Cook IJ.The relationships amongcaecalﬁlling and ileal and colonic propagating sequences.Neurogastroenterol Mot2003;15:591–681:A299.Neurogastroenterol Motil(2008)20,512–520doi:10.1111/j.1365-2982.2007.01060.xÓ2008The AuthorsJournal compilationÓ2008Blackwell Publishing Ltd512these technical deﬁciencies limit the precision with which any temporal relationship between pressure and movement can be detected.Recently we developed and validated a quantitative, scintigraphic method able to accurately detect discrete episodic movements of colonicﬂow on a moment-by-moment basis over prolonged periods.20Therefore, by combining this high-resolution scintigraphic technique with closely spaced recording sites in the proximal human colon we aimed to determine the relative propulsive nature of colonic PSs.Speciﬁcally, we hypothesized that;(i)all colonic PSs propel luminal content;and(ii)episodes of movements of content that are associated with low amplitude PSs are detectable by high resolution scintigraphy.METHODSSubjectsWe studied six healthy subjects(two male and four female)with a mean age of22.7±2.8years.All had normal bowel habits,deﬁned as between two bowel movements a day and one bowel movement every 2days.None was taking regular medications including laxatives.None had a history of prior abdominal surgery,other than appendectomy.All gave written, informed consent and the study was approved of by the Human Ethics Committees of the South Eastern Area Health Service,Sydney and the University of New South Wales.Manometric techniqueWe used a21lumen(16recording sideholes)4.5m long extruded silicone perfused manometric assembly (DentSleeve,Wayville,South Australia,Australia), with an overall diameter of 4.2mm.Each of the recording lumina had an internal diameter of0.4mm with an inter-sidehole distance of7.5cm.Tantalum slugs imbedded with silicone adjacent to each of the sideholes permittedﬂuoroscopic localization of each sidehole.In addition2–4MBq of57Co was sealed within the silicone catheter at sideholes1,4,7,10, 13and16to allow localization of sideholes during scintiscanning.A silicone balloon at the tip of the catheter,which could be inﬂated and deﬂated with water through two0.4mm lumina,assisted catheter passage through the gut.The catheter was rendered radio-opaque along its entire length byﬁlling the larger centre core(I.D. 1.9mm)with a1:1mixture of barium sulphate and sylgard(silicone surfactant, Wilbur-Ellis Company,San Francisco,CA,USA).The recording lumina were perfused with degassed distilled water preceded by a CO2ﬂush to ensure removal of all air bubbles from the catheter,thus minimizing compliance.21A low compliance pneu-mohydraulic perfusion pump drove the perfusate at 0.15mL/min(DentSleeve).Pressures were measured from each sidehole with16external pressure trans-ducers(Abbott Critical Care Systems,North Chicago, IL,USA).Signals were ampliﬁed and digitized at 10Hz by preampliﬁers(AqcKnowledge III Soft-ware,BIOPAC Systems,Inc.Santa Barbara,CA, USA).We have previously demonstrated that the rise rate characteristics afforded by a catheter of this nature is adequate for recording colonic pressure waves.9Experimental protocolThe technique of nasocolonic manometry of the unprepared colon has been described in detail else-where.9,19All subjects ate standard meals(breakfast: 500Kcal,15%protein,34%fat,51%carbohydrate; lunch and dinner:1000kCal,24%protein,43%fat, 33%carbohydrate)during the positioning of the cath-eter on days1and 2.On day3a continuous manometric recording commenced at0800h after a breakfast500kCal taken at0745h.With subjects supine point markers were placed on the right iliac crest and xiphisternum in order to monitor body movement during scintiscanning.Anterior and poster-ior isotope images were then recorded in13-min time intervals at10s/frame(80frames per13min)by a scintillation camera(Picker Prisim2000XP;Picker International,Cleveland,OH,USA)with a medium energy collimator.The camera was linked to a dedi-cated computer.There was a2-min gap between each 13-min recording interval,during which data was saved.At the commencement of each13min scinti-graphic recording period a mark was registered on the manometric trace,which facilitated synchronization of scintigraphic and manometric recording.Data were recorded for4–6h.Distinction between ileal and colonic pressure trac-ings was readily apparent from the characteristic morphology of the tracings from the two regions. Compressing the tracings to better appreciate the characteristic ileal motor pattern facilitated this pro-cess.At the commencement of the scintigraphic recording,30MBq of99m Tc sulphur colloid in1mL of saline was instilled into the terminal ileum,via a recording sidehole in the manometry catheter,approx-imately22.5cm proximal to the ileocolonic junction. The total effective radiation dose to each subject fromVolume20,Number5,May2008Determinants of colonicﬂow Ó2008The AuthorsJournal compilationÓ2008Blackwell Publishing Ltd513ﬂuoroscopy and scintigraphy including the cobalt markers within the catheters was equivalent to 4.8mSv.Data analysisManometric deﬁnitions and analysis The proximal colon,for the purposes of analysis,was divided into eight regions extending from cecum(region1),through hepaticﬂexure(region4)to the splenicﬂexure(region 8).A PS was deﬁned as an array of three or more pressure waves recorded from adjacent recording sites which displayed a conduction velocity of0.2–12cm/s. Propagating sequences were further qualiﬁed by the terms antegrade or retrograde,depending upon the direction of propagation.9,22Antegrade PSs were sub-classiﬁed as high amplitude propagating sequences (HAPS)if the amplitude of at least one component propagating pressure wave was‡116mmHg.This value represents the mean plus two standard devia-tions of propagating pressure waves from the mid colon in our previously collected healthy control data.19In order to address the hypothesis that low ampli-tude PSs are propulsive of content,a further sub-group was deﬁned asÔlow amplitude PSÕif the PS comprised two or more component pressure waves with a trough to peak amplitude between2mmHg and5mmHg but which satisﬁed the same conduction velocity criteria as for PS described above.Quantitative(high resolution)identiﬁcation of isotope movement Quantitative analysis of isotope movement was performed using the MrVoxel image analysis soft-ware.23This technique has been validated and is described in detail previously.20Brieﬂy,a composite image was obtained by combining all80frames cap-tured during a13-min epoch by the camera positioned anterior to the abdomen.This yielded an overall image of isotope distribution within the small and large bowel, enabling it to be divided anatomically into regions of interest(ROI).Using this composite image,ROI were constructed to measureﬂow across the(i)mid-ascend-ing colon and(ii)mid-transverse colon(Fig.1).Con-struction of the ROIs was established for each subject. Isotopeﬂow through each region was measured inde-pendent ofﬂow through adjacent regions.Applying these ROIs to the geometric mean of the anterior and posterior images,produced time-activity curves.In the absence of noise and gross movement of the subject,the slope of the regional time-activity curves may be interpreted as the rate of net inﬂow or outﬂow from the corresponding region.For the pur-poses of this study,declining or increasing activity over three or more consecutive frames were considered to indicate a potential movement of isotope into or out of that region respectively.For each of these potential movements,the index of theﬁrst and last frame were recorded on a grid,on which each box of the grid represented a10s frame.Grids from adjacent ROIs within a13-min recording period were compared. Where three or more consecutive frames demonstrat-ing declining counts coincided with three or more frames demonstrating increasing counts in an aboral adjacent region,it was deemed that antegradeﬂow had occurred while the converse indicated retrogradeﬂow. False-positiveﬂow episodes were removed from anal-ysis through a previously validatedﬂow value index.20 Correlation between scintigraphically-determined ﬂow and propagating sequences The next step was to assign the labelÔpropulsiveÕorÔnon-propulsiveÕto each PS by determining whetherﬂow occurred as a direct consequence of each PS.The time of onset of each PS was superimposed on theﬂow grids(refer quantitative scintigraphic analysis above).A scintigraphically-determinedﬂow episode was considered to be associ-ated temporally with a PS if the onset of the PS occurred not more than three frames(30s)prior to and not more than two frames(20s)after the onset of the isotope movement.This range was determined from our previous study in which we simulatedﬂow between adjacent ROI in a phantom model and ran the computer analytical algorithm to detect these simu-lated isotope movements.Our data showed that the computer algorithm placed the onset of95%these movements within a time window commencing30s Figure1Composite images of the technician in the(A) terminal ileum and proximal colon and(B)throughout the colon.The composite image was used to mark the regions of interest(ROI).These ROI were used to measureﬂow across (C)the mid-ascending colon and(D)mid-transverse colon.P.G.Dinning et al.Neurogastroenterology and MotilityÓ2008The AuthorsJournal compilationÓ2008Blackwell Publishing Ltd514prior and up to20s following the actual isotope movement(i.e.,total time window of50s).20All PSs found to be temporally associated with isotope move-ment in this manner were deﬁned asÔpropulsive PSsÕ, the remainder were consideredÔnon-propulsiveÕ. Following stratiﬁcation of PS in this manner,the pressure wave amplitude,conduction velocity of component propagating pressure waves and extend of propagation were compared between propulsive and non-propulsive PS in order to deﬁne how these vari-ables relate to the phenomenon of propulsion by a PS. Statistical analysis and determination of association probability To test whether a genuine association existed between PSs and episodicﬂow the association probability was calculated as previously described.24,25 Each of the13min grids upon whichﬂow episodes and PSs were detailed(see above)were divided into con-secutive60s periods(six10s frames).The number of these60s periods varied amongst regions and volun-teers and was dependant upon the presence of isotope within the region.The selection of a60s period was based upon the50s window in which our computer algorithm placed the onset of95%of isotope move-ments in our pilot studies20(see previous section). However,as the scintigraphic computer retrieving data from the collimator had to acquire data over13min intervals it was necessary to choose a time window that was a multiple of13and as close to50s as pos-sible.Hence,a60s time window was adopted.The number of these60s periods varied amongst regions and volunteers and was dependant upon the presence of isotope within the region.All60s periods were then evaluated for the existence ofﬂow(F+:ﬂow present;F):ﬂow absent)and for the presence of a PS (PS+:propagating sequence present;PS):propagating sequence absent).A60s period was considered to be ﬂow positive ifﬂow occurred over a minimum of two consecutive frames within the60s period.A60s period was considered to be PS positive if the point in time at which the PS was initiated occurred within the period.From these data a2·2contingency table was constructed with fourﬁelds:Field one contained the number of PSﬂow positive60s periods(PS+F+),ﬁeld two the number ofﬂow positive episodes without PSs (PS)F+),ﬁeld3the number of PS positive periods withoutﬂow(PS+F))andﬁeld4the number of periodswithoutﬂow or PSs(PS)F))(Fig.2).Individual contin-gency tables were constructed for each of the colonic regions(ascending and transverse colon)in each of the subjects.The next stage of the analysis determined the X1 relation between the number of PS positive60s periods and the total number of60s periods and X2 relation between the number of PSﬂow positive periods and the total number of Flow positive periods. If X2<X1it indicated that PSs coincided signiﬁcantly less than could be expected if the association between PSs andﬂow occurred by chance.In such instancesthevalue of1was assigned to the P value resulting in a ﬂow associated probability of0.If X2>X1then the calculation ofﬂow associated probability was possible. FisherÕs Exact Test was then determined from the contingency table to test the probability(P value)that the observed association between PSs andﬂow occurred by chance.Theﬂow associated probability was calculated as(1)P value)·100%.A value of>95% deﬁned as an association between the PS andﬂow,a value<95%indicated no association.The statistical comparison of the degree of associa-tion between PSs andﬂow was performed with|2 parison between the amplitude and velocity of PS associated withﬂow and those that were not associated was performed with a standard two-tailed T test.Data are presented as mean±SD. RESULTSCatheter placementThe catheter tip was located at the mid-transverse colon(n=2),mid descending colon(n=2)and sigmoid colon(n=2)prior to recording.As a result these data represent combined manometry and scintigraphy from the ascending colon in n=6and from the transverse colon in n=4.Propagating sequences:distribution,frequency and amplitudeIn all,585min of combined manometric and isotopic ﬂow data were collated across the mid-ascending colon and273min across the mid-transverse colon.Ante-grade PSs were recorded in each of the colonic regions, with96originating at or proximal to the mid-ascending colon and41PSs originating from a site proximal to the mid-transverse colon and extending up to or through the mid-transverse colon.Of the41PSs which extended through the mid-transverse colon,17had their site of origin in the cecum,the remaining24had their site of origin at or distal to the hepaticﬂexure. There were no HAPSs detected during the recording time.In comparison to antegrade PSs,retrograde PSs were signiﬁcantly less prevalent(P<0.0001)(Table1). Eight low amplitude antegrade PS were identiﬁed with a site of origin proximal to the mid-ascending colon and four with a site of origin distal to the mid-ascending colon and proximal to the mid-transverse colon(Table1).Five low amplitude retrograde PSs were identiﬁed,with three distal to the mid-ascending and two distal to the mid-transverse colon. Propagating sequences:relationship toluminalﬂowOver93%of antegrade(Fig.3)and retrograde(Fig.4) PSs were temporally associated with the propulsion of luminal contents(Table1).There was no apparent difference in PS amplitude or velocity between those PS associated withﬂow and those that were not.The antegrade low amplitude PSs also displayed a strong temporal association with episodicﬂow(91.6%)(Fig.3) while only two ofﬁve(40%)low amplitude retrograde PS were temporally associated with retrogradeﬂow (Table1).Theﬂow-associated probability could be calculated for all of the regions in which a contingency table was constructed.Aﬂow associated probability of ‡95%was calculated inﬁve of the six subjects in the ascending colon and four of four subjects in the transverse colon,indicating a genuine association between PSs and episodicﬂow across both the mid-ascending and mid-transverse colon(Table2). Episodic luminalﬂow and association with propagating sequencesIn total292discrete episodes of antegradeﬂow and175 discrete episodes of retrogradeﬂow were detected.In the ascending colon episodes of antegradeﬂow travers-ing the mid-ascending colon(226episodes;37.6±16.5 episodes/subject)were more prevalent than retrogradeTable1Total number of propagatingsequences(PS)recorded from the sixvolunteers and their temporal associationwithﬂow.Cecal PS are associated withﬂow across the mid-ascending colonicand transverse colonic PSs are associatedwithﬂow across the mid-transversecolonRegion Polarity Numberof PSsAssociatedwithﬂowAscending PS PSs Antegrade9689(93%)Retrograde1514(93%)Low amplitude PSs Antegrade88(100%)Retrograde32(67%)Transverse PS PSs Antegrade4138(93%)Retrograde99(100%)Low amplitude PSs Antegrade43(75%)Retrograde20P.G.Dinning et al.Neurogastroenterology and MotilityÓ2008The AuthorsJournal compilationÓ2008Blackwell Publishing Ltd516ﬂow (127episodes;21.2±7.3episodes/subject)(P <0.0001).In the transverse colon antegrade ﬂow(66episodes;16.5±5.2episodes/subject)and retro-grade ﬂow (48episodes;12.0±5.1episodes/subject)were equally prevalent.For the 273min in which isotope was present inboth the ascending and transverse colon,nearly half(43.5%)of antegrade ﬂow episodes that crossed themid-ascending colon continued beyond the mid-trans-verse colon.Indeed the majority (66%)of ﬂow episodesthat crossed the mid-transverse colon originated in theproximal ascending colon.The remaining episodes of ﬂow originating in the proximal ascending colon terminated at the hepatic ﬂexure.Of the 292episodes of antegrade ﬂow,45%were temporally associated with PSs.The degree of associ-ation between ﬂow and PSs varied considerably amongst colonic regions [ascending colon 41.6%vs 60.6%transverse colon;(P =0.006)]and amongst sub-jects (range:8%to 91%).Of the 175episodes of retrograde ﬂow only 10.8%were associated with retrograde PSs.Just under half (49%)of theretrograde Figure 3Temporal association between propagating sequences and ﬂow in the proximal colon.The hatched arrows link theindividual scintiscan image corresponding to the time along the horizontal axis at which acquisition of the frame was completed.Each solid arrow indicates the location of the manometric sidehole from which the corresponding pressure trace was recorded.The solid bars indicate episodes of ﬂow across the mid-ascending colon and mid-transverse colon.The ﬁrst cecal propagating sequence (PS1)contains individual propagating pressure waves with a trough to peak amplitude <5mmHg.The corresponding scintigraphic images clearly show content moving from the cecum to the hepatic ﬂexure.The second propagating sequence (PS2)originates in the cecum and propagates throughout the proximal colon.It was temporally linked with ﬂow across both the mid-ascending and mid-transversecolon.Figure 4An example of a retrograde propagating sequence temporarily associated with retrograde ﬂow across the mid-transverse colon.The hatched arrows link the individual scintiscan image corresponding to the time along the horizontal axis at which acquisition of the frame was completed.Each solid arrows indicates the location of the manometric sidehole from which the corresponding pressure trace was recorded.The solid bar represents the ﬂow period across the mid-point of the transverse colon.Volume 20,Number 5,May 2008Determinants of colonic ﬂow Ó2008The AuthorsJournal compilation Ó2008Blackwell Publishing Ltd 517ﬂow episodes were recorded immediately following episodes of antegradeﬂow.DISCUSSIONThis study has shown that the vast majority of proximal colonic PSs,regardless of site of origin, amplitude or polarity,are temporally linked to luminal propulsion.These data are in contrast to the only previously study which has attempted to detail the relationship between proximal colonic PSs and luminal ﬂow.1In that study only30%of antegrade PSs and less than10%of retrograde PSs originating distal to hepatic ﬂexure were deemed propulsive.In addition that study showed a correlation between propulsion and the amplitude of PSs.The difference in the reported ﬁndings between that study and ours can be explained by our use of high resolution scintigraphy and our development of a quantitative technique for analyzing the data.20In human colonic studies of combined manometry and scintigraphy,motor patters are recorded in real time while the image capture is limited by the frame rate of the scintiscanning equipment.As the majority of colonic propulsive events are thought to involve subtle movements of colonic content over short distances,26–28the larger the time window between captured scintigraphic images the more likely is the chance of overlooking these subtle movements.The scintigraphic frame rates used in the majority of previous human studies,have ranged from successive 1min frames5,7,29to1min frames every5min.6Such frame rates would limit detailed correlation between motor patterns andﬂow.Even when the image capture rate is improved to1 frame every15s two-thirds of the identiﬁed PSs were not associated with any detectable movement of isotope.1The incomplete correlation described in that study might be explained by that studyÕs reliance upon visual identiﬁcation of individual isotopic movements. Such analysis can be problematic because it is difﬁcult to pick up discreet movements over short distances, especially if image quality is poor.Visual analysis is more likely to detect high volume movements over large distances.To overcome problems associated with visual detec-tion ofﬂow we designed a quantitative technique to identify discrete,episodic movements of colonic con-tent on a moment-to-moment basis.20This quantita-tive technique increased our ability to detectﬂow when compared to visual analysis and therefore allowed us for theﬁrst time to examine in detail the propulsive nature of all PSs,including those with low amplitude pressure ing this technique we have shown that PSs are a major determinant of luminalﬂow.Thisﬁnding allows us to readdress previously conceived hypotheses regarding the physi-ology of the colon.For example Cook et al.1had described the relative propulsiveÔinefﬁciencyÕof ante-grade PS which was hypothesized to help retard antegrade transit,thus allowing time for optimal water and electrolyte absorption and reduce the challenge to continence.As our data indicates that PSs are propulsive,other factors are likely play a role in slowing antegrade transit.We propose that these factors include;(i)short extent PSs with component pressure waves between 2mmHg and40mmHg,(previously shown to make up the vast majority of PSs recorded in the human the proximal colon9,30);(ii)retrograde PSs;and(iii)the reﬂux of50%of the aboral proximal colonic bolus movement.Together,these factors are likely to con-tribute to the subtleÔto and froÕmotion helping to maintain maximal absorption and retard transit and stool frequency.Despite the propulsive properties of PSs,just over half of the identiﬁedﬂow episodes occurred in the absence of any identiﬁable PSs.While other factors are also attributed to propulsion of colonic content such as the tone of colon wall4,31and the viscosity of faecal material,27it is also likely that colonic manometry, will fail to detect a proportion of the colonic activity. Manometry is less sensitive when the diameter of the gut exceeds5.6cm14and as the cecum is the most capacious of the proximal colonic regions,it may explain why the degree of association betweenﬂow and PSs was signiﬁcantly reduced in the ascending compared to the transverse colon.In addition it has been reported that manometric pressure patterns can extend over relatively limitedTable2In each patient for each region the X2relation was greater than the X1relation allowing for calculation of the ﬂow associated probability.A value<95%indicates that any association between propagating sequences andﬂow could occur by chance.A value>95%indicates that there is a gen-uine association between propagating sequences andﬂowSubject Ascendingcolon(%)Transversecolon(%)196NA299.7NA39399.7499.9999.99599.9999.7699.9999.3P.G.Dinning et al.Neurogastroenterology and MotilityÓ2008The AuthorsJournal compilationÓ2008Blackwell Publishing Ltd518。

High-gain, polarization-preserving, Yb-doped fiber

High-gain,polarization-preserving,Yb-dopedﬁberampliﬁer for low-duty-cycle pulse ampliﬁcationJ.R.Marciante and J.D.ZuegelAmpliﬁed spontaneous emission(ASE)suppression techniques were utilized to fabricate a double-pass,Yb-doped ampliﬁer with the noise properties of a single-pass ampliﬁer.Simulations based on a rateequation model were used to analyze the ASE and the effectiveness of the suppression techniques.Thesetechniques were implemented in an alignment-free,double-pass,Yb-dopedﬁber ampliﬁer with a26dBgain at a wavelength23nm off the gain peak and aϪ48dB noiseﬂoor while amplifying linearly polarizedoptical pulses with a low-duty cycle.©2006Optical Society of AmericaOCIS codes:060.2320,140.3280,060.2340,060.2420,140.3510,140.4480.1.IntroductionEr-dopedﬁber ampliﬁers have become commonplace in telecommunications systems.1High-bit-rate pulse trains mean that continuous pumping can be utilized. Additionally,the unknown polarization state of the arriving pulses is ideal for commonﬁber-optic com-ponents and Er-dopedﬁbers,which typically do not preserve the polarization state of the light passing through them.For other applications,however,such conditions do not apply.Low-duty-cycle pulses leave gain available in theﬁber for long durations between pulses,which can lead to parasitic lasing or destructive self-pulsations.The ampliﬁcation of signals with wave-lengths far from the gain peak only enhances this prob-lem,since the gain can be substantially lower for such signals.Additionally,the ampliﬁcation of linear polar-izations requires not only a polarization-maintaining (PM)activeﬁber but also PM wavelength-division multiplexers(WDMs)and otherﬁber components that can be difﬁcult to fabricate.Utilizing a double-pass conﬁguration allows for sig-niﬁcantly higher gains to be obtained in a single-ﬁber ampliﬁer than can be achieved in a single pass.2–6 Additionally,using a Faraday rotator just before the end mirror ensures that the pulse returning from the second pass through the cavity has a polarization state that is orthogonal to that of the input pulse. Thus when used in conjunction with a polarizing beam splitter,the output pulse can be separated from the input pulse with highﬁdelity.3Double-pass con-ﬁgurations are ripe for parasitic lasing or destructive self-pulsations in a highly pumped unsaturated am-pliﬁer,however,since half a resonator is created in-tentionally.Special care must be taken to minimize reﬂections from components,connectors,and splices. Utilizing a timed gate,e.g.,an acousto-optic modula-tor(AOM),at the end of theﬁrst pass can ensure stable operation3but signiﬁcantly adds to the com-plexity of the system.In this work,a double-pass,Yb-dopedﬁber ampli-ﬁer is presented that overcomes these hurdles to pro-vide high gain for low-duty-cycle pulse repetition rates while preserving the linear polarization state in a single-spatial-mode package that requires no align-ment.In Section2,the ampliﬁed spontaneous emis-sion(ASE)is modeled in Yb-dopedﬁber ampliﬁers.In particular,the effects of ASEﬁltering are studied for use in a double-pass ampliﬁer conﬁguration.In Sec-tion3,results are presented on the measurements of a double-passﬁber ampliﬁer built with the various ASE-suppression schemes described in Section2.Ad-ditional discussions regarding the experimental con-ﬁguration,modeling,noise,and nonlinear effects are given in Section4,along with concluding remarks.2.Ampliﬁed Spontaneous Emission Considerations The ASE ofﬁber ampliﬁers can be studied via rate-equation modeling.7Such a model represents the op-tical power resolved in wavelength along the length of theﬁber and the Yb atomic states as a homoge-The authors are with the Laboratory for Laser Energetics, University of Rochester,250East River Road,Rochester,New York 14623-1299.J.Marciante’s e-mail address is johnm@lle.rochester. edu.Received19January2006;revised21April2006;accepted24 April2006;posted24April2006(Doc.ID67379).0003-6935/06/266798-07$15.00/0©2006Optical Society of America6798APPLIED OPTICS͞Vol.45,No.26͞10September2006neously broadened inversion.The resultant equa-tions are given byϮѨPϮѨzϩ1v gѨPϮѨtϭ⌫͓␴e N2Ϫ␴a N1͔PϮϪ␣PϮϩ2␴e N2hc␭3⌬␭ϩS␣RS Pϯ,(1)ѨN2Ѩtϭ1͵⌫͓␴e N2Ϫ␴a N1͔ϫ͑PϩϩPϪ͒d␭ϪN2␶,(2)where PϮ͑z,t,␭͒is the forward͑ϩ͒or backward͑Ϫ͒propagating power as a function of wavelength,time, and axial position along theﬁber.N2and N1are the upper and lower state population densities,respec-tively,as a function of time and axial position along the ﬁber and are related by the total ion concentration as N tϭN2ϩN1,which is constant throughout theﬁber. Wavelength-dependent parameters in Eqs.(1)and (2)include the geometrical overlap of theﬁber mode with the core⌫,the modal area A,the absorption͞emission cross section of the active ion␴a͞e,the group velocity v g,theﬁber attenuation␣,and the Rayleigh scattering coefﬁcient␣RS.Additional parameters in-clude the upper state lifetime␶,theﬁber-core capture coefﬁcient S,and the optical sampling bandwidth⌬␭. Equation(1)represents the bidirectional power ﬂow through theﬁber including stimulated emission, spontaneous emission,and absorption from the ac-tive ions;loss due to the inherentﬁber attenuation; and Rayleigh scattering.Since the parameters all have wavelength dependence,only a single equation is mathematically required to represent the behavior of the pump,the signal,and the ASE.Equation(2)represents the excited-state popula-tion density,which is governed by the absorption and emission of optical power as well as nonradiative decay.One notable omission in Eq.(2)is the wave-length dependence of the excited state governed by the details of the atomic transition manifold.This leads to,for example,excitation due to the absorption of long-wavelength light that can then be used to amplify shorter-wavelength light.While this effect issmall in the presence of a highly invertedﬁber,theimpact on the current work is to overestimate theASE at the gain peak.Since the current goal is tosuppress this feature,the model will yield a worsecase than is expected experimentally.For simplicity,the wavelength dependences ofA,v g,␣,and␣RS are neglected.Since the currentwork considers core-pumped(as opposed to cladding-pumped)activeﬁbers,⌫can also be well approxi-mated by a constant.The Yb cross sections areobtained from the data in Ref.8.For convenience,these cross sections areﬁt to a series of Gaussians of the form␴jϭ͚m A j,m exp͕Ϫ͓͑␭Ϫ␭j,m͒͞w j,m͔2͖with the coefﬁcients listed in Table1.Other parametersutilized are listed in Table2.Data provided with the Yb-dopedﬁber(Nufern),along with our own measurements on theﬁber,de-termined the values of N t,⌫,and A.The other valueswere obtained from Ref.7.The parameters usedare compiled in Table2.A simpleﬁnite-differencemethod is utilized to calculate the power and inver-sion distributions in theﬁber.Initial conditions as-sume no optical power or inversion within theﬁber.The pump power is included as a boundary condition.A double-pass conﬁguration can also be realized byapplying the appropriate boundary conditions.Unsaturatedﬁber ampliﬁers cannot be pumped toarbitrarily high levels because of self-pulsations andoscillation,which limit the length ofﬁber that can bepractically used in a single-pass ampliﬁer.In an un-saturated ampliﬁer,this translates to limited avail-able gain.Figure1shows the forward and reverseampliﬁed spontaneous emission for the case of re-verse pumping and very weak signal ampliﬁcation(no gain depletion)for a3.5m length of Yb:ﬁber withthe characteristics in Table2.Also shown in thisﬁgure are the small-signal gain,deﬁned as the ratioof output energy to input energy,for a1053nm signaland the pump leakage.After140mW of pump power,theﬁber is almost completely inverted,and the re-maining pump is lost out of the opposite end of theﬁber.The small-signal gain of23nm off the gain peakis therefore limited to approximately20dB.Since the gain is in fact unsaturated,simply sendingthe signal back through for a second pass increasesthe ampliﬁcation without additional pumping.Such aTable1.Gaussian Coefﬁcients for Yb Emission and AbsorptionCross SectionsjA j,m(10Ϫ27m2)␭j,m(nm)wj,m(nm)a18095070 a36089524 a51091822 a16097112 a,e23259754 e16097812 e340102520 e175105060 e150103090Table2.Parameters Used in SimulationsParameter ValueNt9.4ϫ1024mϪ3⌫0.85A30␮m2vgc͞1.5␣0.003mϪ1S␣RS1.2ϫ10Ϫ7mϪ1␶0.84ms⌬␭1nm␭pump976nm␭signal1053nm10September2006͞Vol.45,No.26͞APPLIED OPTICS6799double-pass conﬁguration,however,also allows the ASE to make a second trip through the gain,which can lead to undesirable oscillation and self-pulsations. Therefore the ASE must beﬁltered to use a double-pass conﬁguration.Two simple methods ofﬁltration are investigated,as depicted in Fig.2.Theﬁrst is a bandpassﬁlter,which is inserted between subsequent trips through the activeﬁber to deny double-pass ASE except in a small bandwidth around theﬁlter peak. The second is a WDM designed to split the ASE peak from the signal,which removes a signiﬁcant fraction of the ASE power from the signal,provided the signal is not near the gain peak.Four different ampliﬁer con-ﬁgurations are modeled,as shown in Fig.2.Theﬁrst two are simple single-and double-pass conﬁgurations through theﬁber.In the third conﬁguration,a band-passﬁlter is added between subsequent passes through theﬁber.The fourth conﬁguration adds a WDMﬁlter to the output of the third conﬁguration. For simplicity,theﬁlters are assumed to be lossless at the transmission peaks with zero transmission at the nulls.The mirror adds1dB of loss to the double-pass ampliﬁers.Assuming pump and signal wave-lengths of976and1053nm,respectively,the total (spectrally integrated)ASE power out of the different ampliﬁer conﬁgurations is shown in Fig.3as a func-tion of pump power.Because of undepleted gain and pump leakage,the single-pass ASE has a maximum power below6mW.In the double-pass conﬁguration, the ampliﬁer becomes saturated by the ASE,which extracts a signiﬁcant fraction of the gain in theﬁber. This linear trend in the ASE growth with pump powerto an ASE level that is greater than25times thecase and means that there will be veryreduced gain available for the signal.In con-the dotted trace shows that the insertion of thebandpassﬁlter prohibits the ASE closest to the gainpeak from experiencing double-pass gain.The totalASE is therefore limited to small-signal ampliﬁcation,even with double-pass ampliﬁcation far off the gainpeak,and accumulates no more than twice the powerof the single-pass conﬁguration.The insertion of thisﬁlter then allows for exponential signal gain from thedouble-pass ampliﬁer.The addition of the WDMﬁlter(solid curve)reduces the total ASE output power ofthe double-passﬁber ampliﬁer to only30%more thanthe single-pass ASE.The ASE spectra for the four conﬁgurations areshown in Fig.4at a pump power of250mW.The ASE Fig.1.Total(spectrally integrated)forward(solid curve)andreverse(dashed curve)ASE,976nm pump leakage,and1053nmsmall-signal gain as a function of pump power for a3.5m length ofYb:ﬁber.Fig.2.Depiction of modeled conﬁgurations:(a)single pass,(b)double pass,(c)double pass with an intracavity bandpassﬁlter,and(d)same as(c)with a WDMﬁlter at the ampliﬁer output.Fig.3.Total(spectrally integrated)ASE power at the ampliﬁeroutput as a function of pump power for the conﬁgurations shown inFig.2using theﬁber from Fig.1.6800APPLIED OPTICS͞Vol.45,No.26͞10September2006within the bandpass ﬁlter is actually stronger thanunﬁltered double-pass ASE since the gain is un-Nonetheless,this lack of gain saturation for double-pass gain of the signal in the am-The bandpass ﬁlter is therefore critical for the operation of this double-pass conﬁguration for the ampliﬁcation of signals off the gain peak.3.Experimental ResultsUnder the guidance of Section 2,a double-pass polar-ized ampliﬁer was constructed containing both the bandpass ﬁlter and the WDM,utilizing a Faraday mirror with a polarizing beam splitter (PBS)to sep-arate input from output and preserve the linear polarization state.This ampliﬁer conﬁguration is shown in Fig.5and utilizes both single-mode (SM)and PM ﬁbers.The input signal comes through a PM-pigtailed isolator (Optics for Research)to prevent any ASE or signal from returning to the seed source.The light then passes through a PM-pigtailed polariz-ing beam splitter (OZ Optics)followed by a SM 1030nm ͞1053nm WDM (ITF)used to reduce ASE.A SM WDM (Lightel)combines the seed light with pump light from a ﬁber-Bragg-grating-stabilized pump laser (JDSU)with a second WDM in series for additional isolation of the pump diode from the am-pliﬁed signal.The combined light is sent into 3.6m of SM,single-clad,Yb-doped ﬁber (Nufern)with an un-saturated absorption coefﬁcient of approximately 70dB ͞m at 975nm.After the ﬁrst pass of ampliﬁca-tion,the signal passes through a Faraday mirror (Op-tics for Research),a factory-aligned,ﬁber-coupled package containing a Faraday rotator,a 10nm band-pass ﬁlter at 1053nm (Andover),and a mirror.The reﬂected light passes back through the Faraday mir-ror package,the Yb-doped ﬁber,and the WDMs,one of which acts to ﬁlter out the ASE centered at 1030nm (WDM1).Since the polarization at the PBS is orthogonal to that which entered the PBS because of the Faraday mirror,the light is ejected out of a different port and sent through a PM 90͞10splitter (Lightel)to provide polarized signal and sample ports.All ﬁbers were spliced using a Furukawa S183PM fusion splicer,and there are no alignment knobs in the system.The pump was operated contin-uously,and there is no AOM gate,so no temporal alignment is required between the seed pulses and the ampliﬁer.The seed pulse used in this ampliﬁer was a 2ns square pulse with 1.56pJ of energy,resulting in a peak power of 0.78mW.The seed pulse had an opti-cal wavelength of 1053nm and a pulse repetition rate of 300Hz and was linear polarized with a polariza-tion extinction ratio of 20dB.Different lengths of Yb-doped ﬁber were tested in the ampliﬁer to opti-mize the active ﬁber length in terms of maximizing gain while maintaining stability as well as operation free from parasitic lasing or self-pulsations.For a given ﬁber length,a fraction of the ﬁber remains unpumped,and thus lossy to the signal wavelength.If the pumped portion of the ﬁber has sufﬁcient gain,then the ampliﬁer can Q switch because of the satu-rable absorption of the unpumped section of ﬁber andFig.4.Ampliﬁed spontaneous emission spectra at the ampliﬁer output for the conﬁgurations shown in Fig.2using the ﬁber from Fig.1with 250mW of pump power.Fig.5.Schematic of a high-gain double-pass ampliﬁer consisting of input and output isolators,a PBS,an ASE–signal wavelength division multiplexer (WDM1),two pump–signal wavelength division multiplexers (WDM2),3.6m of Yb-doped ﬁber,a Faraday mirror,and a 90͞10splitter.The Faraday mirror was a factory-aligned package containing a lens (L),a Faraday rotator (FR),a bandpass ﬁlter (F),and a mirror (M).The PM ﬁber is notated by dotted curves,while the SM ﬁber is notated by solid curves.10September 2006͞Vol.45,No.26͞APPLIED OPTICS6801minute reﬂections in the system.This can be reme-died by making the ﬁber sufﬁciently short;however,there is an optimum ﬁber length for which the ﬁber provides maximum gain without self-pulsations.For several ampliﬁers that were constructed,the opti-mum ﬁber length was determined to be near 3.5m,regardless of the variability between components or splice quality.The pump utilized had a threshold of 16mA and a slope efﬁciency of 0.68mW ͞mA.Figure 6shows the signal gain,output energy,and total ASE power as a function of pump current.As the pump current is increased beyond 200mA,the am-pliﬁer gain rolls off since the additional pump light is mostly not absorbed in the active ﬁber,a feature that is also reﬂected in the ASE curve.It is important to note that because of the low seed energy and repetition rate,the ampliﬁer is unsaturated.Thus the small-signal gain afforded by this ampliﬁer 23nm off the gain peak is nearly 27dB.Self-pulsations were not observed at any pump cur-rents because of the bandpass ﬁlter in the Faraday mirror assembly and the 1030nm ͞1053nm WDM that suppresses the stronger gain at shorter wave-lengths.Figure 7shows the ASE spectra of the double-pass ampliﬁer for various pump current levels.The seed wavelength is depicted by the vertical dashed curve.The traces clearly show the double-pass gain of the wavelengths in the ﬁlter passband compared to the single-pass gain of those outside the passband.The top trace in Fig.7,which is offset for clarity,is the ASE spectra for the double-pass ampliﬁer without the 1030nm ͞1053nm WDM.This WDM,combined with the 10nm bandpass ﬁlter,has a signiﬁcant impact both on the output of the ampliﬁer as well as the stability against lasing and self-pulsations,as evi-denced in Fig.7,and shows results very similar to the modeling results shown in Fig.4.It is difﬁcult to deﬁne the noise ﬂoor of a system with regard to a low-repetition-rate signal.Simply comparing the strength of the signals on an optical spectrum analyzer requires an optical gate such that the spectrum is only integrated for the duration of the pulse.One alternative method compares the peak power of the ampliﬁed signal to the ASE power.Amore meaningful metric that more accurately repre-sents an optical signal-to-noise ﬁgure of merit com-pares the peak power of the ampliﬁed pulse to the ASE power in a limited spectral bandwidth around the seed-pulse wavelength.While the total noise ﬂoor is less than Ϫ28dB across the entire operating range,the bandwidth-limited noise ﬂoor is better than Ϫ48dB in a 0.1nm bandwidth around the signal wavelength.Since the ampliﬁer runs unsaturated,the noise ﬂoor is essentially constant as a function of the pump current.The combination of a Faraday mirror and PBS al-lows for high-ﬁdelity separation of the input and out-put signals at the front end of the ampliﬁer while maintaining the linear polarization state of the seed.The polarization extinction of the ampliﬁed signal was measured to be 19.9dB,which is identical to that of the input signal.4.Discussion and ConclusionsThe pump is a grating-stabilized diode centered at 975.5nm.The feedback from the ﬁber Bragg grating maintains stable laser operation even in the presence of optical feedback,which can otherwise destabilize a diode laser.9However,the ampliﬁed seed can be a problem in damaging the diode.Starting with a 2pJ seed signal of 2ns duration,the ampliﬁed signal be-comes ϳ1nJ.By some spurious reﬂection,a second round trip through the double-pass ampliﬁer is pos-sible,leading to 0.5␮J.Even in the presence of a pump–signal WDM,the fraction split off from this energetic pulse ͑ϳ15dB ͒leads to a pulse impinging on the face of the pump diode with an energy of 15nJ and a peak power over 7W.This can cause cata-strophic optical damage on the facet of the laser di-ode.Two components in our system serve to eliminate this problem.The ﬁrst is the output isolator,which signiﬁcantly reduces the risk of a high-energy back-reﬂection into the ampliﬁer.The second is thesecondFig.7.Ampliﬁed spontaneous emission spectra from a dual-pass ﬁber ampliﬁer for various pumping levels.Also shown are the seed wavelength at 1053nm and the ASE trace for the ampliﬁer with-out the 1030nm ͞1053nm WDM,which has been offset forclarity.Fig.6.Gain,output energy,and ASE power of the experimental ampliﬁer described in Fig.5as a function of pump current.6802APPLIED OPTICS ͞Vol.45,No.26͞10September 2006pump–signal WDM,which serves as anϳ15dB iso-lator of the signal pulse into the diode.The agreement between measurements and simu-lations is favorable in the prediction of the noise char-acteristics of the ampliﬁer,as shown in Table3.In particular,using the ratio of small-signal gain over ASE power leads to an ampliﬁer performance metric of1161͞mW,which agrees exceedingly well with the measurements.Both the signal gain and the ASE power levels were approximately6dB too high in the simulations,even when accounting for realistic component loss.There are several reasonable expla-nations for the discrepancy.First,there is the uncer-tainty in the loss owing to the components and the splices.Second,the addition of the10nmﬁlter in the Faraday mirror assembly may cause additional in-sertion loss due to slight perturbations in propaga-tion length and͞or angle between the mirror and the ﬁber within the assembly.Finally,some of the parameters used in the simulations were simply taken from previous work.7,8In particular,the emis-sion and absorption cross sections can play a large role in determining the output performance of the ampliﬁer.By varying the simulated emission cross section at1053nm,it is found that the ASE and gain change by9.5%and8.1%,respectively,for a1% change in the emission cross section alone.Given the variability between measurements of Yb absorption and emission cross sections,8,10this is likely a strong contributor to the mismatch in absolute values of gain and ASE power.The unpumped ampliﬁer has a passive loss of approximately15dB.The simulations show an un-pumped ampliﬁer loss of13.9dB.While some of this loss is due to absorption in the unpumped Yb-doped ﬁber,almost9dB of this loss is insertion loss of the constituent components.Many of these components are free-space optics that are packaged withﬁber pigtails at the vendor.All-ﬁber components would certainly help to increase the gain of the system,as well as the noiseﬁgure.The noiseﬁgure of the am-pliﬁer is given by11NFϭ2P ASEh␯⌬␯opt G,(3)where P ASE is the ASE power measured on a band-width⌬␯opt,G is the signal gain,and␯is the optical frequency of the ing measured parameters, the noiseﬁgure of the double-pass ampliﬁer in a 0.1nm bandwidth is6.6dB.Considering that the seed is degraded by2.8dB because of the insertion loss of the components before the activeﬁber,this result leads us to conclude that,in spite of operating far from the gain peak in a double-pass conﬁguration, the ampliﬁer is of extremely high quality because of the ASE suppression techniques utilized.Further, our model shows that our double-pass ampliﬁer does not add any penalty to the noiseﬁgure,as has been observed in Er-doped ampliﬁers used in a simple double-pass conﬁguration.4Multiple path interference(MPI)can also lead to a degradation of the signal-to-noise ratio inﬁber ampliﬁers.12–14In conventional single-pass ampliﬁ-ers,Rayleigh scattering can reﬂect a portion of the signal backward.A second Rayleigh event can sub-sequently reﬂect that portion back into the signal path.This effect is particularly important when the scattering occurs at locations that allow the scattered photons to see additional gain,as compared to the signal gain in the ampliﬁer.For double-pass ampli-ﬁers,the more signiﬁcant contribution comes from a single Rayleigh scattering event that occurs in the return trip through the ampliﬁer.This backscattered light will see additional gain and a deterministic sec-ond reﬂection(via the mirror)back through the am-pliﬁer into the signal path.The strength of the double Rayleigh scattering(DRS)compared to the signal is generally given by15P DRS͑L͒S͑͒ϭ͵0L d z2͵0z2d z1͑S␣RS͒2Gជ͑z1,z2͒Gឈ͑z1,z2͒,(4)where the G terms represent the gain in the for-ward and reverse directions.In applying this for-malism to our double-pass case,one of the Rayleigh scattering terms,S␣RS,is replaced by a determinis-tic reﬂection from the back end of the double-pass ampliﬁer,R2␦͑z2ϪL͞2͒,where R2is the power re-ﬂection coefﬁcient of the ampliﬁer end mirror.The highest gain case is where the signal has completed its trip through the gainﬁber and the scattered light then sees a complete additional double pass through the gainﬁber.This can be formally written for our unsaturated ampliﬁer asP DRS͑L͒P S͑L͒ϭR2S␣RS͵L͞2Ld z2e g͑z2ϪL͞2͒ϭR2S␣RS2g͑G DPϪͱG DP͒,(5)where G DP is the double-pass ampliﬁer gain.For the ampliﬁer described in Section3,the resultant rela-tive noise strength due to MPI isϪ47dB.It is important to note,however,that for low-duty-cycle pulse-train ampliﬁcation,this MPI is temporally sep-arable from the desired signal pulse and can there-fore be eliminated by temporal gating.Consideringparison of Measured and Simulated Values for theDouble-Pass Fiber AmpliﬁerCharacteristic Measured Value Simulation ResultSmall-signal gain͞ASE1175͞mW1161͞mWTotal noiseﬂoorϪ28dBϪ30.6dBNoiseﬂoor in0.1nmbandwidthϪ48dBϪ43.5dBSmall-signal gain26.6dB32.3dB10September2006͞Vol.45,No.26͞APPLIED OPTICS6803only MPI noise that contributes during the signal implies signiﬁcantly shorter gain paths for the MPI photons.This effect will,therefore,be more pro-nounced at the trailing edge of the pulse.The gain length of the backscattered light is conﬁned to half the pulse duration.This is simply represented in Eq.(5)by substituting the gainﬁber length L with␶v g͞2, where␶is the pulse duration.For a2ns pulse,the relative noise is,therefore,Ϫ80dB on the back end of the pulse.From these calculations,it is evident that MPI will not present a signiﬁcant impairment,par-ticularly if temporalﬁltering is utilized.Finally,nonlinear effects in this ampliﬁer need to be considered to understand the scaling to high output energy.For narrowband signals,stimulated Brillouin scattering(SBS)is generally the dominant limitation.In the application of low-duty-cycle pulse trains where only one pulse exists in the ampliﬁer at any given time,however,the Brillouin gain only ex-ists during the pulse.Since SBS produces light that propagates backward with respect to the signal,the effectiveﬁber length is limited to␶v g͞2.The SBS gain threshold16for the6␮m coreﬁber is approximately 20W m,which leads to a peak power limitation of 100W or100nJ per nanosecond of pulse duration. Stimulated Raman scattering(SRS)has a higher threshold,but is not limited to the pulse duration since the photons are scattered in the forward direc-tion.For the case of exponential gain where the Raman pump changes during propagation,an effec-tive length of the SRS effect can be calculated,similar to what is done for pump loss in conventional Raman ampliﬁers.16For the ampliﬁer described in Fig.5, however,theﬁber length of interest is the span from the output end of the Yb-dopedﬁber to the output of the ampliﬁer(approximately3m),during which the signal strength(Raman pump)is at its highest level and is effectively constant.Given that the SRS thresh-old16is approximately500W m,this leads to a limi-tation of166W of peak power,or166nJ͞ns of pulse duration.For2ns pulses,energies below1nJ were reported in Section3,indicating that nonlinear ef-fects were not important for the performance of the ampliﬁer.However,scaling the pulse energy by a factor of100is possible before running into nonlinear limitations in this ampliﬁer.In conclusion,ampliﬁed spontaneous emission sup-pression techniques were utilized to fabricate a double-pass,Yb-doped ampliﬁer with the noise properties of a single-pass ampliﬁer.Simulations based on a rate-equation model were used to analyze the ASE and the impact of the suppression techniques.These tech-niques were implemented in an alignment-free,double-pass,Yb-dopedﬁber ampliﬁer with a26dB gain at a wavelength23nm off the gain peak and aϪ48dB noiseﬂoor while amplifying linearly polarized optical pulses with a low-duty cycle.The authors thank Jake Bromage for technical discussions.This work was supported by the U.S. Department of Energy(DOE)Ofﬁce of Inertial Con-ﬁnement Fusion under Cooperative Agreement DE-FC52-92SF19460,the University of Rochester,and the New York State Energy Research and Develop-ment Authority.The support of the DOE does not constitute an endorsement by the DOE of the views expressed in this paper.References1.E.Desurvire,Erbium-Doped Fiber Ampliﬁers:Principles andApplications(Wiley,1994).2.S.Hwang,K.W.Song,H.J.Kwon,J.Koh,Y.J.Oh,and K.Cho,“Broad-band erbium-dopedﬁber ampliﬁer with double-pass conﬁguration,”IEEE Photon.Technol.Lett.13,1289–1291(2001).3.A.Galvanauskas,G.C.Cho,A.Hariharan,M.E.Fermann,and D.Harter,“Generation of high-energy femtosecond pulses in multimode-core Yb-ﬁber chirped-pulse ampliﬁcation sys-tems,”Opt.Lett.26,935–937(2001).4.S.W.Harun,P.Poopalan,and H.Ahmad,“Gain enhancementin L-band EDFA through a double-pass technique,”IEEE Photon.Technol.Lett.14,296–297(2002).5.M.Tang,Y.D.Gong,and P.Shum,“Dynamic properties ofdouble-pass discrete Raman ampliﬁer with FBG-based all-optical gain clamping techniques,”IEEE Photon.Technol.Lett.16,768–770(2004).6.L.L.Yi,L.Zhan,J.H.Ji,Q.H.Ye,and Y.X.Xia,“Improve-ment of gain and noiseﬁgure in double-pass L-band EDFA by incorporating aﬁber Bragg grating,”IEEE Photon.Technol.Lett.16,1005–1007(2004).7.Y.Wang and H.Po,“Dynamic characteristics of double-cladﬁber ampliﬁers for high-power pulse ampliﬁcation,”J.Light-wave Technol.21,2262–2270(2003).8.R.Paschotta,J.Nilsson,A.C.Tropper,and D.C.Hanna,“Ytterbium-dopedﬁber ampliﬁers,”IEEE J.Quantum Elec-tron.33,1049–1056(1997).ach and A.R.Chraplyvy,“Regimes of feedback effectsin1.5-␮m distributed feedback lasers,”J.Lightwave Technol.LT-4,1655–1661(1986).10.N.A.Brilliant,R.J.Beach,A.D.Drobshoff,and S.A.Payne,“Narrow-line ytterbiumﬁber master-oscillator power ampli-ﬁer,”J.Opt.Soc.Am.B19,981–991(2002).11.P.C.Becker,N.A.Olsson,and J.R.Simpson,Erbium-DopedFiber Ampliﬁers:Fundamentals and Technology(Academic, 1999).12.S.Faralli and F.Di Pasquale,“Impact of double Rayleighscattering noise in distributed higher order Raman pumping schemes,”IEEE Photon.Technol.Lett.15,804–806(2003).13.S.W.Harun,N.Tamchek,P.Poopalan,and H.Ahmad,“Double-pass L-band EDFA with enhanced noiseﬁgure char-acteristics,”IEEE Photon.Technol.Lett.15,1055–1057 (2003).14.M.Tang,P.Shum,and Y.D.Gong,“Design of double-passdiscrete Raman ampliﬁer and the impairments induced by Rayleigh backscattering,”Opt.Express11,1887–1893(2003).15.J.Bromage,P.J.Winzer,and R.-J.Essiambre,“Multiple pathinterference and its impact on system design,”in Raman Am-pliﬁers for Telecommunications:2,Subsystems and Systems, M.N.Islam,ed.,Springer Series in Optical Science(Springer, 2004),Chap.15.16.G.P.Agrawal,Nonlinear Fiber Optics,2nd ed.,Optics andPhotonics Series(Academic,1995).6804APPLIED OPTICS͞Vol.45,No.26͞10September2006。

SY88149CL中文资料

SY88149CL3.3V, 1.25Gbps PECL Limiting PostAmplifier w/High Gain TTL Signal DetectGeneral DescriptionThe SY88149CL is a high-sensitivity limiting postamplifier designed for use in fiber-optic receivers. Thesedevices connect to typical transimpedance amplifiers(TIAs). The linear signal output from TIAs can containsignificant amounts of noise and may vary in amplitudeover time. The SY88149CL quantizes these signals andoutputs PECL level waveforms.The SY88149CL operates from a single +3.3V powersupply, over temperatures ranging from –40o C to +85o C.With its wide bandwidth and high gain, signals with datarates up to 1.25Gbps, and as small as 5mV pp, can beamplified to drive devices with PECL inputs.The SY88149CL generates a high-gain signal-detect(SD) open-collector TTL output. The SD function has ahigh gain input stage for increased sensitivity. Aprogrammable Signal Detect level set pin (SD LVL) setsthe sensitivity of the input amplitude detection. SDasserts high if the input amplitude rises above thethreshold set by SD LVL and de-asserts low otherwise.The enable input (EN) de-asserts the true output signalwithout removing the input signal. The SD output can befed back to the EN input to maintain output stabilityunder a loss-of-signal condition. Typically, 3.4dB SDhysteresis is provided to prevent chattering.All support documentation can be found on Micrel’s website at: .Features•Single 3.3V power supply•Fast SD enable/disable time•622Mbps to 1.25Gbps operation•Low-noise PECL data outputs• High-gain SD•Chatter-free Open-Collector TTL signal detect (SD)output with internal 4.75kΩ pull-up resistor•TTL EN input•Programmable SD level set (SD LVL)•Available in a tiny 10-pin MSOP packageApplications• GE-PON/GPON/EPON• Gigabit Ethernet• 1062Mbps Fibre Channel• OC-12/24 SONET/SDH•High-gain line driver and line receiver•Low-gain TIA interfaceMarkets• FTTH/FTTP• Datacom/Telecom• Optical transceiverTypical Application CircuitOrdering Information(1)Part Number PackageType OperatingRangePackage MarkingLead FinishSY88149CLKG K10-1 IndustrialSY88149CLwithPb-Free bar line indicator NiPdAu Pb-Free SY88149CLKGTR(1)K10-1 Industrial SY88149CL with Pb-Free bar line indicator NiPdAu Pb-Free Notes:1. Tape and Reel.Pin Configuration10-Pin MSOP (K10-1)Pin DescriptionPin Number Pin Name Type Pin Function1 ENTTL Input: Default isHIGH. Enable: This input enables the outputs when it is HIGH. Note that this input is internally connected to a 25kΩ pull-up resistor and will default to a logic HIGH state if left open.2 DIN Data Input True data input.3 /DIN Data Input Complementary data input.4 VREF Reference voltage: Placing a capacitor here to V CC helpsstabilize SD LVL.5 SDLVL Input Signal Detect Level Set: a resistor from this pin to V CC sets thethreshold for the data input amplitude at which SD will beasserted.6 GND GroundDeviceground.7 SDOpen-collector TTLoutput w/internal 4.75kΩpull-up resistor Signal-Detect asserts high when the data input amplitude rises above the threshold set by SD LVL.8 /DOUT PECL Output Complementary data output.9 DOUT PECL Output True data output.10 VCC Power Supply Positive power supply.Absolute Maximum Ratings(1)Supply Voltage (V CC).......................................0V to +7.0V Input Voltage (DIN, /DIN).......................................0 to V CC Output Current (I OUT)Continuous........................................................±50mA Surge..............................................................±100mA EN Voltage.............................................................0 to V CC V REF Current..........................................-800µA to +500µA SD LVL Voltage...................................................V REF to V CC Lead Temperature (soldering, 20sec.).....................260°C Storage Temperature (T s).......................–65°C to +150°C Operating Ratings(2)Supply Voltage (V CC).............................+3.0V to +3.6V Ambient Temperature (T A)...................–40°C to +85°C Junction Temperature (T J).................–40°C to +120°C Junction Thermal ResistanceMSOP(θJA) Still-air...................................113°C/WDC Electrical CharacteristicsV CC = 3.0 to 3.6V; R L = 50Ω to V CC-2V; T A = –40°C to +85°C, typical values at V CC = 3.3V, T A = 25°C.Symbol Parameter Condition Min Typ Max Units I CC Power Supply Current No output load 26 39 mASD LVL SD LVL Voltage V REF V CC V V OH PECL Output HIGH Voltage V CC-1.085 V CC-0.955 V CC-0.880VV OL PECL Output LOW Voltage V CC-1.830 V CC-1.705 V CC-1.555VV IHCMR Common Mode Range GND+2.0 V CC V V REF ReferenceVoltage V CC-1.48 V CC-1.32 V CC-1.16VTTL DC Electrical CharacteristicsV CC = 3.0 to 3.6V; R L = 50Ω to V CC-2V; T A = –40°C to +85°C, typical values at V CC = 3.3V, T A = 25°C.Symbol Parameter Condition Min Typ Max Units V IH EN Input HIGH Voltage 2.0 VV IL EN Input LOW Voltage 0.8 VI IH EN Input HIGH Current V IN = 2.7VV IN = V CC20 100µAµAI IL EN Input LOW Current V IN = 0.5V -0.3 mAV OH SD Output HIGH Level V CC > 3.3V, I OH-MAX < 160µAV CC < 3.3V, I OH-MAX < 160µA 2.42.0VVV OL SD Output LOW Level I OL = +2mA 0.5 V Notes:1. Permanent device damage may occur if absolute maximum ratings are exceeded. This is a stress rating only and functional operation is notimplied at conditions other than those detailed in the operational sections of this data sheet. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.2. The data sheet limits are not guaranteed if the device is operated beyond the operating ratings.3. Thermal performance assumes the use of a 4-layer PCB. Exposed pad must be soldered (or equivalent) to the device’s most negative potentialon the PCB.AC Electrical CharacteristicsV CC = 3.0V to 3.6V; R LOAD = 50Ω to V CC–2V; T A = –40°C to +85°C.Symbol Parameter Condition MinTypMaxUnitst r, t f Output Rise/Fall Time(20% to 80%) Note 4 260pst JITTER DeterministicRandom Note 5Note 6155ps PPps RMSV ID Differential Input Voltage Swing Figure 1 5 1800 mV PPV OD Differential Output VoltageSwing V ID > 18mV PP Figure 11500mV PPT OFF SD Release Time 100 500 nsT ON SDAssertTime 100500ns SD AL Low SD Assert Level R SDLVL = 15kΩ, Note 8 3.4 mV PPSD DL Low SD De-assert Level R SDLVL = 15kΩ, Note 8 2.3 mV PPHYS L Low SD Hysteresis R SDLVL = 15kΩ, Note 7 3.4 dBSD AM Medium SD Assert Level R SDLVL = 5kΩ, Note 8 6.2 8 mV PPSD DM Medium SD De-assert Level R SDLVL = 5kΩ, Note 8 3 4.2 mV PPHYS M Medium SD Hysteresis R SDLVL = 5kΩ, Note 7 2 3.4 5 dBSD AH High SD Assert Level R SDLVL = 100Ω, Note 8 16.4 20 mV PPSD DH High SD De-assert Level R SDLVL = 100Ω, Note 8 8 10.8 mV PPHYS H High SD Hysteresis R SDLVL = 100Ω, Note 7 2 3.4 5 dBB-3dB 3dBBandwidth 1 GHz A V(Diff)Differential Voltage Gain 42 dBS21Single-ended Small-Signal Gain 36 dBNotes:4. Amplifier in limiting mode. Input is a 200MHz square wave.5. Deterministic jitter measured using 1.25Gbps K28.5 pattern, V ID = 10mV PP.6. Random jitter measured using 1.25Gbps K28.7 pattern, V ID = 10mV PP.7. This specification defines electrical hysteresis as 20log(SD Assert/SD De-assert). The ratio between optical hysteresis andelectrical hysteresis is found to vary between 1.5 and 2 depending upon the level of received optical power and ROSAcharacteristics. Based upon that ratio, the optical hysteresis corresponding to the electrical hysteresis range 2dB-5dB, shownin the AC characteristics table, will be 1dB-4dB optical Hysteresis.8. See “Typical Operating Characteristics” for a graph showing how to choose a particular R SDLVL for a particular SD assert andits associated de-assert amplitude.Typical Operating CharacteristicsV CC = 3.3V, T A = 25°C, R L = 50Ω to V CC–2V, unless otherwise stated.R SDLVL (kΩ) R SDLVL (kΩ)Functional Block DiagramDetailed DescriptionThe SY88149CL high-sensitivity limiting post amplifier operates from a single +3.3V power supply, over temperatures from –40°C to +85°C. Signals with data rates up to 1.25Gbps and as small as 5mV PP can be amplified. Figure 1 shows the allowed input voltage swing. The SY88149CL generates an SD output, allowing feedback to EN for output stability. SD LVL sets the sensitivity of the input amplitude detection.Input Amplifier/BufferFigure 2 shows a simplified schematic of the input stage. The high-sensitivity of the input amplifier allows signals as small as 5mV PP to be detected and amplified. The input amplifier allows input signals as large as 1800mV PP . Input signals are linearly amplified with a typically 42dB differential voltage gain. Since it is a limiting amplifier, the SY88149CL outputs typically 1500mV PP voltage-limited waveforms for input signals that are greater than 12mV PP . Applications requiring the SY88149CL to operate with high-gain should have the upstream TIA placed as close as possible to the SY88149CL’s input pins. This ensure the best performance of the device.Output BufferThe SY88149CL’s PECL output buffer is designed to drive 50Ω lines. The output buffer requires appropriate termination for proper operation. An external 50Ω resistor to V CC –2V for each output pin provides this. Figure 3 shows a simplified schematic of the output stage.Signal DetectThe SY88149CL generates a chatter-free Signal-Detect (SD) open-collector TTL output with internal 4.75k Ω pull-up resistor, as shown in Figure 4. SD is used to determine that the input amplitude is too small to be considered a valid input. SD asserts high if the input amplitude rises above threshold set by SDLVL and de-asserts low otherwise. SD can be fed back to the enable (EN) input to maintain output stability under a SDs of signal condition. EN de-asserts low the true output signal without removing the input signals. Typically, 3.4dB SD hysteresis is provided to prevent chattering.Signal Detect Level SetA programmable SD level set pin (SD LVL ) sets the threshold of the input amplitude detection. Connecting an external resistor between V CC and SD LVL sets the voltage at SD LVL . This voltage ranges from V CC to V REF . The external resistor creates a voltage divider between V CC and V REF , as shown in Figure 5.HysteresisThe SY88149CL provides typically 3.4dB SD electrical hysteresis. By definition, a power ratio measured in dB is 10log (power ratio). Power is calculated as V 2IN /R for an electrical signal. Hence the same ratio can be stated as 20log (voltage ratio). While in linear mode, the electrical voltage input changes linearly with the optical power and hence the ratios change linearly. Therefore, the optical hysteresis in dB is half the electrical hysteresis in dB given in the data sheet. The SY88149CL is an electrical device; this data sheet refers to hysteresis in electrical terms. With 3.4dB SD hysteresis, a voltage factor of 1.5 is required to assert or de-assert SD.Figure 1. VIS and VID DefinitionFigure 2. Input Structure Figure 3. Output StructureFigure 4. SD Output StructureFigure 5. SD LVL Setting CircuitNote: Recommended value for R SDLVL is 15kΩ or less.Related Product and Support DocumentationPart Number Function Data Sheet Link/product-info/sy88933al.shtmlSY88933AL 3.3V/5V 1.25Gbps PECL High-SensitivityLimiting Post Amplifier with TTL SD/product-info/app_hints+notes.shtml Application Notes Notes on Sensitivity and Hysteresis in MicrelPost AmplifiersPackage Information10-Pin MSOP (K10-1)。

普芦卡必利和莫沙必利分别联用小剂量聚乙二醇治疗老年难治性功能性便秘的短期疗效比较

普芦卡必利和莫沙必利分别联用小剂量聚乙二醇治疗老年难治性功能性便秘的短期疗效比较黄海辉;张小敏;赵亮【摘要】目的比较普芦卡必利、莫沙必利分别联用小剂量聚乙二醇治疗老年难治性功能性便秘的短期疗效.方法对2014年5月至2016年2月在该院门诊诊治的老年难治性功能性便秘患者90例进行回顾性分析.入选病例分为两组,每组45例.普芦卡必利组:琥珀酸普芦卡必利片,2 mg,每日1次;莫沙必利组:枸橼酸莫沙必利胶囊,5 mg,每日3次.两组均联用复方聚乙二醇电解质散(PEG)13.125 g,每日2次,疗程4周.观察两组患者首次排便和排便困难缓解时间、每周完全自发排便(SCBM)平均次数、排便困难、大便性状、不良反应及生命质量的变化.结果两组慢传输型、排便障碍型的治疗有效率比较,差异均有统计学意义(P<0.05),混合型差异无统计学意义(P>0.05).与莫沙必利组比较,普卢卡必利组首次排便时间和排便困难缓解时间均较短,差异有统计学意义(P<0.05).治疗后4周,两组患者SCBM>3次,普卢卡必利组次数更多;普芦卡必利组排便困难改善更为明显,差异有统计学意义(P<0.05).大便性状改善组间比较,差异无统计学意义(P>0.05).两组总有效率比较差异有统计学意义(P<0.05),不良反应发生率比较差异无统计学意义(P>0.05).治疗后4周,两组患者便秘患者生存质量自评量表(PAC-QOL)评分的总均分均有下降,普芦卡必利组降低更为明显,差异有统计学意义(P<0.05).结论普芦卡必利+PEG治疗老年难治性功能性便秘起效更快,且在总体疗效及生命质量改善方面有优势,尤适用于慢传输型和排便障碍型.%Objective To compare the short-term curative effects of prucalopride and mosapride respectively combined with low dose polyethylene glycol in treating elderly refractory functional constipation.Methods Ninety patients with elderly refractory functionalconstipation in the outpatient department of our hospital from May 2014 to February 2016 were retrospectively analyzed and divided into two groups randomly,45 cases in each group.the prucalopridegroup:Prucalopride Succinate Tablets,2mg,4 times daily;the mosapride group:Mosapride Citrate Capsules,5mg,3 times daily.Polyethylene Glycol Electrolytes Powder(PEG) was also used in the two groups,13.125g,twice daily.The course of treatment was 4 weeks.The first defecating time and defecation difficulty relief time,average weekly spontaneous complete bowel movements (SCBM),defecating difficulty,stool character,adverse reactions and change of life quality were observed in the twogroups.Results The treatment effective rate of slow transit constipation(STC) and defecatory disorder had the statistical difference between the two groups (P<0.05).The comparison of the effective rates in mixed type showed no statistical difference between the twogroups(P>0.05).Compared with the mosapride group,the first defecating time and defecation difficulty relief time in the prucalopride group were shorter with statistical difference (P<0.05).After 4-week treatment,SCBM times per week in the two groups were more than 3 times;the times of the prucalopride group were even more.In the prucalopride group,the defecation difficulty improvement was more obvious,the difference between the two groups had statistical significance (P<0.05).As for the comparison of the stool character improvement,the difference had no statistical significance(P>0.05).The total effective rate had statistical difference between the two groups(P<0.05).The incidence rate of adversereactions had no statistical difference between the two groups(17.78%vs.15.56%,P<0.05).The total average score of PAC-QOL after treatment in the two groups were both decreased,moreover the decrease in the prucalopride group was more obvious;the difference between the two groups had statistical significance(P<0.05).Conclusion Prucalopride +PEG take effect faster in the treatment of elderly refractory functional constipation and has the advantages in the aspects of overall curative effect and life quality improvement,which is specially suitable for STC and defecatory disorder type.【期刊名称】《重庆医学》【年(卷),期】2017(046)020【总页数】4页(P2793-2796)【关键词】难治性功能性便秘;老年;聚乙二醇;琥珀酸普芦卡必利;枸橼酸莫沙必利【作者】黄海辉;张小敏;赵亮【作者单位】广东省河源市源城区人民医院消化内科 517000;广东省河源市源城区人民医院消化内科 517000;广东省河源市源城区人民医院消化内科 517000【正文语种】中文【中图分类】R57我国60岁以上人群慢性便秘的患病率高达22%，其中约50%为功能性便秘[1-3]。

胃结肠反射的研究概况

胃结肠反射的研究概况黄敏;陈继红;谭诗云;罗和生;JanDHUIZINGA【摘要】Gastrocolic reflex is one of physiological phenomenon in gastrointestinal tract,which is involved in the development of functional gastrointestinal disorders and has potential clinical value. Main methods for research on gastrocolic reflex are electromyography and colon manometry,increases of colonic spike bursts and colonic motility index occur after meal. Recent studies showed specific postprandial colonic motor patterns by using manometry,however,the mechanism of gastrocolic reflex is still uncertain. This article reviewed the study on gastrocolic reflex.%胃结肠反射是胃肠道生理现象之一，参与功能性胃肠疾病的发生和发展，具有潜在的临床价值。

胃结肠反射的研究方法主要为肌电图和结肠测压法，可见餐后结肠尖峰突发波或结肠动力指数增加。

近年应用测压法发现餐后结肠具有特定的运动模式，但胃结肠反射的表现方式和机制仍不清楚。

本文就胃结肠反射的研究概况作一综述。

【期刊名称】《胃肠病学》【年(卷),期】2016(021)008【总页数】4页(P505-508)【关键词】胃结肠反射;进食;肌电描记术;测压法【作者】黄敏;陈继红;谭诗云;罗和生;JanDHUIZINGA【作者单位】武汉大学人民医院消化内科消化系统疾病湖北省重点实验室430060;武汉大学人民医院消化内科消化系统疾病湖北省重点实验室 430060;武汉大学人民医院消化内科消化系统疾病湖北省重点实验室 430060;武汉大学人民医院消化内科消化系统疾病湖北省重点实验室 430060;麦克马斯特大学健康科学学院医学【正文语种】中文结肠功能紊乱性疾病如便秘为最常见的疾病之一，但其发病机制仍未完全阐明。

声音是一种由物体振动引发的物理现象(课堂PPT)

6
• 次声 • 人耳能听见的声音 • 超声 • 特超声
0～20Hz 20Hz～20KHz 20KHz～1GHz 1GHz～10THz
• Infra-sound
from 0 to 20 Hz
• Human hearing frequency range from 20Hz to 20kHz
• Ultrasound
1
•Байду номын сангаас声音和音频技术涉及的主要是声波(声学信号)的处理。
编码、录音机或数字音频带的存储、音乐及语音处理是这一领域的主要课题。
• Sound methodology and audio techniques engage in processing these sound waves (acoustic signals). Important topics in this area are coding, storage on recorders or digital audio tapes, music and speed processing.
• The waveform repeats the same shape at regular intervals and this portion is called a period. Since sound waves occur naturally, they are never perfectly smooth or uniformly periodic. However, sounds that display a recognizable periodicity tend to be more musical than those that are nonperiodic. Examples of periodic sound sources are musical instruments, vowel sounds, the whistling wind and bird songs. Nonperiodic sound sources include unpitched percussion instruments, coughs and sneezes and rushing water.

皇冠 XLS系列功放

Obteniendo otras Versiones de Idioma: Para obtener información en otro idioma acerca del uso de este producto, contacte por favor a su Distribuidor Crown local. Si necesita asistencia para localizar a su distribuidor local, favor de contactar a Crown al 574-294-8000. Este manual no incluye todos los detalles de diseño, producción, o variaciones del equipo. Tampoco cubre cualquier posible situación que pueda surgir durante la instalación, operación o mantenimiento. La información provista en este manual fue considerada precisa a la fecha de publicación. Sin embargo, pueden haber ocurrido actualizaciones a esta información. Para obtener la última versión de este manual, por favor visite la página de internet de Crown en . Nota de Marcas Registradas: Crown, Crown Audio y Amcron son marcas registradas de Crown International. Otras marcas son de la propiedad de sus respectivos dueños. Algunos modelos pueden ser exportados bajo el nombre de . Amcron ® Derechos reservados por Crown Audio Inc, ©2008., 1718 W. Mishawaka Rd., Elkhart Indiana 46517-9439 U.S.A® 001-574-294-8000 .,

杰出的理论与实验物理学家——瑞利勋爵（Lord

杰出的理论与实验物理学家——瑞利勋爵（Lord Rayleigh）1904年诺贝尔物理学奖授予英国皇家研究所的瑞利勋爵（Lord Rayleigh， 1842～1919），以表彰他在研究⼀些⽓体的密度中发现了惰性⽓体氩这⼀重要成就。

瑞利原名约翰·威廉·斯特拉特（John William Strutt），尊称瑞利男爵三世（Third Baron Rayleigh），1842年11⽉12⽇出⽣于英国埃塞克斯郡莫尔登（Malden）的朗弗德林园。

他的⽗亲是第⼆世男爵约翰·詹姆斯·斯特拉特，母亲叫克拉腊·伊丽莎⽩·拉图哲，是理查德·维卡斯海军上校的⼩⼥⼉。

出⾝名望贵族的瑞利以严谨、⼴博、精深著称，并善于⽤简单的设备作实验⽽能获得⼗分精确的数据。

⽓体密度测量本来是实验室中的⼀件常规⼯作，但是瑞利不放过常⼈不当回事的实验差异，终于作出了惊⼈的重⼤发现。

这就是1892年瑞利从密度的测量中发现了第⼀个惰性⽓体——氩。

⾃从门捷列夫周期表提出以后，科学家对寻找新的元素以填补周期表上的空缺，表现出了很⼤的积极性。

但是，⼈们没有想到，竟然在周期表上遗漏了整整⼀族性质特殊的惰性⽓体。

1882年，瑞利为了证实普劳特假说，曾经测过氢和氧的密度。

经过⼗年长期的测定，他宣布氢和氧的原⼦量之⽐实际上不是1：16，⽽是1：15.882。

他还测定了氮的密度，他发现从液态空⽓中分馏出来的氮，跟从亚硝酸铵中分离出来的氮，密度有微⼩的但却是不可忽略的偏差。

从液态空⽓中分馏出来的氮，密度为1.2572 g/cm3，⽽⽤化学⽅法从亚硝酸铵直接得到的氮，密度却为 1.2505 g/cm3。

两者数值相差千分之⼏，在⼩数点后第三位不相同。

他认为，这⼀差异远远超出了实验误差范围，⼀定有尚未查清的因素在起作⽤。

为此他先后提出过⼏种假说来解释造成这种不⼀致的原因。

其中有⼀种是认为在⼤⽓中的氮还含有⼀种同素异形体，就像氧和臭氧那样，这种同素异形体混杂在⼤⽓氮之中，⽽从化学⽅法所得应该就是纯净的氮。

2010-PE-07---Effects of Discretization Methods on the Performance of Resonant Controllers

Effects of Discretization Methods on the Performance of Resonant ControllersAlejandro G.Yepes,Student Member,IEEE,Francisco D.Freijedo,Member,IEEE, Jes´u s Doval-Gandoy,Member,IEEE,´Oscar L´o pez,Member,IEEE,Jano Malvar,Student Member,IEEE,and Pablo Fernandez-Comesa˜n a,Student Member,IEEEAbstract—Resonant controllers have gained signiﬁcant impor-tance in recent years in multiple applications.Because of their high selectivity,their performance is very dependent on the ac-curacy of the resonant frequency.An exhaustive study about dif-ferent discrete-time implementations is contributed in this paper. Some methods,such as the popular ones based on two integrators, cause that the resonant peaks differ from expected.Such inac-curacies result in signiﬁcant loss of performance,especially for tracking high-frequency signals,since inﬁnite gain at the expected frequency is not achieved,and therefore,zero steady-state error is not assured.Other discretization techniques are demonstrated to be more reliable.The effect on zeros is also analyzed,establishing the inﬂuence of each method on the stability.Finally,the study is extended to the discretization of the schemes with delay compensa-tion,which is also proved to be of great importance in relation with their performance.A single-phase active powerﬁlter laboratory prototype has been implemented and tested.Experimental results provide a real-time comparison among discretization strategies, which validate the theoretical analysis.The optimum discrete-time implementation alternatives are assessed and summarized.Index Terms—Current control,digital control,power condition-ing,pulsewidth-modulated power converters,Z transforms.N OMENCLATUREVariablesC Capacitance.f Frequency in hertz.G(s)Model in the s domain.G(z)Model in the z domain.H(s)Resonant controller in the s domain.H(z)Resonant controller in the z domain.i Current.K Gain of resonant controller.L Inductance value.m Pulsewidth modulation(PWM)duty cycle. N Number of samples to compensate with com-putational delay compensation.n Highest harmonic to be compensated. Manuscript received September17,2009;revised December29,2009.Date of current version June18,2010.This work was supported by the Spanish Min-istry of Education and Science under Project DPI2009-07004.Recommended for publication by Associate Editor P.Mattavelli.The authors are with the Department of Electronic Technology,University of Vigo,Vigo36200,Spain(e-mail:agyepes@uvigo.es;fdfrei@uvigo.es;jdoval@ uvigo.es;olopez@uvigo.es;janomalvar@uvigo.es;pablofercom@uvigo.es). Color versions of one or more of theﬁgures in this paper are available online at .Digital Object Identiﬁer10.1109/TPEL.2010.2041256R Equivalent series resistance value.R(s)Resonant term in the s domain.R(z)Resonant term in the z domain.T Period.θPhase of grid voltage.V V oltage.ωAngular frequency in radians per second.u(s)Input value.y(s)Output value.Subscripts1Fundamental component.a Actual value(f).c Generic current controller(G).d Degree of freedom in the zero-pole matchingdiscretization method(K).dc Relative to the dc link(V).f Relative to the passive inductiveﬁlter(V,i,L,R,and G).I Equivalent to the double of the integral gainof a proportional+integral(PI)controller indq frame(K).k Relative to the k th harmonic(H,R,K P,andK I).L Relative to the load(i).Lh Relative to the harmonics of the load(i).o Resonant frequency of a continuous resonantterm or resonant controller(f andω).P Equivalent to the double of the proportionalgain of a PI controller in dq frame(K). PCC Relative to the point of common coupling(V).PL Relative to the plant(G).rms Root mean square.s Relative to sampling(f and T).src Relative to the voltage source(V,i,and L). sw Relative to switching(f).T Sum of the gains for every value of harmonicorder k(K P).X Resonant term R or resonant controller Hdiscretized with method X,where X∈{zoh,foh,f,b,t,tp,zpm,imp}.X&Y Resonant term R or resonant controller Himplemented with two discrete integrators,with the direct one discretized with method Xand the feedback one with method Y,whereX,Y∈{zoh,foh,f,b,t,tp,zpm,imp}.0885-8993/$26.00©2010IEEEX−Y Resonant controller H VPI(z),in whichR1(s)is discretized with method X andR2(s)with method Y,where X,Y∈{zoh,foh,f,b,t,tp,zpm,imp}. Superscripts∗Reference value.1Resonant term R of the form s/(s2+ω2o). 2Resonant term R of the form s2/(s2+ω2o).d Including delay compensation(H and R). PR Resonant controller H of the PR type.VPI Resonant controller H of the VPI type. Others∆x Difference between x and its target value(i f).ˆx Estimated value of x(θ1andω1).I.I NTRODUCTIONI N recent years,resonant controllers have gained signiﬁcantimportance in a wide range of different applications due to their overall good performance.They have been applied with satisfactory results to cases such as distributed power generation systems[1],[2],dynamic voltage regulators[3],[4],wind tur-bines[5],[6],photovoltaic systems[7],[8],fuel cells[9],[10], active rectiﬁers[11],active powerﬁlters(APFs)[12]–[17], microgrids[18],and permanent magnet synchronous motors [19].Resonant controllers allow to track sinusoidal references of arbitrary frequencies with zero steady-state error for both single-phase and three-phase applications.An important saving of computational burden and complexity is obtained due to their implementation in stationary frame,avoiding the coordinates transformations,and providing perfect tracking of both positive and negative sequences[1],[13],[14],[20]–[22].Resonant con-trollers in synchronous reference frame(SRF)have been also proposed to control pairs of harmonics simultaneously when no unbalance exist[7],[15]–[17],[22],[23].An essential step in the implementation of resonant digital controllers is the discretization.Because of the narrow band and inﬁnite gain of resonant controllers,they are specially sensitive to this process.Actually,a slight displacement of the resonant poles causes a signiﬁcant loss of performance.In the case of proportional+resonant(PR)controllers[14],[20]–[22],even for small frequency deviations,the effect of resonant terms becomes minimal,and the PR controller behaves just as a proportional one[14].The resonant regulator proposed in[16]is less sensitive to these variations when cross coupling due to the plant appears in the dq frame,but if these deviations in the resonant poles are present,it does not achieve zero steady-state error either. Furthermore,if selectivity is reduced to increase robustness to frequency variations,undesired frequencies and noise may be ampliﬁed.Thus,an accurate peak position is preferable to low selectivity.Therefore,it is of paramount importance to study the effectiveness of the different alternatives of discretization for implementing digital resonant controllers,due to the critical characteristics of their frequency response.As proved in this paper,many of the existing discretization techniques cause a displacement of the poles.This fact results in a deviation of the frequency at which the inﬁnite gain occurs with respect to the expected resonant frequency.This error becomes more signiﬁcant as the sampling time and the desired peak frequency increase.In practice,it can be stated that most of these discretization methods result in suitable implementations when tracking50/60Hz(fundamental)references and even for low-order harmonics.However,as shown in this paper,some of them do not perform so well in applications in which signals of higher frequencies should be tracked,such as APFs and ac motor drives. This error has special relevance in the case of implementations based on two integrators,since it is a widely employed option mainly due to its simplicity for frequency adaptation[8],[13], [15],[23]–[25].Discretization also has an effect on zeros,modifying their distribution with respect to the continuous transfer function. These discrepancies should not be ignored because they have a direct relation with stability.In fact,resonant controllers are often preferred to be based on the Laplace transform of a cosine function instead of that of a sine function because its zero im-proves stability[13],[19].In a similar way,the zeros mapped by each technique will affect the stability in a different man-ner.Consequently,it is also convenient to establish which are the most adequate techniques from the point of view of phase versus frequency response.However,for large values of the resonance frequency,the computational delay affects the system performance and may cause instability.Therefore,a delay compensation scheme should be implemented[14],[15],[17],[23].It can be per-formed in the continuous domain as proposed in[15].However, the discretization of that scheme leads to several different expressions.A possible implementation in the z domain was posed in[14],but there are other possibilities.Consequently,it should be analyzed how each method affects the effectiveness of the computational delay compensation.This aspect has a signiﬁcant relevance since it will determine the stability at the resonant frequencies.The study of these effects of the discretization on resonant controllers has not been analyzed in the existing literature. Therefore,it is of paramount importance to analyze how each method affects the performance in relation with these aspects.A single-phase APF laboratory prototype has been built to check the theoretical approaches,because it is an application very suitable for proving the controllers performance when tracking different frequencies,and results can be extrapolated to other single-phase and three-phase applications where a perfect tracking/rejection of references/disturbances is sought through resonant controllers.The paper is organized as follows.Section II presents alterna-tive digital implementations of resonant controllers.The reso-nant peak displacement depending on the discretization method, as well as its inﬂuence on stability,is analyzed in Section III. Several discrete-time implementations including delay compen-sation,and a comparison among them,are posed in Section IV. Section V summarizes the performance of the digital imple-mentations in each aspect and establishes the most optimum alternatives depending on the existing requirements.Finally, experimental results of Section VII validate the theoreticalanalysis regarding the effects of discretization on the perfor-mance of resonant controllers.II.D IGITAL I MPLEMENTATIONS OF R ESONANT C ONTROLLERS A.Resonant Controllers in the Continuous DomainA PR controller can be expressed in the s domain as[14],[20]–[22]H PR(s)=K P+K Iss2+ω2o=K P+K I R1(s)(1)withωo being the resonant angular frequency.R1(s)is the resonant term,which has inﬁnite gain at the resonant frequency (f o=ωo/2π).This assures perfect tracking for components rotating at f o when implemented in closed-loop[21].R1(s) is preferred to be the Laplace transform of a cosine function instead of that of a sine function,since the former provides better stability[13],[19].H PR(s)in stationary frame is equivalent to a propor-tional+integral(PI)controller in SRF[21].However,if cross coupling due to the plant is present in the dq frame,unde-sired peaks will appear in the frequencies around f o in closed loop[17].This anomalous behavior worsens even more the per-formance when frequency deviates from its expected value.An alternative resonant regulator,known as vector PI(VPI)con-troller,is proposed in[16]:H VPI(s)=K P s2+K I ss2+ω2o.(2)The VPI controller cancels coupling terms produced when the plant has the form1/(sL f+R f)[16],[17],[23],such as in shunt APFs and ac motor drives,with L f and R f being, respectively,the inductance and the equivalent series resistance of an R–Lﬁlter.Parameters detuning due to estimation errors in the values of L f and R f has been proved in[17]to have small inﬂuence on the performance.H VPI(s)can be decomposed as the sum of two resonant terms,R1(s)and R2(s),as follows:H VPI(s)=K Ps2s2+ω2o+K Iss2+ω2o=K P R2(s)+K I R1(s).(3) Equation(3)permits to discretize R1(s)and R2(s)with dif-ferent methods.In this manner,the most optimum alternative for H VPI(z)will be the combination of the most adequate discrete-time implementation for each resonant term.B.Implementations Based on the Continuous Transfer Function DiscretizationTable I shows the most common discretization methods.The Simpson’s rule approximation has not been included because it transforms a second-order function to a fourth-order one,which is undesirable from an implementation viewpoint[26].The techniques reﬂected in Table I have been applied to R1(s) and R2(s),leading to the discrete mathematical expressions shown in Table II.T s is the controller sampling period and f s=1/T s is the sampling rate.From Table II,it can be seen thatTABLE IR ELATIONS FOR D ISCRETIZING R1(s)AND R2(s)BY D IFFERENT METHODS the effect of each discretization method on the resonant poles displacement will be equal in both R1(s)and R2(s),since each method leads to the same denominator in both resonant terms. It should be noted that zero-pole matching(ZPM)permits a degree of freedom(K d)to maintain the gain for a speciﬁc frequency[26].C.Implementations Based on Two Discrete IntegratorsThe transfer function H PR(s)can be discretized by decom-posing R1(s)in two simple integrators,as shown in Fig.1(a) [13].This structure is considered advantageous when imple-menting frequency adaptation,since no explicit trigonometric functions are needed.Whereas other implementations require the online calculation of cos(ωo T s)terms,in Fig.1schemes the parameterωo appears separately as a simple gain,so it can be modiﬁed in real time according to the actual value of the frequency to be controlled.Indeed,it is a common practice to implement this scheme due to the simplicity it permits when frequency adaptation is required[13],[15],[24],[25].An analogous reasoning can be applied to H VPI(s),leading to the block diagram shown in Fig.1(b).Instead of developing an equivalent scheme to the total transfer function H VPI(s), it could be obtained as an individual scheme for implementing each resonant term R1(s)and R2(s)could be obtained,but in this case the former is preferable because of the saving of resources.It has been suggested in[8]to discretize the direct integrator of Fig.1(a)scheme using forward Euler method and the feedback one using the backward Euler method.Additional alternatives of discretization for both integrators have been analyzed in[25], and it was also proposed to use Tustin for both integrators,or to discretize both with backward Euler,adding a one-step delay in the feedback line.Nevertheless,using Tustin for both integrators poses implementation problems due to algebraic loops[25].In this paper,these proposals have been also applied to the block diagram shown in Fig.1(b).Table III shows these three discrete-time implementations of the schemes shown in Fig.1.TABLE IIz -D OMAIN T RANSFER F UNCTIONS O BTAINED BY D ISCRETIZING R 1(s )AND R 2(s )BY D IFFERENT METHODSFig.1.Block diagrams of frequency adaptive resonant controllers (a)H P R (s )and (b)H V P I (s )based on two integrators.It should be noted that H jt&t (z )and H j t (z )are equivalent for both j =PR and j =VPI ,since the Tustin transformation is based on a variable substitution.The same is true for the rest of methods that consist in substituting s as a function of z .However,zero-order hold (ZOH),ﬁrst-order hold (FOH),and impulse invariant methods applied separately to each integratordo not lead to H j zoh,H j foh ,and H jimp ,respectively.Indeed,to dis-cretize an integrator with ZOH or FOH results in the same way as a forward Euler substitution,while to discretize an integrator with the impulse invariant is equivalent to employ backward Euler.III.I NFLUENCE OF D ISCRETIZATION M ETHODSON R OOTS D ISTRIBUTIONA.Resonant Poles DisplacementThe z domain transfer functions obtained in Section II can be grouped in the sets of Table IV,since some of them present an identical denominator,and therefore,coinciding poles.Fig.2represents the pole locus of the transfer functions in Table IV.Damped resonant controllers do not assure perfect tracking [21];poles must be placed in the unit circumference,which corresponds to a zero damping factor (inﬁnite gain).All discretization techniques apart from A and B lead to undamped poles;the former maps the poles outside of the unit circle,whereas the latter moves them toward the origin,causing a damping factor different from zero,so both methods should be avoided.This behavior ﬁnds its explanation in the fact that these two techniques do not map the left half-plane in the s domain to the exact area of the unit circle [26].However,there is an additional issue that should be taken into account.Although groups C ,D ,and E achieve inﬁnite gain,it can be appreciated that,for an identical f o ,their poles are located in different positions of the unit circumference.This fact reveals that there exists a difference between the actual resonant frequency (f a )and f o ,depending on the employed implementation,as also observed in Fig.3(d).Consequently,the inﬁnite gain may not match the frequency of the controller references,causing steady-state error.Fig.3(a)–(c)depicts the error f o −f a in hertz as a function of f o and f s for each group.The poles displacement increases with T s and f o ,with the exception of group E .The slope of the error is also greater as these parameters get higher.Actually,the denominator of group D is a second-order Tay-lor series approximation of group E .This fact explains the in-creasing difference between them as the product ωo T s becomes larger.Some important outcomes from this study should be highlighted.1)The Tustin transformation,which is a typical choice in digital control due to its accuracy in most applications,features the most signiﬁcant deviation in the resonant frequency.2)The error exhibited by the methods based on two dis-cretized integrators becomes signiﬁcant even for highTABLE IIID ISCRETE T RANSFERF UNCTIONS H P R (s )AND H V P I (s )O BTAINED BY E MPLOYING T WO D ISCRETIZED INTEGRATORSTABLE IVG ROUPS OF E XPRESSIONS W ITH I DENTICAL P OLES IN THE z DOMAINFig.2.Pole locus of the discretized resonant controllers at f s =10kHz (fundamental to the 17th odd harmonics).sampling frequencies and low-order harmonics.For in-stance,at f s =10kHz,group D exhibits an error of +0.7Hz for the seventh harmonic,which causes a consid-erable gain loss [see Fig.3(d)].When dealing with higher harmonic orders (h ),such as 13and 17,it raises to 4.6and 10.4Hz,respectively,which is unacceptable.3)Group E leads to poles that match the original continuous ones,so the resonant peak always ﬁts the design frequency f o .B.Effects on Zeros DistributionOnce assured inﬁnite gain due to a correct position of the poles,another factor to take into account is the displacement of zeros caused by the discretization.Resonant controllers that be-long to group E have been proved to be more suitable for an op-timum implementation in terms of resonant peak displacement.However,the numerators of these discrete transfer functions are not the same,and they depend on the discretization method.This aspect has a direct relation with stability,so it should not be ignored.On the other hand,although group D methods produce a resonant frequency error,they avoid the calculation of explicit cosine functions when frequency adaptation is needed.This fact may imply an important saving of resources.Therefore,it is also of interest to establish which is the best option of that set.The analysis will be carried out by means of the frequency response.The inﬁnite gain at ωo is given by the poles po-sition,whereas zeros only have a visible impact on the gain at other frequencies.Concerning phase,the mapping of zeros provided by the discretization may affect all the spectrum,in-cluding the phase response near the resonant frequency.Due to the high gain around ωo ,the phase introduced by the reso-nant terms at ω≈ωo will have much more impact on the phase response of the whole controllers than at the rest of the spec-trum [14].Therefore,the inﬂuence of discretization on the stabil-ity should be studied mainly by analyzing the phase lag caused at ω≈ωo .1)Displacement of R 1(s )Zeros by Group E Discretiza-tions:Fig.4compares the frequency response of a resonant controller R 1(s ),designed for the seventh harmonic,when dis-cretization methods of group E are employed at f s =10kHz.An almost equivalent magnitude behavior is observed,eventhough R 1imp(z )has a lower attenuation in the extremes,and both R 1tp (z )and R 1foh (z )tend to reduce the gain at high fre-quencies.However,the phase versus frequency plot differs more signiﬁcantly.From Fig.4,it can be appreciated that R 1tp(z )and R 1foh (z )are the most accurate when comparing with R 1(s ).On thecontrary,the phase lag introduced by R 1zoh (z )and R 1zpm (z )is higher than for the continuous model.This fact is particu-larly critical at ω≈ωo ,even though they also cause delay for higher frequencies.As shown in Fig.4,they introduce a phase lag at f o =350Hz of 6.3◦.For higher values of ωo T s ,it be-comes greater.For instance,if tuned at a resonant frequency of f o =1750Hz with f s =10kHz,the delay is 32◦.There-fore,the implementation of R 1zoh (z )and R 1zpm (z )may lead to instability.On the other hand,R 1tp (z ),R 1foh (z ),and R 1imp(z )accurately reproduce the frequency response at the resonance frequency,maintaining the stability of the continuous controllerat ωo .Fig.4also shows that R 1imp(z )can be considered the most advantageous implementation of R 1(s ),since it maintains the stability at ω≈ωo and introduces less phase lag in open-loop for the rest of the spectrum,thereby allowing for a larger phase margin.Fig.3.Deviation of the resonance frequency of the discretized controller f a from the resonance frequency f o of the continuous controller.(a)Group C transfer functions.(b)Group D transfer functions.(c)Group E transfer functions.(d)Discretized seventh harmonic resonant resonant controller at f s= 10kHz.Fig.4.Bode plot of R1(s)discretized with group E methods for a seventh harmonic resonant controller at f s=10kHz.In any case,the inﬂuence of the discretization atω=ωo is not as important as its effect on the stability atω≈ωo,since the gain of R1(z)is much lower at those frequencies.Consequently, this aspect can be neglected unless low sample frequencies, high resonant frequencies,and/or large values of K I/K P are employed.In these cases,it can be taken into account in order to avoid unexpected reductions in the phase margin that could affect the stability,or even to increase its value over the phase margin of the continuous system by means of R1imp(z).2)Displacement of R2(s)Zeros by Group E Discretizations: The frequency response of R2(s)discretizations is shown in Fig.5(a).It can be seen that ZOH produces a phase lag near the resonant frequency that could affect stability.Among the rest of possibilities of group E,the impulse invari-ant method is also quite unfavorable:it provides much less gain after the resonant peak than the rest of the discretizations.This fact causes that the zero phase provided by R2(z)forω>ωo has much less impact on the global transfer function H VPI(z), in comparison to the phase delay introduced by R1(z).In this manner,the phase response of H VPI(z)would show a larger phase lag if R2(s)is discretized with impulse invariant instead of other methods,worsening the stability atω>ωo. Actually,as shown in Fig.5(b),if R2imp(z)is used,the delay of H VPI(z)can become close to−45◦for certain frequencies, which is certainly not negligible.This is illustrated,as an exam-ple,in Fig.5(b),in which Bode plot of H VPI(z)is shown when it is implemented as R1imp(z),and R2(s)is discretized with the different methods.Fixed values of K I and K P have been employed to make the comparison possible.K I=K P R f/L f has been chosen,so the cross coupling due to the plant is can-celed[16],[17],and an arbitrary value of1has been assigned to K P as an example.According to the real parameters of the laboratory prototype,L f=5mH and R f=0.5Ω.If the ra-tio K I/K P is changed,the differences will become more or less notable,but essentially,each method will still affect in the same manner.It should be remarked that the phase responseFig.5.Study of group E discretizations effect on R2(s)zeros.(a)Frequency response of R2(s)discretized with group E methods for a seventh harmonic resonant controller at f s=10kHz.(b)Frequency response of H V P I(z)for a third harmonic resonant controller at f s=10kHz,with R1im p(z),when R2(s) is discretized by each method of group E.K P=1and K I=K P R f/L f, with R f=0.5Ωand L f=5mH.of H VPI(z)atω≈ωo is not modiﬁed by R1imp(z),but only by the discretization of R2(s).Fig.5(b)also shows that some implementations introduce less phase at low frequencies than H VPI(s),but the inﬂuence of this aspect on the performance can be neglected.In conclusion,any of the discretization methods of group E, with the exception of impulse invariant and ZOH,are adequate for the implementation of R2(z).Actually,the inﬂuence of these two methods is so negative that they could easily lead to instability continuous resonant controllers with considerable stability margins.3)Displacement of Zeros by Group D Discretizations: Fig.6(a)shows the Bode plot of R1(s)implemented with setD schemes.R1f&b (z)produces a phase lead in comparisontoFig.6.Frequency response of R1(s)and H V P I(s)implemented with groupD methods for a seventh harmonic resonant controller at f s=10kHz.(a)R1(s).(b)H V P I(s),K P=1,and K I=K P R f/L f,with R f=0.5Ωand L f=5mH.R1(s),whereas R1b&b(z)causes a phase lag.This is also trueatω≈ωo,which are the most critical frequencies.Therefore,R1f&b(z)is preferable to R1b&b(z).On the other hand,as can beappreciated in Fig.6(b),the Bode plot of H VPIf&b(z)and H VPIb&b(z)scarcely differ.They both achieve an accurate reproduction ofH VPI(s)frequency response.Actually,atω≈ωo,they provideexactly the same phase.Consequently,they can be indistinctlyemployed with satisfactory results.IV.D ISCRETIZATION I NFLUENCE ON C OMPUTATIONALD ELAY C OMPENSATIONA.Delay Compensation in the Continuous DomainFor large values ofωo,the delay caused by T s affects the sys-tem performance and may cause instability.Therefore,a delaycompensation scheme should be implemented[14],[15],[17], [23],[27].1)Delay Compensation for H PR(s):Concerning resonant controllers based on the form H PR(s),a proposal was posed in[15]for performing the compensation of the computational delay.The resulting transfer function can be expressed in the s domain asH PR d(s)=K P+K I s cos(ωo NT s)−ωo sin(ωo NT s)s2+ω2o=K P+K I R1d(4) with N being the number of sampling periods to be compen-sated.According to the work of Limongi et al.[23],N=2is the most optimum value.2)Delay Compensation for H VPI(s):Because of H VPI(s) superior stability,it only requires computational delay for much greater resonant frequencies than H PR(s)[16],[17],[23]. Delay compensation could be obtained by selecting K P= cos(ωo NT s)and K I=−ωo sin(ωo NT s).However,this ap-proach would not permit to choose the parameters so as to satisfy K I/K P=R f/L f;thus,it would not cancel the cross coupling terms as proposed in[16]and[17].Therefore,an alternative approach is proposed shortly. R1d(s)and R2d(s)are individually implemented with a de-lay compensation of N samples each,so K P and K I can be still adjusted in order to cancel the plant pole:H VPI d(s)=K P s2cos(ωo NT s)−sωo sin(ωo NT s)s2+ω2o+K I s cos(ωo NT s)−ωo sin(ωo NT s)s2+ω2o=K P R2d+K I R1d.(5)3)Delay Compensation for R1d(s)and R2d(s):If the res-onant terms are decomposed by the use of two integrators,it is possible to perform the delay compensation by means of the block diagrams depicted in Fig.7(a)and(b)for R1d(s)and R2d(s),respectively.Fig.8illustrates the effect of the computational delay com-pensation for both R1d(s)and R2d(s),setting f o=350Hz and f s=10kHz as an example.As N increases,the180◦phase shift at f o rises,compensating the phase lag that would be caused by the delay.B.Discrete-Time Implementations of Delay Compensation SchemesAs stated in the previous section,the delay compensation should be implemented for each resonant term separately.For this reason,it is convenient to study how each discretization method affects the effectiveness of the delay compensation for R1d(z)and R2d(z)individually.Effects on groups E and D implementations,due to their superior performance,are ana-lyzed.Tables V and VI reﬂect the discrete transfer functions obtained by the application of these methods to R1d(s)andR2d(s),respectively.R1df&b (z)and R1db&b(z)result of apply-ing the corresponding discretization transforms to theschemeFig.7.Implementations of(a)R1d(s)and(b)R2d(s)based on twointegrators.Fig.8.Frequency response of(a)R1d(s)and(b)R2d(s)for different valuesof N;f o=350Hz and f s=10kHz.shown in Fig.7(a).On the other hand,R2df&b(z)and R2db&b(z) are obtained by discretizing the integrators shown in Fig.7(b).Substituting N=0in Tables V and VI leads to the expres-sions of Tables II and III,respectively.It can be also noted that。

结肠慢传输型便秘诊治进展

结肠慢传输型便秘诊治进展目前，我们将慢性顽固性便秘（chronic intractable consti-pation）已专指慢性传输便秘（slowtransit constipation，STC），又称结肠无力（colonic interia），是因结肠传输功能减弱致使肠内容物滞留于结肠而引起的顽固性便秘。

病因不清，症状顽固，治疗困难。

1986年由Preton和Lennard-Joners提出，临床以结肠通过时间延长和对纤维素、缓泻剂治疗反应差为特征［1］。

Surrenti 报道慢传输型便秘占慢性便秘的37% ［2］，Johanson报告美国家庭慢性传输性便秘的发病率为3% ［3］。

Degen认为终末大便形态与结肠通过时间有关［4］。

Probert提出根据排便次数、大便形态评分和大便间隔时间，可应用公式对全胃肠通过时间进行粗略估算［5］。

如何评价便秘的诊断与治疗，是当今一个重要的课题。

1 病理生理学慢性传输型便秘是由结肠无效推进所致的一系列以便秘为主要表现的临床综合征。

结肠的运动无一定规律，乙状结肠内压力显著高于横结肠和降结肠。

健康人的结肠运动方式有传送性收缩和非传送性收缩2种，仅有1/3的传送性收缩可对肠内容物发挥推进作用［6］。

对结肠推进有意义的是收缩幅度≥75～90mmHg的高幅推进性收缩（high-amplitude propagated contractions，HAPCs），常发生于白天进食后，可能与结肠的激发排便有关。

结肠是否对进餐和HAPCs产生有效的收缩性反应是鉴别正常人和便秘患者的关键。

近年来发现部分STC患者存在全胃肠道动力异常，如Gunay等用同位素方法检测研究发现44.4%的STC有胆囊功能障碍，50%胃排空及小肠转运功能减慢。

多数人认为是自主神经支配机制紊乱导致STC。

Automare等用乙酰胆碱汗点实验自主神经完整性证实，便秘患者存在自主神经功能异常。

研究还证明，便秘与心理因素相关。

溃疡性结肠炎的结肠动力异常

【摘要】结肠动力异常是引起溃疡性结肠炎患者腹泻的一个重要因素。

在溃疡性结肠炎活动期，结肠表现出收缩力下降和传输时间延长。

在结肠的不同部位，其传输时间并不相同。

缓解期溃疡性结肠炎患者的结肠动力明显好转，但仍未完全恢复正常。

引起溃疡性结肠炎结肠动力异常的机制尚不完全明确，除了炎症反应以外，肠神经系统的异常也参与其中。

【关键词】溃疡性结肠炎；结肠动力；结肠传输；肠神经系统DOI：10.3760/cma.j.issn.2096⁃367X.2019.03.016Colonic motility disorders in ulcerative colitisDong Jinpei，Zhang Lu，He Shengduo，Zhang Shangqing，Wang HuahongDepartment of Gastroenterology，Peking University First Hospital，Beijing100034，ChinaCorresponding author：Wang Huahong，Email：wwwanghuahong @，Tel：0086⁃10⁃83572226【Abstract】Colonic motility disorder is an important pathogenesis of diarrhea in ulcerative colitis patients.In the active stage of ulcerative colitis，colon appears decreased contractility and prolonged transmission time.The transmission time various in different colon segments.Colonic motility in remission of ulcerative colitis improves apparently，but not returns to normal.The pathogenesis of colonic motility disorder is unclear，inflammation and dysfunction of enteric nervous system may be involved in this process.【Key words】Ulcerative colitis；Colonic motility；Colonic transit；Enteric nervous systemDOI：10.3760/cma.j.issn.2096⁃367X.2019.03.016溃疡性结肠炎（ulcerative colitis，UC）是一种病因尚不清楚的慢性非特异性肠道炎性疾病，表现为腹痛、腹泻和黏液血便。

幅度相位误差纠正英语

幅度相位误差纠正英语Amplitude and Phase Error Correction.Amplitude and phase error correction are crucial processes in various fields of engineering and science, particularly in signal processing, telecommunications, and electronics. These errors can arise due to various factors such as noise, distortion, and equipment limitations. Correcting these errors is essential to ensure accurate signal transmission and reliable system performance.Amplitude error correction involves adjusting the amplitude of a signal to its original or desired value. Amplitude errors can occur due to attenuation, gain variations, and nonlinearities in the signal path. To correct these errors, various techniques can be employed, such as gain adjustment, equalization, and amplitude compensation. Gain adjustment involves adjusting the amplifier gain to compensate for attenuation losses. Equalization techniques are used to correct frequency-dependent amplitude distortion by adjusting the signal's amplitude at different frequencies. Amplitude compensation techniques involve adjusting the signal amplitude to compensate for nonlinearities in the system.Phase error correction, on the other hand, involves adjusting the phase of a signal to align it with its original or desired phase. Phase errors can occur due to propagation delays, phase shifts introduced by components, and phase noise. To correct these errors, various phase correction techniques can be applied, such as phase alignment, phase compensation, and phase locking. Phase alignment techniques involve adjusting the phase of the signal to align it with a reference signal. Phase compensation techniques involve introducing an opposite phase shift to cancel out the phase error introduced by the system. Phase locking techniques involve using a phase-locked loop (PLL) or a frequency-locked loop (FLL) to maintain a constant phase relationship between two signals.Amplitude and phase error correction techniques are widely used in various applications, such as radar systems,satellite communications, audio processing, and image processing. In radar systems, amplitude and phasecorrection techniques are essential for accurate target detection and ranging. In satellite communications, these techniques are crucial for reliable signal transmission over long distances. In audio processing, amplitude and phase correction can improve sound quality by reducing distortion and noise. In image processing, these techniques can enhance image clarity and resolution by correcting distortions introduced during image acquisition or transmission.In conclusion, amplitude and phase error correction are fundamental processes in signal processing and system design. Correcting these errors is essential to ensure accurate signal transmission and reliable system performance. Various techniques and algorithms have been developed to address amplitude and phase errors, and their application depends on the specific application and system requirements. As technology continues to evolve, so will the methods and techniques used for amplitude and phaseerror correction, leading to more efficient and accurate systems in the future.。

ACOUSTIC MEMBRANE OSCILLATOR

专利名称：ACOUSTIC MEMBRANE OSCILLATOR发明人：CHARBONNEAUX, Marc,PERRICHON, Claude, Annie,PICCALUGA, Pierre申请号：FR1999000255申请日：19990205公开号：WO99/041942P1公开日：19990819专利内容由知识产权出版社提供摘要：In the field of sound reproduction the movements of membranes have too high amplitude levels relative to the deformations of low amplitude levels propagated on the surfaces of materials of the objects which have emitted the original sounds. To remedy this, the method proposed by the invention is not concerned with higher sound efficiency, but it rather aims at exciting the particles of air by adding at least a membrane vibrating without emitting audible sound in the ambient medium. The method is characterised it that it uses low acoustic efficiency with high oscillating dissipation of low amplitude in the membrane material, and air, it is a physiological electro-acoustic transducer of air. An apparatus is produced characterised in that it has no magnet. Said physiological electro-acoustic transducer of air does not require yokes, membrane suspensions. The coil is electrically coupled to a sound reproducing system. The assembly is mounted fixed so as to vibrate freely on a support. Said method and apparatus are applicable in all types of sound, audio and audio-visual reproduction.申请人：CHARBONNEAUX, Marc,PERRICHON, Claude, Annie,PICCALUGA, Pierre地址：6, rue Dumange F-69004 Lyon FR,6, rue des Escoffiers F-38080 L'Isle d'Abeau FR,6, rue des Escoffiers F-38080 L'Isle d'Abeau FR国籍：FR,FR,FR代理机构：CHARBONNEAUX, Marc 更多信息请下载全文后查看。

Instruction manual.pdf_1693225695.0476854说明书

I. M EASUREMENT OF DC AND AC VOLTAGE AND CURRENT , MEASUREMENTUNCERTAINTY AND ERRORS.M ESUREMENT OF THE PARAMETERS OF DIODES ANDTRANSISTORSTheory:Theory of errors and uncertainty in the measurement. Uncertainty of type A ,type B and C. Definitions of the instrument precision by the producers. Principle of multimeters. Measurement of DC and AC voltage and current. Connection of the multimeter to the tested circuit. Measurement of the effective value of the voltage and current- definitions & principles. Measurement of the effective value alternating voltage/current with or without superimposed direct voltage/current. Shape coefficient, crest factor. Testing of diodes and transistors using the multimeter Principle of the digital frequency measurement. Exercises:1) Get acquainted with Agilent 33220A waveform generator. Set the appropriate load value according tothe resistor used (Utility > Output Setup> Load> 50Ω). ATTENTION: The generator output must be matched to the load impedance for all laboratory tasks.2) Set the generator for harmonic signal output of 2Vpp amplitude and 100 Hz frequency (setting of thegenerator, not measured value on the voltmeter). Connect the rectifier to loaded output according to the schematic. Measure the rectified voltage by available multimeters (using DC mode). Read at least10 measured values. Estimate measurement uncertainty of type A. Estimate the measurementuncertainty of type B based by parameters from datasheets. Determine overall uncertainty of your measurements (type).3) Generate a harmonic, rectangular, triangular, saw tooth and at least one of embedded arbitrarysignals with arbitrary amplitude from the range 1-5 V and frequency from the range 50-300 Hz with the offset equal to zero. Measure voltages for all shapes using both a TRMS voltmeter and simple multimeter with diode rectifier. Explain why the multimeter readings differ for every waveform and amplitude. Use a multimeter also for frequency measurement of every waveform.4) Repeat task 3 for harmonic, rectangular, triangular, saw-tooth waveform with DC offset set to 1V.Measure the output voltage of the generator by TRMS voltmeter in both AC and DC mode. What is the total dissipated power on the resistor load and what is the effective value of the voltage? Hint -Parceval´s theorem.5) Generate a harmonic signal with amplitude 1V and frequency of 5Hz. What is measured by themultimeter? Gradually adjust the frequency 10, 50, 200, 1k, 10k, 25k, 100k, 500kHz and 1MHz. What is measured by the multimeter? Try to explain the multimeter behavior.6) Set the generator for rectangular pulses of 100 Hz repeating frequency and pulse width of 100 s. Setthe low voltage level to 0V. The high level (pulse amplitude) set gradually to 0.02V, 0.2V, 2V. How does the measured rms value change for different peak values of the signal? What voltage value is shown by the multimeter? Is its variation consistent with the changes of the pulse amplitude?Compare your measurement results acquired with other types of multimeters.7) Repeat task 4 for AC and DC current through the load. How can you calculate total power dissipatedon the resistor load from the measured current and resistor’s value? Compare results with those of the task 4.8) Test available diodes using a multimeter and assess whether they passed. What does thismeasurement tell us about the measured diode? Measure also the Graetz bridge9) Measure PN junctions and h21E of available transistors in the active and inverse mode. Comparemeasured results with datasheet values.10) Switch the multimeter to frequency measurement mode. Set the generator to an arbitrary harmonicwaveform of frequency within kHz range. Gradually rise the amplitude from minimum up to 5V.Observe the measured frequency and determine an amplitude threshold, where multimeter starts to measure correctly. Try to explain the results and behavior of the multimerter in frequency measurement mode.Instruments‘ manuals:Multimeter UT 803Multimeter Agilent 34410AMultimeter Agilent 34405AMultimeter Metex 3640Multimeter METEX 3850DGenerator Agilent 33220AStudy materials:Agilent multimeter simulation installation filesWebsite simulating the function of selected instruments - meas-lab.fei.tuke.sk。

考研英语作文中表明显示的高级表达

考研英语作文中表明显示的高级表达In the field of postgraduate entrance examination, especially in the English writing test, it is essential for candidates to express their ideas clearly and logically. As a result, advanced expressions for indicating or showing are crucial for enhancing the quality of the composition. In this article, we will discuss some high-level expressions that can be used in the context of showing or indicating in the writing of the postgraduate English examination.To begin with, one effective way to indicate or show a point is to use the phrase "it is evident that." This phrase emphasizes the clarity and obviousness of the point being made, making the argument more persuasive and convincing. For example, "It is evident that the rapid development of technology has greatly impacted people's daily lives."Another useful expression to show a relationship between two ideas is "this demonstrates that." This phrase is often used to illustrate a cause-and-effect relationship or to show evidence supporting a claim. For instance, "This demonstrates that environmental protection should be a top priority for governments around the world."Furthermore, the phrase "it is clear that" can also be employed to indicate a point in a straightforward and direct manner. This expression is useful for emphasizing the clarity and certainty of a statement. For example, "It is clear that education plays a crucial role in reducing poverty and inequality."Additionally, the use of the phrase "it is apparent that" can help to highlight the obviousness of a point or situation. This phrase is particularly effective when emphasizing the visibility or significance of a particular aspect. For instance, "It is apparent that social media has revolutionized the way we communicate with each other."Moreover, the expression "it is manifest that" can be utilized to emphasize the clarity and visibility of a concept or idea. This phrase is often used to highlight the obviousness or self-evident nature of a point. For example, "It is manifest that diversity in the workplace leads to greater innovation and productivity."In conclusion, the use of advanced expressions for indicating or showing in the context of postgraduate English writing is crucial for effectively conveying ideas and arguments. By incorporating phrases such as "it is evident that," "this demonstrates that," "it is clear that," "it is apparent that," and "it is manifest that," candidates can enhance the quality andpersuasiveness of their compositions. It is important for candidates to practice using these expressions in their writing in order to improve their overall performance in the English writing test of the postgraduate entrance examination.。

HD49334AF资料

+ −
BIAS
Байду номын сангаасBIAS
AVDD
Rev.1.0, Apr 20, 2004, page 4 of 22
HD49334AF/AHF
Block Diagram
DRDVDD ADCLK SPBLK SPSIG DVDD AVDD DVSS AVSS
16 18 19 Timing generator
31 16 18 19 19 42 OEB
Bias generator
17
44 45 43
35
32 34 33
VRT
VRM
SDATA
Rev.1.0, Apr 20, 2004, page 5 of 22
BIAS
OBP
VRB
SCK
CS
Output latch circuit
HD49334AF/AHF
Internal Functions
Functional Description • CDS input CCD low-frequency noise is suppressed by CDS (correlated double sampling). The signal level is clamped at 14 LSB to 76 LSB by resister during the OB period. *1 Gain can be adjusted using 8 bits of register (0.132 dB steps) within the range from –2.36 dB to 31.40 dB. *2 • ADC input The center level of the input signal is clamped at 512 LSB (Typ). Gain can be adjusted using 8 bits of register (0.01784 times steps) within the range from 0.57 times (–4.86 dB) to 5.14 times (14.22 dB). *1 • Automatic offset calibration of PGA and ADC • DC offset compensation feedback for CCD and CDS • Pre-blanking CDS input operation is protected by separating it from the large input signal. Digital output is set at clamp level by resister. • Digital output enable function Notes: 1. It is not covered by warranty when 14LSB settings 2. Full-scale digital output is defined as 0 dB (one time) when 1 V is input. Operating Description Figure 1 shows CDS/PGA + ADC function block.

高级阅读No Signposts in the Sea海上无航标原文+翻译+生词注解+修辞赏析

No Signposts in the Sea一、In the dining-saloon I sit at a table with three other men, Laura sits some way off with a married couple and their daughter. I can observe her without her knowing and this gives me pleasure, for it is as in a moving picture that I can note the grace of her gestures,①whether she raises a glass of wine to her lips or②turns with a remark to one of her neighbors or takes a cigarette from her casewith those slender fingers（loose sentense松散句）.I have never had much of an eye for noticing the clothes of women, but I get the impression Laura is always in grey and white by day, looking cool when other people are flushed and shinyin the tropical heat; in the evening, she wears soft rich colors（metonymy借代）, dark red, olive green, midnight blue, always of the most supple flowing texture. I ventured to say something of the kind to her, when she laughed at my clumsy compliment and said. I had better take to writing fashion articles instead of political leaders.PS：①soft：ADJ Something that is soft is very gentle and has no force. For example, a soft sound or voice is quiet and not harsh. A soft light or colour is pleasant to look at because it is not bright. (声音、光线或色彩) 柔和的；②rich:Rich smells are strong and very pleasant. Rich colours and sounds are deep and very pleasant. 浓郁的(气味); 浓厚的(色彩)因此soft和rich这里的修辞手法是：oxymoron n. （修词中的）矛盾修饰法在餐厅里我同另外三个男人围坐在一张桌子旁，而劳拉同一对夫妇及他们的女儿一块儿坐在离我不远的地方。

通信专业英语-Lesson 6

The bandwidth (B) of an AM DSBFC wave is equal to the difference between the highest upper side frequency and the lowest side frequency, or two times the highest modulating signal frequency .
Technique
Introduction
Q1: What is the purpose of modulation? Q2: What is modulation? Q3: What is demodulation? Q4: What is AM? Q5: What applications can the AM be used for? Q6: What is the construction of the AM modulator? Q7: What’s the definition of RF?
Lesson 6 Amplitude Modulation Transmission
6.1 Principles of Amplitude Modulation 6.2 Single-side Band Modulation 6.3 Generation of SSB Waves 6.4 Weaver SSB Frequency Shifting
直译与意译
This means that insofar as the transmission of information is concerned, only one sideband is necessary, and if the carrier and the other sidebands are suppressed at the transmitter, no information is lost.