Cross-Slot-Coupled Microstrip Antenna and Dielectric Resonator Antenna for Circular Polarization

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MICROSTRIP ANTENNA

申请人：NIPPON TELEGRAPH & T由知识产权出版社提供
专利名称：MICROSTRIP ANTENNA 发明人：TAGA TOKIO 申请号：J P 4 4 05680 申请日：198004 05 公开号：J P S604 0203B 2 公开日：19850910
摘要：PURPOSE:To realize two input/output terminals which resonate in frequencies being mutually different, by providing feeding points on both the long axis and the short axis of an elliptical radiation conductor element, in the microstrip antenna which is used for the mobile communication system, etc. CONSTITUTION:In the structure which consists of a radiation conductor element 1, a ground conductor 2 and a dielectric 3, and is coupled with coaxial lines 8, 12 on the reverse side of the conductors 2, an elliptical radiation conductor plate is used for the element 1. Feeding points 6, 10 are provided on both the long axis 4 and the short axis 5 of this element, and two energizing modes falling at right angles with each other are energized independently. As a result, it is possible to obtain two input/output terminals 9, 13 which are operated in different frequencies. Also, the energizing except a necessary mode is suppressed by shortcircuiting the element 1 and the conductor 2 in the intersection point of the long axis 4 and the short axis 5 of the element 1. In this way, it is possible to obtain two input/output terminals which are small-sized, light in weight and low in attitude, and resonate in frequencies being mutually different.

Glossary of Satellite Technology 卫星技术词汇总结

天线增益antenna gain等向天线isotropic antenna天线极化antenna polarization双极化缝隙耦合微带天线dual-polarized slot-coupled microstrip antenna线极化linear polarization交叉极化cross polarization交叉极化抑制cross-polarization suppression隔离度isolation极化隔离度polarization isolation口径aperture面天线aperture antenna阵列天线array antenna跟踪精度tracking accuracy侧馈偏置卡塞格伦(SFOC)天线Side-Fed Offset Cassegrain (SFOC) antenna偏馈抛物面天线offset fed paraboloidal antenna副瓣side lobe前轴向区forward axis天线座antenna pedestal俯仰elevation方位azimuth轮轨式俯仰-方位型天线座roller raceway type elevation-azimuth antenna pedestal便携移动卫星站portable mobile satellite station无线图传系统wireless image transmission system频谱分析仪spectrum analyzer光端机optical transceiver功分器power analyzer光波导功分器optical waveguide power analyzer异构卫星通信系统heterogeneous satellite communication system（网络）互联interconnect（业务）互通interworking公用电信网间互联管理规定Administration of the Interconnection of Public Telecommunications Networks Provisions数据格式data format通信协议communication protocol网络融合network convergence应用层application layer无缝连接seamless connection通信接口communication interface用户网络接口user to network interface （UNI）网络节点接口network to network interface（NNI）光纤技术optical fiber technology固网和移动网的融合fixed and mobile network convergence （FMC）车载无线接入网络vehicle-mounted/car-borne wireless network专用接入网关private access gateway互联网关internet gateway多模用户终端multi-module user terminal工作频段working band组网方式network formation星状star网状net点对点point-to-point多址multi-address模拟用户线接口simulated/analogue user interface模拟中继线接口simulated/analogue trunk interface数字中继接口data truck interface地面站ground station地面信关站ground gateway station远端站remote station地球站terrestrial station公共交换电话网public switched telephone network（PSTN）公共交换数据网public switched data network（PSDN）协议数据单元protocol data unit（PDU）无线收发系统wireless transceiver system专用集成电路application-specific integrated circuit（ASIC）协议转换器protocol switcher协议封装与转换protocol encapsulating and switching多协议封装multi-protocol encapsulating（MPE）系统同步system synchronization移动分组数据服务mobile paneled data service（MPDS）前向链路forward link反向链路reverse link备份信道backup channelIP封装机IP encapsulator协议加速器protocol accelerator波段天线band antenna船载天线ship-borne antenna广播链路broadcasting上行站uplink station射频单元frequency unit射频前端RF front end调制器modulator复用器multiplexer频谱资源frequency resource广域覆盖wide area coverage海事终端maritime terminal航空终端aeronautic terminal卫星基站satellite base station（SBS）本土地面站HLES用户鉴权user authentification组播multicasting内部路由表internal routing table子网subnet天线切换器antenna switcher电视解码器TV decoder集线器concentrator数据交换机data switching exchange无线接入点wireless access point主控计算机host computer美军转型卫星通信系统TSAT美军单向广播系统GBS全球信息栅格（美国）Global Information Grid（GIG）移动用户目标系统mobile user objective system（MUOS）宽带全球卫星通信WGS网络战net-centric warfare出向信号outcoming signal入向信号incoming signal输入流input stream封包检测null-packet detection循环校验码CRC微调电容器padder导引插入pilot insertion比特交织bit interleaver正交调制quadrature modulationIP路由IP routing服务接入点SAP卫星链路控制层SLC卫星媒体访问控制层MAC卫星物理层SPHY透明转发器transparent transmitter再生式转发器regenerated transmitter卫星回传信道satellite return channel混合回传信道hybrid return channel拓扑topology点波束spot beam馈电链路feeder link协议栈protocol stack应用场景application scenario地面中继器terrestrial repeater卫星中枢satellite hub单点回传信道unicast return channel卷积码convolutional code基带成型滤波器baseband shaping filter卫星信道适配器satellite channel adapter分组协议packet protocol核心网core network宽带IP卫星技术broadband IP satellite technology天基网络space-based network定长fixed length非定长unfixed lengthTS（时间戳）数据包报头TS packet headerIP datagram IP数据包选路协议routing protocol数据流分类data stream classification通路routing数据时延限制data throughput limit拥塞控制算法congestion control algorithm互联网骨干网连接internet backbone interconnection报文丢失packet loss美国航空航天局NASA欧洲空间局ESA空间数据系统咨询委员会CCSDS潜伏/等待时间latency上链uplink前链forward link下链downlink回链return link多址分配协议multi-address allocation protocol双跳double hop卫星载荷satellite payload转型通信卫星transformational communications satellite (TSAT)比特流bit stream链路层帧link layer frame多点广播路由multicast routing机载路由器space-born/onboard router锚站anchor station多波束天线multi-beam antenna (MBA)室内单元indoor unit (IDU)室外单元outdoor unit（ODU）太比特terabyte流量控制traffic shaping公平接入政策fair access policy（FAP）流媒体视频streaming video对等互联网peer to peer (P2P)信道容量channel capacity低轨道星座通信卫星low-earth orbit （LEO）constellation communication satellite 轨道控制orbit control同步轨道通信卫星geosynchronous communication satellite星上处理onboard processing星间链路inter-satellite link关口站gateway station动态分配信道dynamic channel allocation（DCA）地面蜂窝移动通信系统ground cellular mobile communication system 静止轨道卫星geostationary （GEO）satellite全球波束global beam宽点波束wide spot beam窄点波束narrow spot beam转发器transmitter单向太阳能帆板翼展用户链路user link馈线链路feeder link数字波束成形digital beam forming（DGF）宽带码分多址WCDMA有效载荷分析payload analysis多波束馈源multi-beam feed超短波语音通信ultra-short wave audio communication相控阵天线phased array antenna交叉振子crossed dipole微带贴片microstrip patch介质螺旋medium spiral赋形forming低剖面天线low profile antenna通量密度intensity可编程逻辑器件programmable logic device（PLD）射频单元RF unit中频单元IF unit电源模块power module射频大规模集成电路large scale RF integrated circuit调制解调modulation and demodulation信道编码channel coding信噪比signal to noise ratio（SNR）无码间干扰和抖动的交错正交相移键控IJF-OQPSK载波恢复carrier recovery训练序列training sequence门限threshold射频前端电路RF front-end circuit射频功放RF PA双工器diplexer前端滤波器front-end filter介质加载medium loading四臂螺旋耦合天线quadrifilar helical coupling antenna频率复用frequency reuse中继卫星relay satellite帧结构frame structure时隙time slot亚洲蜂窝卫星系统Asia Cellular Satellite（ACeS）星上交换onboard switching单跳通信single-hop communication射频接口RF interface板到板接口board-to-board（B2B）interface宽带全球局域网系统broadband global area network（BGAN）载波带宽carrier bandwidth地基波速形成GBBF轨位orbital slot；orbital position对地观测载荷earth observation load单片微波集成电路MMIC收发组件T/F component过顶卫星overhead satellite多波束天线multi-beam antenna等通量equivalent flux天线阵元antenna array element交织/解交织interleaving/deinterleaving扩频/解扩spread spectrum/de-spread spectrum低复杂度并行卷积正交调制编码low-complexity parallel convolution orthogonal/quadrature modulation coding 大频偏快速码捕获fast code acquisition under large frequency offset协同分集cooperative diversity功率控制power control易遮蔽occlusion-prone多阴影衰落multi-shadow fading雨衰rain fading电平余量（level）margin算法algorithm拓扑快照静态路由TSSR定长包格式fixed length packet format离散化discretization静态拓扑stationary topology动态拓扑dynamic topology链路代价link cost路由表routing table卫星路由算法satellite routing algorithm卫星交换节点satellite switching node卫星接入节点satellite access node空中接口技术air interface technology频分多址frequency division multiple access（FDMA）码分多址code division multiple access（CDMA）频分复用frequency division multiplexing非相干测量non-coherent measuring时间同步精度time synchronization accuracy短稳short term stability蜂窝小区cellular波束小区beam cell导频强度pilot frequency intensity时间基准time reference单星定位single satellite positioning伪距测量pseudo range measurement多普勒测量Doppler measurement伪距方程pseudo range equation多普勒方程Doppler equation高程方程vertical equation三维坐标tri-dimensional coordinate钟差clock bias频差frequency bias老化率aging rate迭代加权最小二乘法iterative weighted least-square equation卡尔曼滤波法Kalman filter equation高斯分布Gaussian distribution自适应编码和调制self-adaptive coding and modulating (ACM)远端站remote station上行功率控制UPC向前纠错偏码率FECmesh路由器mesh router动中通communication on the move （CoTM）静中通stationary satellite communication system海事卫星通信maritime satellite communication低抖动low jitter移动回传mobile backhaul蜂窝回传cellular backhaul多路接入专网private network with multiple access多协议标签交换MPLS天地一体网络integrated space-ground network卫星应用系统集成satellite application system integration机动遥感卫星mobile remote sensing satellite遥感卫星通用变频器universal converter for remote sensing satellite 合成孔径雷达synthetic aperture radar环境减灾卫星satellite for environmental and disaster monitoring卫星导航终端satellite navigation terminal星载时差测量仪onboard time difference measurer高可靠星载接收机onboard receiver of high reliability三模接收机tri-modulation receiver抗干扰通信技术anti-jamming communication technology高动态抗干扰数字图像传输系统high mobility anti-jamming digital image transmission system 数字式末制导图像传输设备digital terminal guidance image transmission device基于巡航导弹的卫星cruise-missile-based satellite捕控指挥链catching and control chain of command超视距信息链beyond-visual-range information chain近程无人机near-distance pilotless plane超近程无人机ultra-near-distance pilotless plane巡航导弹cruise missile开路图像传输open circuit image transmission增补转发器additional transponder虚拟专网virtual private network （VPN）预提取pre-fetching旁瓣sidelobe在轨卫星satellite in orbit三轴稳定three-axis stability在轨寿命orbital lifetime在轨精度orbit precision转发器冗余配置transponder redundant configuration行波管放大器功率traveling wave tube amplifier power有效全向辐射功率EIRP饱和通量密度saturation flux density（SFD）信标频率beacon frequency砷化镓太阳能单元gallium arsenide solar power unit光电转换效率photoelectric conversion efficiency液体二元推进系统liquid bipropellant propulsion system氙离子推进系统xenon ion propulsion system质子火箭proton rock线性度linearity相位噪声phase noise地球站earth station在轨备份on-orbit backup“动中通”天线COTM antenna扩谱spread spectrum蜂窝架构cellular architecture固定天线fixed antenna低轮廓天线low profile antenna缝隙反射面slotted reflector全向控阵omnidirectional phased array非实时数据交互delayed data interaction数字集群digital trunking多载波跳频multi-carrier frequency hopping自动上行功率控制ULPC时槽分配time slot allocation纠错编码correction coding地址映射address mapping报头压缩header compression信道加密channel security以太网接口Ethernet interface调制器输出modulator output解调器输入demodulator input【我们从一个会场到另一个会场，les voyageurs du temps，我在同传箱里悄悄看你的侧脸，温暖和笃定的表情，多么好。

一种基于开口谐振环的高增益宽带双极化天线设计

一种基于开口谐振环的高增益宽带双极化天线设计任宇辉;丁君;郭陈江【摘要】A wideband and high gain dual-polarized antenna based on split ring resonators is presented. The antenna is composed of two cross printed dipole antennas vertically fixed on an aluminum plate as ground plane, which are excited by two similar micro-strip baluns. In order to further improve the gain with a broad bandwidth, split ring resonators and complementary split ring resonators are loaded on the printed dipole antennas. Measured results show that the proposed antenna achieves-10 dB return loss with bandwidth of 0.98~2.01 GHz (69%), and its port isolation is higher than 20 dB within that band. The maximum gain of antenna is improved up by 4.1 dB because of split ring resonators. Furthermore, the height of the antenna is reduced about 12% than the LPPDs- DPA.%该文设计并制作了一种基于超材料的高增益宽带双极化天线,该天线由两个正交放置的印刷振子单元、馈电巴伦及金属接地板构成.为了进一步展宽带宽、提高增益,在天线上了加载开口谐振环、互补开口谐振环等超材料结构.测试结果表明,该天线回波损耗小于-10 dB的带宽约为69%(0.98~2.01 GHz),在相同的频带内隔离度大于20 dB.由于开口谐振环的引向作用,天线的辐射特性得到改善,增益最大提高了4.1 dB左右.和已有设计相比,该天线的总体高度减小了约12%.其可以当作独立天线使用,也可用作反射面天线的双极化馈源.【期刊名称】《电子与信息学报》【年(卷),期】2017(039)011【总页数】5页(P2790-2794)【关键词】开口谐振环;互补开口谐振环;双极化天线【作者】任宇辉;丁君;郭陈江【作者单位】西北工业大学电子信息学院西安 710129;西北大学信息科学与技术学院西安 710069;西北工业大学电子信息学院西安 710129;西北工业大学电子信息学院西安 710129【正文语种】中文【中图分类】TN823由于双极化天线(Dual-Polarized Antennas, DPAs)可以形成一对极化方式正交，且工作频率相同的电磁波，目前已广泛应用于电子信息领域。

微带耦合器的中英文对照翻译

Design and Analysis of Wideband Nonuniform Branch Line Coupler and Its Application in a Wideband Butler MatrixYuli K. Ningsih,1,2 M. Asvial,1 and E. T. RahardjoAntenna Propagation and Microwave Research Group (AMRG), Department of Electrical Engineering, Universitas Indonesia, New Campus UI, West Java, Depok 16424, Indonesia Department of Electrical Engineering, Trisakti University, Kyai Tapa, Grogol, West Jakarta 11440, IndonesiaReceived 10 August 2011; Accepted 2 December 2011Academic Editor: Tayeb A. DenwdnyCopyright © 2012 Yuli K. Ningsih et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.AbstractThis paper presents a novel wideband nonuniform branch line coupler. An exponential impedance taper is inserted, at the series arms of the branch line coupler, to enhance the bandwidth. The behavior of the nonuniform coupler was mathematically analyzed, and its design of scattering matrix was derived. For a return loss better than 10 dB, it achieved 61.1% bandwidth centered at 9GHz. Measured coupling magnitudes and phase exhibit good dispersive characteristic. For the 1dB magnitude difference and phase error within 3∘, it achieved 22.2% bandwidth centered at 9GHz. Furthermore, the novel branch line coupler was implemented for a wideband crossover. Crossover was constructed by cascading two wideband nonuniform branch line couplers. These components were employed to design a wideband Butler Matrix working at 9.4GHz. The measurement results show that the reflection coefficient between the output ports is better than 18dB across 8.0GHz–9.6GHz, and the overall phase error is less than 7.1. IntroductionRecently, a switched-beam antenna system has been widely used in numerous applications, such as in mobile communication system, satellite system, and modern multifunction radar. This is due to the ability of the switched-beam antenna to decrease the interference and to improve the quality of transmission and also to increase gain and diversity.The switched-beam system consists of a multibeam switching network and antenna array. The principle of a switched-beam is based on feeding a signal into an array of antenna with equal power and phase difference. Different structures of multibeam switching networks have been proposed, such as the Blass Matrix, the Nolen Matrix, the Rotman Lens, and the Butler Matrix .One of the most widely known multibeam switching networks with a linear antenna is the Butler Matrix. Indeed, it seems to be the most attractive option due to its design simplicity and low power loss .In general, the Butler Matrix is an N × N passive feeding network, composed of branch line coupler, crossover, and phase shifter. The bandwidth of the Butler Matrix is greatly dependent on the performance of the components. However, the Butler Matrix has a narrow bandwidth characteristic due to branch line coupler and crossover has a limited bandwidth.As there is an increased demand to provide high data throughput , it is essential that the Butler Matrix has to operate over a wide frequency band when used for angle diversity. Therefore, many papers have reported for the bandwidth enhancement of branch line coupler . In reference , design and realization of branch line coupler on multilayer microstrip structure was reported. These designs can achieve a wideband characteristic. However, the disadvantages of these designs are large in dimension and bulk.Reference introduces a compact coupler in an N-section tandem-connected structure. The design resulted in a wide bandwidth. Another design, two elliptically shaped microstrip lines which are broadside coupled through an elliptically shaped slot, was employed in . This design was used in a UWB coupler with high return loss and isolation. However, these designs require a more complex manufacturing.In this paper, nonuniform branch line coupler using exponential impedance taper is proposed which can enhance bandwidth and can be implemented for Butler Matrix, as shown in Figure 1. Moreover, it is a simple design without needs of using multilayer technology. This will lead in cost reduction and in design simplification.Figure 1:Geometry structure of a new nonuniform branch line coupler design with exponential impedance taper at the series arm.To design the new branch line coupler, firstly, the series arm’s impedance is modified. The shunt arm remains unchanged. Reduced of the width of the transmission line at this arm is desired by modifying the series arm. Next, by exponential impedance taper at the series arm, a good match over a high frequency can be achieved.2. Mathematical Analysis of Nonuniform Branch Line CouplerThe proposed nonuniform branch line coupler use λ/4 branches with impedance of 50Ω at the shunt arms and use the exponential impedance taper at the series arms, as shown in Figure 1. Since branch line coupler has a symmetric structure, the even-odd mode theory can be employed to analyze the nonuniform characteristics. The four ports can be simplified to a two-port problem in which the even and odd mode signals are fed to two collinear inputs [22]. Figure 2 shows the schematic of circuit the nonuniform branch line coupiers.Figure 2:Circuit of the nonuniform branch line coupler.The circuit of Figure 2 can be decomposed into the superposition of an even-mode excitation and an odd-mode excitation is shown in Figures and .Figure 3:Decomposition of the nonuniform branch line coupler into even and odd modes of excitation.The ABCD matrices of each mode can be expressed following . In the case of nonuniform branch line coupler, the matrices for the even and odd modes become:A branch line coupler has been designed based on the theory of small reflection, by the continuously tapered line with exponential tapers , as indicated in Figure 1, wherewhich determines the constant as:Useful conversions for two-port network parameters for the even and odd modes of S11and S21 can be defined as follows :whereSince the amplitude of the incident waves for these two ports are ±1/2, the amplitudes of the emerging wave at each port of the nonuniform branch line coupler can be expressed asParameters even and odd modes of S11 nonuniform branch line coupler can be expressed as and as follows:An ideal branch line coupler is designed to have zero reflection power and splits the input power in port 1 (P1) into equal powers in port 3 (P3) and port 4 (P4). Considering to , anumber of properties of the ideal branch line coupler maybe deduced from the symmetry and unitary properties of its scattering matrix. If the series and shunt arm are one-quarter wavelength, by using , resulted in S11 = 0.As both the even and odd modes of S11 are 0, the values of S11 and S21 are also 0. The magnitude of the signal at the coupled port is then the same as that of the input port.Calculating and under the same , the even and odd modes of S21 nonuniform branch line coupler will be expressed as follows inBased on ,S11 can be expressed as follows Following ,S41 nonuniform branch line coupler can be calculating as followsFrom this result, both S31 and S41 nonuniform branch line couplers have equal magnitudes of −3dB. Therefore, due to symmetry property, we also have thatS11=S22=S33=S44=0,S13=S31，S14=S41，S21=S34, and . Therefore, the nonuniform branch line coupler has the following scattering matrix in3. Fabrication and Measurement Result of Wideband Nonuniform Branch Line CouplerTo verify the equation, the nonuniform branch line coupler was implemented and its -parameter was measured. It was integrated on TLY substrate, which has a thickness of 1.57mm. Figure 4shows a photograph of a wideband nonuniform branch line coupler. Each branch at the series arm comprises an exponentially tapered microstrip line which transforms the impedance from ohms to ohms. This impedance transformation has been designed across a discrete step length mm.Figure 4:Photograph of a proposed nonuniform branch line coupler.Figure 5 shows the measured result frequency response of the novel nonuniform branch line coupler. For a return loss and isolation better than 10dB, it has a bandwidth of about 61.1%; it extends from 7 to 12.5GHz. In this bandwidth, the coupling ratio varies between 2.6 dB up to 5.1dB. If the coupling ratio is supposed approximately 3 ±1dB, the bandwidth of about 22.2% centered at 9GHz.Figure 5:Measurement result for nonuniform branch line coupler.As expected, the phase difference between port 3 (P3) and port 4 (P4) is 90°. At 9 GHz, thephases of and are 85.54° and 171°, respectively. These values differ from ideal value by 4.54°. The average phase error or phase unbalance between two branch line coupler outputs is about 3°. But even the phase varies with frequency; the phase difference is almost constant and very close to ideal value of 90° as shown in Figure 6.Figure 6:Phase characteristic of nonuniform branch line coupler.4. Design and Fabrication of the Wideband Butler MatrixFigure 7 shows the basic schematic of the Butler Matrix . Crossover also known as 0dB couplers is a four-port device and must provide for a very good matching and isolation, while the transmitted signal should not be affected. In order to achieve wideband characteristic crossover, this paper proposes the cascade of two nonuniform branch line couplers.Figure 7:Basic schematic of the Butler Matrix .Figure 8shows the microstrip layout of the optimized crossover. The crossover has a frequency bandwidth of 1.3GHz with VSWR = 2, which is about 22.2% of its centre frequency at 9 GHz. Thus, it is clear from these results that a nonuniform crossover fulfills most of the required specifications, as shown in Figure 9.Figure 8:Photograph of microstrip nonuniform crossover.Figure 9:Measurement result for nonuniform crossover.Figure 10 shows the layout of the proposed wideband Butler Matrix. This matrix uses wideband nonuniform branch line coupler, wideband nonuniform crossover, and phase-shift transmission lines.Figure 10:Final layout of the proposed wideband Butler Matrix .The wideband Butler Matrix was measured using Network Analyzer. Figure 11 shows the simulation and measurement results of insertion loss when a signal was fed into port 1, port 2, port 3, and port 4, respectively. The insertion loss are varies between 5dB up to 10dB. For the ideal Butler matrix, it should be better than 6dB. Imperfection of fabrication could contribute to reduction of the insertion loss.Figure 11:Insertion loss of the proposed Butler Matrix when different ports are fed.The simulated and measured results of the return loss at each port of the widedend Butler Matrix is shown in Figure 12. For a return loss better than 10dB, it has a bandwidth about17% centered at 9.4GHz.Figure 12:Return loss of the proposed Butler Matrix when different ports are fed.Figure 13 shows the phase difference of measured results when a signal was fed into port 1, port 2, port 3, and port 4, respectively. The overall phase error was less than 7°. There are several possible reasons for this phase error. A lot of bends in high frequency can produce phase error. Moreover, the imperfection of soldering, etching, alignment, and fastening also could contribute to deviation of the phase error.Figure 13:Phase difference of the proposed Butler Matrix when different ports are fed.Table 1shows that each input port was resulted a specific linear phase at the output ports. The phase differences each between the output ports are of the same value. The phase difference can generate a different beam ( θ). If port 1 (P1) is excited, the phase difference was 45°, the direction of generated beam ( θ) will be 14.4°for 1L. It is summarized in Table 1.Table 1:Output phase difference and estimated direction of generated beam.5. ConclusionA novel nonuniform branch line coupler has been employed to achieve a wideband characteristic by exponential impedance taper technique. It is a simple design without needs of using multilayer technology and this will lead to cost reduction and design simplification. The scattering matrix of the nonuniform branch line coupler was derived and it was proved that the nonuniform branch line coupler has equal magnitude of −3dB. Moreover, the novel nonuniform branch line coupler has been employed to achieve a wideband 0dB crossover. Furthermore, these components have been implemented in the Butler Matrix and that achieves wideband characteristics.ReferencesT. A. Denidni and T. E. Libar, “Wide band four-port butler matrix for switched multibeam antenna arrays,” in Proceedings of the IEEE International Symposium on Personal, Indoor and Mobile Radio Communications (PIMRC '03), vol. 3, pp. 2461–2464, 2003. View at Publisher ·View at Google Scholar2 E. Siachalou, E. Vafiadis, S. S. Goudos, T. Samaras, C. S. Koukourlis, and S. Panas, “On the design of switched-beam wideband base stations,” IEEE Antennas and Propagation Magazine, vol. 46, no. 1, pp. 158–167, 2004. View at Publisher ·View at Google Scholar ·View at Scopus3P. S. Hall and S. J. Vetterlein, “Review of radio frequency beamforming techniques for scanned and multiple beam antennas,”IEE Proceedings H, vol. 137, no. 5, pp. 293–303, 1990. View at Scopus4W-D. Wirth, Radar Techniques Using Array Anney, IEE Publishers, Stevenage, UK, 2001.5S. Y. Zheng, S. H. Yeung, W. S. Chan, and K. F. Man, “Broadband butler matrix optimized using jumping genes evolutionary algorithm,” in Proceedings of the IEEE International Conference on Industrial Technology (IEEE ICIT '08), Hong Kong, April 2008. View at Publisher ·View at Google Scholar6K. Wincza and K. Sachse, “Broadband Butler matrix in microstrip multilayer technology designed with the use of three-section directional couplers and phase correction Networks,” in Proceedings of the 18th International Conference on Microwave Radar and Wireless Communications(MIKON '10), Cracow, Poland, June 2010.7 A. M. El Tager and M. A. Eleiwa, “Design and implementation of a smart antenna using butler matrix for ISM band,”in Proceedings of the Progress in Electromagnetics Research Symposium (PIERS '09), pp. 571–575, Beijing, China, March 2009.8Y. S. Jeong and T. W. Kim, “Design and analysis of swapped port coupler and its application in a miniaturized butler matrix,”IEEE Transactions on Microwave Theory and Techniques, vol. 58, no. 4, pp. 764–770, 2010. View at Publisher ·View at Google Scholar ·View at Scopus9 C. Collado, A. Grau, and F. De Flaviis, “Dual-band butler matrix for WLAN systems,”in IEEE MTT-S International Microwave Symposium Digest, vol. 2005, pp. 2247–2250, 2009. View at Publisher ·View at Google Scholar10K. Wincza, S. Gruszczynski, and K. Sachse, “Integrated four-beam dual-band antenna array fed by broadband Butler matrix,” Electronics Letters, vol. 43, no. 1, pp. 7–8, 2007. View at Publisher ·View at Google Scholar ·View at Scopus11T. N. Kaifas and J. N. Sahalos, “On the design of a single-layer wideband Butler matrix for switched-beam UMTS system applications,” IEEE Antennas and Propagation Magazine, vol. 48, no. 6, pp. 193–204, 2006. View at Publisher ·View at Google Scholar ·View at Scopus12K. Wincza and S. Gruszczynski, “A broadband 4 × 4 butler matrix for modern-day antennas,” in Proceedings of the 35th European Microwave Conference, pp. 1331–1334, Paris, France, October 2005. View at Publisher ·View at Google Scholar13S. Gruszczyński, K. Wincza, and K. Sachse, “Reduced sidelobe four-beam N-element antenna arrays fed by 4 × 4 N butler matrices,” IEEE Antennas and Wireless PropagationLetters, vol. 5, no. 1, pp. 430–434, 2006. View at Publisher ·View at Google Scholar ·View at Scopus14Y. C. Su, M. E. Bialkowski, F. C. E. Tsai, and K. H. Cheng, “UWB switched-beam array antenna employing UWB butler matrix,”in Proceedings of the IEEE International Workshop on Antenna Technology: Small Antennas and Novel Metamaterials (iWAT '08), pp. 199–202, Hsinchu, Taiwan, March 2008. View at Publisher ·View at Google Scholar15J. He, B. Z. Wang, Q. Q. He, Y. X. Xing, and Z. L. Yin, “Wideband x-band microstrip Butler matrix,”Progress in Electromagnetics Rese arch, vol. 74, pp. 131–140, 2007. View at Scopus16Y. Liuqing and G. B. Giannakis, “Ultra wideband communications,”IEEE Signal Proces sing Magazine, vol. 21, no. 6, pp. 26–54, 2004. View at Publisher ·View at Google Scholar ·View at Scopus17S. Banba and H. Ogawa, “Multilayer MMIC directional couplers using thin dielectric layers,” IEEE Transaction s on Micro wave Theory and Techniques, vol. 43, no. 6, pp. 1270–1275, 1995. View at Publisher ·View at Google Scholar ·View at Scopus18J. Sebastien and G. Y. Delisle, “Microstrip EHF butler matrix design and realization,”ETRI Journal, vol. 27, no. 6, pp. 788–797, 2005. View at Scopus19J. H. Cho, H. Y. Hwang, and S. W. Yun, “A design of wideband 3-dB coupler with N-section microstrip tandem structure,” IEEE Microwave and Wireless Components Letters, vol. 15, no. 2, pp. 113–115, 2005. View at Publisher ·View at Google Scholar ·View at Scopus20M. E. Bialkowski, N. Seman, and M. S. Leong, “Design of a compact ultra wideband 3 db microstrip-slot coupler with high return losses and isolation,” in Asia Pacific Microwave Conference (APMC '09), pp. 1334–1337, St. Lucia, Australia, December 2009. View at Publisher ·View at Google Scholar21R. P. Hecken, “A near-optimum matching section without discontinuities,” IEEE Transactions on Microwave Theory and Techniques, vol. 20, no. 11, pp. 734–739, 1972.22 D. M. Pozar, Microwave Engineering, John Wiley& Sons, New York, NY, USA, 2nd edition, 1998.23M. Bona, L. Manholm, J. P. Starski, and B. Svensson, “Low loss compact butler matrix for a microstrip antenna,” IEEE Transactions on Microwave Theory and Techniques, vol. 50, no. 9, pp. 2069–2075, 2002. View at Publisher ·View at Google Scholar24M. Kobayashi and N. Sawada, “Analysis and synthesis of tapered microstrip transmission lines,”IEEE Transactions on Microwave Theory and Techniques, vol. 40, no. 8, pp. 1642–1646, 1992. View at Publisher ·View at Google Scholar宽带非均匀支线耦合器及其应用在宽带巴特勒矩阵的设计与分析协作院校：印尼大学新校区电机工程学系天线传播和微波研究小组（AMRG）。

Microstrip Antenna

1An Introduction to Microstrip Antennas1.1IntroductionDeschamps first proposed the concept of the MSA in1953[1].However, practical antennas were developed by Munson[2,3]and Howell[4]in the 1970s.The numerous advantages of MSA,such as its low weight,small volume,and ease of fabrication using printed-circuit technology,led to the design of several configurations for various applications[5–9].With increas-ing requirements for personal and mobile communications,the demand for smaller and low-profile antennas has brought the MSA to the forefront.An MSA in its simplest form consists of a radiating patch on one side of a dielectric substrate and a ground plane on the other side.The top and side views of a rectangular MSA(RMSA)are shown in Figure1.1.However, other shapes,such as the square,circular,triangular,semicircular,sectoral, and annular ring shapes shown in Figure1.2,are also used.Radiation from the MSA can occur from the fringing fields between the periphery of the patch and the ground plane.The length L of the rectangular patch for the fundamental TM10mode excitation is slightly smaller than␭/2,where␭is the wavelength in the dielectric medium,which in terms of free-space wavelength␭o is given as␭o/√⑀e,where⑀e is the effective dielectric constant of a microstrip line of width W.The value of ⑀e is slightly less than the dielectric constant⑀r of the substrate because thefringing fields from the patch to the ground plane are not confined in the12Broadband Microstrip AntennasFigure1.1MSA configuration.Figure1.2Different shapes of microstrip patches.dielectric only,but are also spread in the air.To enhance the fringing fields from the patch,which account for the radiation,the width W of the patch is increased.The fringing fields are also enhanced by decreasing the⑀r or by increasing the substrate thickness h.Therefore,unlike the microwave integrated circuit(MIC)applications,MSA uses microstrip patches with larger width and substrates with lower⑀r and thicker h.The details of various substrates used for MSA are given in Appendix A.For MSA applications in the microwave frequency band,generally h is taken greater than or equal to 1/16th of an inch(0.159cm).A typical comparison of MSA with MIC in the microwave frequency range is given in Table1.1.An Introduction to Microstrip Antennas3Table1.1A Comparison of MIC and MSAMIC MSAh≤0.159cm≥0.159cm⑀r≥9.8≤9.8W Small LargeRadiation Minimized Maximized1.2Characteristics of MSAsThe MSA has proved to be an excellent radiator for many applications because of its several advantages,but it also has some disadvantages.The advantages and disadvantages of the MSA are given in Sections1.2.1and 1.2.2.1.2.1AdvantagesMSAs have several advantages compared to the conventional microwave antennas.The main advantages of MSAs are listed as follows:•They are lightweight and have a small volume and a low-profileplanar configuration.•They can be made conformal to the host surface.•Their ease of mass production using printed-circuit technology leadsto a low fabrication cost.•They are easier to integrate with other MICs on the same substrate.•They allow both linear polarization and CP.•They can be made compact for use in personal mobile communica-tion.•They allow for dual-and triple-frequency operations.1.2.2DisadvantagesMSAs suffer from some disadvantages as compared to conventional micro-wave antennas.They are the following:•Narrow BW;•Lower gain;•Low power-handling capability.4Broadband Microstrip AntennasMSAs have narrow BW,typically1–5%,which is the major limiting factor for the widespread application of these antennas.Increasing the BW of MSAs has been the major thrust of research in this field,and broad BW up to70%has been achieved[9,10].Various broadband MSA configurations are summarized in this chapter,and they are detailed in the following chapters.1.2.3Applications of MSAsThe advantages of MSAs make them suitable for numerous applications [6–11].The telemetry and communications antennas on missiles need to be thin and conformal and are often MSAs.Radar altimeters use small arrays of microstrip radiators.Other aircraft-related applications include antennas for telephone and satellite communications.Microstrip arrays have been used for satellite imaging systems.Patch antennas have been used on commu-nication links between ships or buoys and satellites.Smart weapon systems use MSAs because of their thin profile.Pagers,the global system for mobile communication(GSM),and the global positioning system(GPS)are major users of MSAs.Some of the applications of MSAs are listed in Table1.2.1.3Feeding TechniquesThe MSA can be excited directly either by a coaxial probe or by a microstrip line.It can also be excited indirectly using electromagnetic coupling or aperture coupling and a coplanar waveguide feed,in which case there is noTable1.2Typical Applications of MSAsSystem ApplicationAircraft and ship antennas Communication and navigation,altimeters,blindlanding systemsMissiles Radar,proximity fuses,and telemetrySatellite communications Domestic direct broadcast TV,vehicle-basedantennas,communicationMobile radio Pagers and hand telephones,man pack systems,mobile vehicleRemote sensing Large lightweight aperturesBiomedical Applicators in microwave hyperthermiaOthers Intruder alarms,personal communication,and soforthAn Introduction to Microstrip Antennas5direct metallic contact between the feed line and the patch[9–12].Feeding technique influences the input impedance and characteristics of the antenna, and is an important design parameter.The coaxial or probe feed arrangement is shown in Figure1.1.The center conductor of the coaxial connector is soldered to the patch.The main advantage of this feed is that it can be placed at any desired location inside the patch to match with its input impedance.The disadvantages are that the hole has to be drilled in the substrate and that the connector protrudes outside the bottom ground plane,so that it is not completely planar.Also, this feeding arrangement makes the configuration asymmetrical.A patch excited by microstrip line feed is shown in Figure1.3(a).This feed arrangement has the advantage that it can be etched on the same substrate,so the total structure remains planar.The drawback is the radiation from the feed line,which leads to an increase in the cross-polar level.Also, in the millimeter-wave range,the size of the feed line is comparable to the patch size,leading to increased undesired radiation.For thick substrates,which are generally employed to achieve broad BW,both the above methods of direct feeding the MSA have problems.In the case of a coaxial feed,increased probe length makes the input impedance more inductive,leading to the matching problem.For the microstrip feed, an increase in the substrate thickness increases its width,which in turn increases the undesired feed radiation.The indirect feed,discussed below, solves these problems.An electromagnetically coupled RMSA is shown in Figure1.3(b).The electromagnetic coupling is also known as proximity coupling[9,12,13].The feed line is placed between the patch and the ground plane,which is separated by two dielectric media.The advantages of this feed configuration include the elimination of spurious feed-network radiation;the choice between two different dielectric media,one for the patch and the other for the feed line to optimize the individual performances; and an increase in the BW due to the increase in the overall substrate thickness of the MSA.The disadvantages are that the two layers need to be aligned properly and that the overall thickness of the antenna increases.Another method for indirectly exciting a patch employs aperture cou-pling[14].In the aperture-coupled MSA configuration,the field is coupled from the microstrip line feed to the radiating patch through an electrically small aperture or slot cut in the ground plane,as shown in Figure1.3(c). The coupling aperture is usually centered under the patch,leading to lower cross-polarization due to symmetry of the configuration.The shape,size, and location of the aperture decide the amount of coupling from the feed line to the patch[15–17].The slot aperture can be either resonant or6Broadband Microstrip AntennasFigure1.3Rectangular MSA fed by(a)microstrip line,(b)electromagnetic coupling,(c)aperture coupling,and(d)coplanar waveguide(CPW).An Introduction to Microstrip Antennas7nonresonant[10,11].The resonant slot provides another resonance in addition to the patch resonance thereby increasing the BW at the expense of an increase in back radiation.As a result,a nonresonant aperture is normally used.The performance is relatively insensitive to small errors in the alignment of the different layers.Similar to the electromagnetic coupling method,the substrate parameters of the two layers can be chosen separately for optimum antenna performance.This feeding method gives increased BW as described in Chapter4.The coplanar waveguide feed,shown in Figure1.3(d),has also been used to excite the MSA[18].In this method,the coplanar waveguide is etched on the ground plane of the MSA.The line is excited by a coaxial feed and is terminated by a slot,whose length is chosen to be between0.25 and0.29of the slot wavelength.The main disadvantage of this method is the high radiation from the rather longer slot,leading to the poor front-to-back ratio.The front-to-back ratio is improved by reducing the slot dimension and modifying its shape in the form of a loop[19].1.4Methods of AnalysisThe MSA generally has a two-dimensional radiating patch on a thin dielectric substrate and therefore may be categorized as a two-dimensional planar component for analysis purposes.The analysis methods for MSAs can be broadly divided into two groups.In the first group,the methods are based on equivalent magnetic current distribution around the patch edges(similar to slot antennas).There are three popular analytical techniques:•The transmission line model;•The cavity model;•The MNM.In the second group,the methods are based on the electric current distribution on the patch conductor and the ground plane(similar to dipole antennas,used in conjunction with full-wave simulation/numerical analysis methods).Some of the numerical methods for analyzing MSAs are listed as follows:8Broadband Microstrip Antennas•The method of moments(MoM);•The finite-element method(FEM);•The spectral domain technique(SDT);•The finite-difference time domain(FDTD)method.This section briefly describes these methods.1.4.1Transmission Line ModelThe transmission line model is very simple and helpful in understanding the basic performance of a MSA.The microstrip radiator element is viewed as a transmission line resonator with no transverse field variations(the field only varies along the length),and the radiation occurs mainly from the fringing fields at the open circuited ends.The patch is represented by two slots that are spaced by the length of the resonator.This model was originally developed for rectangular patches but has been extended for generalized patch shapes.Many variations of this method have been used to analyze the MSA[9,20–22].Although the transmission line model is easy to use,all types of configu-rations can not be analyzed using this model since it does not take care of variation of field in the orthogonal direction to the direction of propagation.1.4.2Cavity ModelIn the cavity model,the region between the patch and the ground plane is treated as a cavity that is surrounded by magnetic walls around the periphery and by electric walls from the top and bottom sides.Since thin substrates are used,the field inside the cavity is uniform along the thickness of the substrate[23–25].The fields underneath the patch for regular shapes such as rectangular,circular,triangular,and sectoral shapes can be expressed as a summation of the various resonant modes of the two-dimensional resonator.The fringing fields around the periphery are taken care of by extending the patch boundary outward so that the effective dimensions are larger than the physical dimensions of the patch.The effect of the radiation from the antenna and the conductor loss are accounted for by adding these losses to the loss tangent of the dielectric substrate.The far field and radiated power are computed from the equivalent magnetic current around the periphery.An alternate way of incorporating the radiation effect in the cavity model is by introducing an impedance boundary condition at the walls of the cavity.The fringing fields and the radiated power are not included insideAn Introduction to Microstrip Antennas9the cavity but are localized at the edges of the cavity.However,the solution for the far field,with admittance walls is difficult to evaluate[9].1.4.3MNMThe MNM for analyzing the MSA is an extension of the cavity model[9, 26,27].In this method,the electromagnetic fields underneath the patch and outside the patch are modeled separately.The patch is analyzed as a two-dimensional planar network,with a multiple number of ports located around the periphery.The multiport impedance matrix of the patch is obtained from its two-dimensional Green’s function.The fringing fields along the periphery and the radiated fields are incorporated by adding an equivalent edge admittance network.The segmentation method is then used to find the overall impedance matrix.The radiated fields are obtained from the voltage distribution around the periphery.Appendix C details this method.The above three analytical methods offer both simplicity and physical insight.In the latter two methods,the radiation from the MSA is calculated from the equivalent magnetic current distribution around the periphery of the radiating patch,which is obtained from the corresponding voltage distribution.Thus,the MSA analysis problem reduces to that of finding the edge voltage distribution for a given excitation and for a specified mode. These methods are accurate for regular patch geometries,but—except for MNM with contour integration techniques—they are not suited for arbitrary shaped patch configurations.For complex geometries,the numerical tech-niques described below are employed[9].1.4.4MoMIn the MoM,the surface currents are used to model the microstrip patch, and volume polarization currents in the dielectric slab are used to model the fields in the dielectric slab.An integral equation is formulated for the unknown currents on the microstrip patches and the feed lines and their images in the ground plane[28].The integral equations are transformed into algebraic equations that can be easily solved using a computer.This method takes into account the fringing fields outside the physical boundary of the two-dimensional patch,thus providing a more exact solution.This book makes extensive use of a commercially available software IE3D[29] based on MoM to analyze various MSA configurations.10Broadband Microstrip Antennas1.4.5FEMThe FEM,unlike the MoM,is suitable for volumetric configurations.In this method,the region of interest is divided into any number of finite surfaces or volume elements depending upon the planar or volumetric structures to be analyzed[30].These discretized units,generally referred to as finite elements,can be any well-defined geometrical shapes such as triangular elements for planar configurations and tetrahedral and prismatic elements for three-dimensional configurations,which are suitable even for curved geometry.It involves the integration of certain basis functions over the entire conducting patch,which is divided into a number of subsections.The problem of solving wave equations with inhomogeneous boundary conditions is tackled by decomposing it into two boundary value problems,one with Laplace’s equation with an inhomogeneous boundary and the other corre-sponding to an inhomogeneous wave equation with a homogeneous boundary condition[10].1.4.6SDTIn the SDT,a two-dimensional Fourier transform along the two orthogonal directions of the patch in the plane of substrate is employed.Boundary conditions are applied in Fourier transform plane.The current distribution on the conducting patch is expanded in terms of chosen basis functions, and the resulting matrix equation is solved to evaluate the electric current distribution on the conducting patch and the equivalent magnetic current distribution on the surrounding substrate surface.The various parameters of the antennas are then evaluated[31].1.4.7FDTD MethodThe FDTD method is well-suited for MSAs,as it can conveniently model numerous structural inhomogenities encountered in these configurations[10].It can also predict the response of the MSA over the wide BW witha single simulation.In this technique,spatial as well as time grid for the electric and magnetic fields are generated over which the solution is required. The spatial discretizations along three Cartesian coordinates are taken to be same.The E cell edges are aligned with the boundary of the configuration and H-fields are assumed to be located at the center of each E cell.Each cell contains information about material characteristics.The cells containing the sources are excited with a suitable excitation function,which propagates along the structure.The discretized time variations of the fields are deter-mined at desired ing a line integral of the electric field,the voltage across the two locations can be obtained.The current is computed by a loop integral of the magnetic field surrounding the conductor,where the Fourier transform yields a frequency response.The above numerical techniques,which are based on the electric current distribution on the patch conductor and the ground plane,give results for any arbitrarily shaped antenna with good accuracy,but they are time-consuming.These methods can be used to plot current distributions on patches but otherwise provide little of the physical insight required for antenna design.1.5Review of Various Broadband Techniques for MSAsAs mentioned earlier,the most serious limitation of the MSA is its narrow BW.The BW could be defined in terms of its VSWR or input impedance variation with frequency or in terms of radiation parameters.For the circularly polarized antenna,BW is defined in terms of the axial ratio(AR)[9]. Therefore,before describing the various methods for increasing the BW,the various definitions of the BW are described.1.5.1Definition of BWThe VSWR or impedance BW of the MSA is defined as the frequency range over which it is matched with that of the feed line within specified limits. The BW of the MSA is inversely proportional to its quality factor Q and is given by[32]BW=VSWR−1Q√VSWR(1.1)where VSWR is defined in terms of the input reflection coefficient⌫as:VSWR=1+|⌫|1−|⌫|(1.2)The⌫is a measure of reflected signal at the feed-point of the antenna. It is defined in terms of input impedance Z in of the antenna and the characteristic impedance Z0of the feed line as given below:⌫=Z in −Z 0Z in +Z 0(1.3)The BW is usually specified as frequency range over which VSWR is less than 2(which corresponds to a return loss of 9.5dB or 11%reflected power).Sometimes for stringent applications,the VSWR requirement is specified to be less than 1.5(which corresponds to a return loss of 14dB or 4%reflected power).Conversion of BW from one VSWR level to another can be accomplished byBW 1BW 2=(VSWR 1−1)√VSWR 1√VSWR 2(VSWR 2−1)(1.4)where BW 1and BW 2correspond to VSWR 1and VSWR 2,respectively.The variation of percentage BW for VSWR ≤2and efficiency ␩of a square MSA with normalized substrate thickness h /␭0for two different values of ⑀r (2.2and 10)are given in Figure 1.4(a).Also,the variation of percentage BW with frequency for three commonly used values of h and ⑀r =2.32is given in Figure 1.4(b).The BW of a single-patch antenna increases with an increase in the substrate thickness and a decrease in the ⑀r of the substrate [9,33,34].The BW is approximately 15%for ⑀r =2.2and h =0.1␭0.The ⑀r can be chosen close to 1to obtain a broader BW.The larger thickness of the substrate givesrise to an increase in probe reactance Figure 1.4(a)Variation of percentage BW and efficiency of a square MSA versus h /␭0.(——)⑀r =2.2,(---)⑀r =10and (b)variation of percentage BW with frequency for three values of h and ⑀r =2.32:(——)0.318,(---)0.159,(–-–)0.079cm.for the coaxial feed and the excitation of surface waves,which reduces the efficiency␩of the antenna as can be seen from Figure1.4(a).The efficiency of the MSA is defined in Appendix C.The BW enhancement of the single regularly shaped MSA is discussed in Chapter2.The expressions for approximately calculating the percentage BW of the RMSA in terms of patch dimensions and substrate parameters is given by%BW=Ah␭0√⑀r√W L(1.5)whereA=180forh␭0√⑀r≤0.045A=200for0.045≤h␭0√⑀r≤0.075A=220forh␭0√⑀r≥0.075where W and L are the width and length of the RMSA.With an increase in W,BW increases.However,W should be taken less than␭to avoid excitation of higher order modes.For other regularly shaped patches,values of equivalent W can be obtained by equating the area with that of the RMSA as described in Chapter2[35,36].Another simplified relation for quick calculation of BW(in megahertz) for VSWR=2of the MSA operating at frequency f in gigahertz,with h expressed in centimeters,is given by[37]BW≅50hf2(1.6) The BW can also be defined in terms of the antenna’s radiation parame-ters.It is defined as the frequency range over which radiation parameters such as the gain,half-power beamwidth(HPBW),and sidelobe levels are within the specified minimum and maximum limits.This definition is more complete as it also takes care of the input impedance mismatch,which also contributes to change in the gain.The expression for approximately calculating the directivity D of the RMSA is given byD≅0.2W+6.6+10log΀1.6/√⑀r΁dB(1.7) For other geometries,the values of equivalent W can be obtained by equating its area with that of the RMSA as described in Chapter2[35,36].The above definitions for BW are mainly for a linearly polarized MSA. For a circularly polarized MSA,the BW is generally limited by its AR.This BW is the frequency range over which AR is less than a maximum limit (e.g.,3or6dB).There are various techniques for increasing the BW of the MSAs.The main techniques used to increase the BW are presented briefly in the following sections.1.5.2Modified Shape PatchesThe regular MSA configurations,such as rectangular and circular patches have been modified to rectangular ring[38]and circular ring[39],respectively,to enhance the BW.The larger BW is because of a reduction in the quality factor Q of the patch resonator,which is due to less energy stored beneath the patch and higher radiation.When a U-shaped slot is cut inside the rectangular patch,it gives a BW of approximately40%for VSWR≤2[40]. Similar results are obtained when a U-slot is cut inside a circular or a triangular MSA[41,42].These configurations are discussed in detail in Chapter6.1.5.3Planar Multiresonator ConfigurationsThe planar stagger–tuned coupled multiple resonators yield wide BW in the same way as in the case of multistage tuned circuits.Several configurations are available yielding BW of5–25%[43–49].Various parasitic patches like narrow strips,shorted quarter-wavelength rectangular patches,and rectangu-lar resonator patches have been gap-coupled to the central-fed rectangular patch.Three combinations of gap-coupled rectangular patches are shown in Figure1.5.To reduce the criticality of the gap coupling,direct coupling as depicted in Figure1.6has been used to obtain broad BW.Both gap and direct(hybrid)coupling have been used with circular MSAs(CMSAs)and equilateral triangular MSAs(ETMSAs)to yield broad BW.Figure1.5Various gap-coupled multiresonator RMSA configurations:(a)three RMSAs gap-coupled along radiating edges,(b)three RMSAs gap-coupled along non-radiating edges,and(c)five gap-coupled RMSAs.These planar multiresonator configurations yield broad BW but have the following disadvantages:•The large size,which makes them unsuitable as an array element;•The variation in the radiation pattern over the impedance BW.A modification of the multiresonator patches—to avoid the above-mentioned problems—entails using five or six narrow strips that are gap-coupled along the width[49].This yielded wide BW with a relatively small variation in pattern over the BW.Various broadband planar multiresonator configurations are covered in Chapter3.1.5.4Multilayer ConfigurationsIn the multilayer configuration,two or more patches on different layers of the dielectric substrate are stacked on each other.Based on the coupling mechanism,these configurations are categorized as electromagnetically cou-pled or aperture-coupled MSAs.Figure1.6Various direct-coupled multiresonators:(a)three RMSAs direct-coupled along radiating edges,(b)three RMSAs direct-coupled along nonradiating edges,and(c)five direct-coupled RMSAs.1.5.4.1Electromagnetically Coupled MSAsIn the electromagnetically coupled MSA,one or more patches at the different dielectric layers are electromagnetically coupled to the feed line located at the bottom dielectric layer as shown in Figure1.3(b).Alternatively,one of the patches is fed by a coaxial probe and the other patch is electromagnetically coupled.Either the bottom or top patch is fed with a coaxial probe as shown in Figure1.7.The patches can be fabricated on different substrates,and accordingly the patch dimensions are to be optimized so that the resonance frequencies of the patches are close to each other to yield broad BW.TheseFigure1.7An electromagnetically coupled MSA,in which(a)the bottom patch is fed and(b)the top patch is fed.two layers may be separated by either air-gap or foam yielding BW of15–30% [50–56].1.5.4.2Aperture-Coupled MSAsIn the aperture-coupled MSA,the field is coupled from the microstrip feed line placed on the other side of the ground plane to the radiating patch through an electrically small aperture/slot in the ground plane,as shown in Figure1.3(c).Two different dielectric substrates could be chosen,one for the patch and the other for the feed line to optimize the individual perfor-mances.The coupling to the patch from the feed line can be maximized by choosing the optimum shape of the aperture[14–16].Two patches of rectangular or circular shapes,which are stacked on each other in different dielectric layers yield around30%BW[57–60].A BW of nearly70%has been obtained by stacking patches with resonant apertures[61].The multilayer broadband MSAs,unlike single-layer multiresonator configurations,show a very small degradation in radiation pattern over the complete VSWR BW.The drawback of these structures is the increased height,which is not desirable for conformal applications and increased back radiation for aperture-coupled MSAs.Multilayered configurations using both electromagnetic as well as aperture coupling are described in Chapter4.1.5.5Stacked Multiresonator MSAsThe planar and stacked multiresonator techniques are combined to further increase the BW and gain.A probe-fed single rectangular or circular patch located on the bottom layer has been used to excite multiple rectangular or circular patches on the top layer,respectively[62,63].Besides increasing the BW,these configurations also provide an increase in gain as described in Chapter5.1.5.6Impedance-Matching Networks for Broadband MSAsThe impedance-matching networks are used to increase the BW of the MSA. Some examples that provide about10%BW are the rectangular MSA with a coplanar microstrip impedance-matching network and an electromagnetically coupled MSA with single-stub matching as shown in Figure1.8[12,64–66].1.5.7Log-Periodic MSA ConfigurationsThe concept of log-periodic antenna has been applied to MSA to obtain a multi-octave BW.In this configuration,the patch dimensions are increased。

Glossary of Satellite Technology 卫星技术词汇总结

非辐射边馈电的宽带双层微带贴片天线

非辐射边馈电的宽带双层微带贴片天线卢晓鹏;张玉梅;李昂【期刊名称】《雷达科学与技术》【年(卷),期】2011(9)5【摘要】This paper proposes a broadband double-layer rectangular microstrip antenna(DL-RMSA) fed from the non-radiating edge of a bottom patch. The characteristics of the antenna has been analyzed via simulation tool. This design shows that cutting 3 to 5 slots along the co-polarization into the parasitic patch can effectively suppress the cross polarization radiation as well as enhances the impedance bandwidth. As compared to the conventional DL-RMSAs, the proposed antenna presents a 30% broader impedance bandwidth) while maintaining the cross polarization level. Using this antenna as an element in the construction ofa corresponding array can simply the feed network and yield a co-planar-fed microstrip array with broad operation bandwidth, high radiation efficiency and wide-angle scanning capability. A X-band demo array of 8 X 8 elements has been fabricated and tested. The measurement results show that this array antenna exhibits low cross-polarization level and high radiation efficiency within a relative bandwidth of 17. 6%.%介绍了一种非辐射边馈电的宽带双层微带贴片天线单元,并对其参数特性进行了仿真研究,结果表明,通过在寄生贴片上开3～5个与极化方向相平行的缝,可有效抑制天线的交叉极化,同时改善天线的阻抗带宽.相比传统双层微带贴片,该天线单元的阻抗带宽可提高3％以上,而交叉极化指标相当.当该单元应用于阵列天线设计时,可简化馈电网络,便于实现宽带、高效、大扫描角的微带共面馈电天线阵.对X波段8×8单元实验小阵的测试结果表明,该天线在17.6％的频段内具有良好的交叉极化性能及较高的工作效率.【总页数】5页(P479-483)【作者】卢晓鹏;张玉梅;李昂【作者单位】中国电子科技集团公司第三十八研究所,安徽合肥230088;中国电子科技集团公司第三十八研究所,安徽合肥230088;中国电子科技集团公司第三十八研究所,安徽合肥230088【正文语种】中文【中图分类】TN821+.4;TN957.2【相关文献】1.双层宽带微带贴片天线 [J], 李倩;吕晓德2.基于HFSS的双层宽带微带贴片天线的研究 [J], 闵刚;武永刚3.用于宽带高功率微波辐射的双层贴片天线特性 [J], 徐刚;廖勇;孟凡宝;唐传祥;杨周炳;谢平4.基于HFSS的双层宽带微带贴片天线的研究 [J], 武永刚;邢光龙;楚玉焕5.一种面向导航应用的电容耦合四点馈电的宽带双层微带天线 [J], 王超;艾永强;张振杰;郑伟;杨文丽因版权原因，仅展示原文概要，查看原文内容请购买。

正交缝隙耦合馈电宽带圆极化微带天线设计

正交缝隙耦合馈电宽带圆极化微带天线设计张昭;曹祥玉;李思佳;郭蓉【摘要】为了实现圆极化微带天线的频带拓宽和增益提高,在缝隙耦合天线的基础上,设计了一种Ku频段正交缝隙耦合馈电的宽带圆极化微带天线.该天线以双层方形贴片为辐射单元,在拓展天线阻抗带宽的同时提高了增益;采用微带线结合正交左旋缝隙结构实现耦合馈电,通过优化缝隙结构改善了天线轴比特性.测量结果表明:阻抗带宽(VSWR＜2)和轴比带宽(AR＜3 dB)分别达到22.5％和16.2％,轴比带宽内天线增益均大于9 dBi.该结构天线以其简单的馈电设计为宽带圆极化微带天线设计提供了一定的参考价值.【期刊名称】《空军工程大学学报（自然科学版）》【年(卷),期】2014(015)002【总页数】5页(P57-61)【关键词】微带天线;宽频带;圆极化;正交左旋缝隙【作者】张昭;曹祥玉;李思佳;郭蓉【作者单位】空军工程大学信息与导航学院,陕西西安,710077;空军工程大学信息与导航学院,陕西西安,710077;空军工程大学信息与导航学院,陕西西安,710077;空军工程大学信息与导航学院,陕西西安,710077【正文语种】中文【中图分类】TN82现阶段圆极化微带天线具有体积小、剖面低、易共形、能接收任意线极化来波等优点在卫星通信领域中拥有广阔的应用前景,因此对其研究具有重要意义[1-2]。

但是由于圆极化微带天线阻抗带宽和轴比带宽通常较窄、增益较低需要进一步的研究。

针对拓展圆极化轴比带宽、提高天线增益的问题,文献[3]提出一种双层贴片结构,利用威尔金森功分器通过H形口径耦合馈电,轴比带宽虽达到30%,但增益仅大于6 dBi;文献[4～5]深入介绍了通过微带巴伦为4个L型探针馈电的单贴片圆极化微带天线,3 dB轴比带宽均达到了80%,但带内增益仅在3 dBi以上;文献[6～7]应用三馈电方式实现圆极化辐射,3 dB轴比带宽分别达到19.8%和33%,带内增益大于3 dBi;以上设计虽然轴比带宽得到很大扩展,但是馈电网络的设计都很复杂,且带内增益较低。

Microstrip antenna

专利名称：Microstrip antenna发明人：Iwasaki, Hisao, c/o Int. Prop. Div., K.K.Toshiba,Sawada, Hisashi, c/o Int. Prop. Div.,K.K. Toshiba申请号：EP92304879.7申请日：19920528公开号：EP0516440A1公开日：19921202专利内容由知识产权出版社提供专利附图：摘要：A microstrip antenna is disclosed which comprises a ground conductor plate (5)and a patch (2) opposed to the ground conductor plate (5) with a particular distance (h), atransmission feed line and a reception feed line (3,4) being disposed between the ground conductor plate (5) and the patch (2). Signals are fed from these feed lines (3,4) to the patch (2) by electromagnetic coupling. The angle made by the extended lines of these feed lines (3,4) is nearly 90°. When four patches are disposed in a square arrangement, the transmission feed line feeds signals in directions of first lines which pass through the center point of each patch in such a way that the directions are line-symmetrical with respect to a horizontal line and a vertical line which pass through the center point of the square arrangement. On the other hand, the reception feed line feeds signals in the directions of second lines which pass through the center point of each patch and intersect with each first line at right angle. As a result, the mutual coupling between transmission and reception can be suppressed to a low level. In addition, when the transmission feed line is radiately connected from the center point of the square arrangement to each patch, the length thereof can be reduced, thereby decreasing the transmission loss.申请人：KABUSHIKI KAISHA TOSHIBA地址：72, Horikawa-cho, Saiwai-ku Kawasaki-shi, Kanagawa-ken 210 JP国籍：JP代理机构：Freed, Arthur Woolf更多信息请下载全文后查看。

MICROSTRIP ANTENNA

专利名称：MICROSTRIP ANTENNA发明人：TAGA TOKIO,MISHIMA AKIRA 申请号：JP8081179申请日：19790628公开号：JPS566503A公开日：19810123专利内容由知识产权出版社提供摘要：PURPOSE:To realize the low posture, a high gain and a wide band each by providing the microslot or the slender microcapacity bar extending to the radius direction on the fringe of the circular radiation element. CONSTITUTION:The microslot (dielectric load impedance) or slender microcapacity bar 7 (capacitive load impedance) extending to the radius direction is provided on the fringe of circular radiation element 1. As a result, the antenna is resonated by the two different frequencies which are separated into the upper and lower parts of the basic resonance frequency decided by the radius of the disk. Thus the 2-point resonance frequencies can be resonated to the transmission and reception bands each in the case of the communication system in which both the transmission band and the reception band are required separately by selecting properly both the measurements and the loading position of load impedance 7 or 8.申请人：NIPPON TELEGRAPH & TELEPHONE更多信息请下载全文后查看。

MICROSTRIP ANTENNA

专利名称：MICROSTRIP ANTENNA 发明人：MATSUNAGA MAKOTO 申请号：JP12759091申请日：19910530公开号：JPH04352503A公开日：19921207专利内容由知识产权出版社提供摘要：PURPOSE:To miniaturize a coupling part between a radiator and a feeder line and to obtain the antenna with high performance by forming the opening part electromagnetically coupling the radiator and the feeder line so as to present induced reactance when being observed from the feeder line. CONSTITUTION:The induced reactance due to a fine gap 5 provided at a ground conductor 4 is loaded to a microstrip line 8 at the position of the fine gap 5. Since capacitic reactance presented by a chip-shaped capacitor 16 grounding one electrode is serially connected to a strip conductor 6 of the line 8 after crossing the fine gap 5, the reactance of the impedance of the line 8 is canceled and serially resonated and a short-circuited state is obtained at the position of the fine gap 5 by suitably selecting the capacity of the capacitor 16. Therefore, the magnetic field of radio waves propagated in the line 8 is made maximum and excites the fine gap 5, the coupling with an antenna 7 is made strong, and the antenna is efficiently electromagnetically coupled with no contact so as to feed power from the line 8.申请人：MITSUBISHI ELECTRIC CORP更多信息请下载全文后查看。

MICROSTRIP ANTENNA

申请人：NIPPON TELEGRAPH & TELEPHONE
更多信息请载全文后查看
专利内容由知识产权出版社提供
专利名称：MICROSTRIP ANTENNA 发明人：TAGA TOKIO 申请号：J P 4 4 05680 申请日：198004 05 公开号：J P S5614 1604 A 公开日：19811105
摘要：PURPOSE:To realize two input/output terminals which resonate in frequencies being mutually different, by providing feeding points on both the long axis and the short axis of an elliptical radiation conductor element, in the microstrip antenna which is used for the mobile communication system, etc. CONSTITUTION:In the structure which consists of a radiation conductor element 1, a ground conductor 2 and a dielectric 3, and is coupled with coaxial lines 8, 12 on the reverse side of the conductors 2, an elliptical radiation conductor plate is used for the element 1. Feeding points 6, 10 are provided on both the long axis 4 and the short axis 5 of this element, and two energizing modes falling at right angles with each other are energized independently. As a result, it is possible to obtain two input/output terminals 9, 13 which are operated in different frequencies. Also, the energizing except a necessary mode is suppressed by shortcircuiting the element 1 and the conductor 2 in the intersection point of the long axis 4 and the short axis 5 of the element 1. In this way, it is possible to obtain two input/output terminals which are small-sized, light in weight and low in attitude, and resonate in frequencies being mutually different.

Slot coupling patch antenna

专利名称：Slot coupling patch antenna发明人：Qian Li,Wladimiro Villarroel申请号：US11025499申请日：20041229公开号：US07126549B2公开日：20061024专利内容由知识产权出版社提供专利附图：摘要：An antenna for receiving and/or transmitting circularly and/or linearly polarized RF signals includes a radiation element, a ground plane, a dielectric substrate, and a feed line. The radiation element is disposed on a pane of glass. The radiation element defines a slot having a first leg and a second leg forming the shape of a cross for generating thecircular and/or linear polarization. The cross-shaped slot includes a center point. The ground plane is disposed substantially parallel to and spaced from the radiation element. The dielectric substrate is sandwiched between the radiation element and the ground plane. The feed line extends within the dielectric substrate and is electromagnetically coupled with the radiation element and the ground plane. The feed line terminates at a distal end short of the center point of the slot. That is, the feed line does not cross the center point. The antenna is compact in size and generally conformal to the pane of glass.申请人：Qian Li,Wladimiro Villarroel地址：Ann Arbor MI US,Worthington OH US国籍：US,US代理机构：Howard & Howard Attorneys, P.C.更多信息请下载全文后查看。

CST微波工作室(CST Microwave Studio)中网格划分(Mesh)和波导端口(Waveguide Port)

摘要—本文简述了在CST微波工作室（CST Microwave Studio）中网格划分（Mesh）和波导端口（Waveguide Port）设立时的基本原理。

针对常见的微波器件结构，简单论述了CST建议的网格划分方法。

简要分析波导端口设置时需要注意的网格长度设置原则。

索引词—CST、仿真I. 简述CST工作室套件（CST Studio Suite）因为多种类的组件和良好的技术支持已经成为很多微波从业者首选的电磁仿真工具。

在使用CST微波工作室（CST Microwave Studio）的过程中，一些使用者在网格划分（Mesh）时会觉得相对比较难于理解和掌握相关的原理和技巧。

也有不少使用者在创建波导端口（Waveguide Port）时会遇到一些错误提示信息。

本文集于CST微波工作室的帮助文件作提供的官方信息，结合作者的使用经验，就上面两个问题作简单的论述与分析。

本部分设定了隐藏，您已回复过了，以下是隐藏的内容II. 网格划分A. 网格划分基本原则CST工作室套件使用有限积分法（Finite Integration Technique）求解麦柯斯韦方程（Maxwell’s Grid Equations）来进行仿真运算[1]，因此，每一个创建的模型都要“翻译”成软件可识别的离散结构，这个“翻译”过程通过网格划分（Mesh Generati on）来实现。

在实际操作中，当一个模型创建好的时候，CST专家系统（Expert System）已经按照默认设置作了初始网格划分，这个初始设置可以打开全局网格设置（Global Mesh Pro perties）来查看。

在不应用任何模板（Template）的前提下，每波长网格线数（Line s per wavelength）、最小网格限制（Lower mesh limit）和网格线比率限制（Mesh l ine ratio limit）这三个参数都为10。

关于这三个参数的意义，鉴于篇幅的原因，这里不再详细阐述，有兴趣的读者请参考CST帮助文件[2]。

微带分支线定向耦合器的小型化

微带分支线定向耦合器的小型化陈佳;张绍洲【摘要】Directional coupler in microwave technology is widely used but the size limits its scope of application. In the article change a quarter wavelength microstrip line into a T-structure by connecting Open-end stub and then the stub is equivalent to a cross-shaped microstrip line. So we get the miniaturized microstrip branch-line coupler. Simulating the microstrip branch-line coupler by software ADS, performance indicators meet the requirements. The size of miniaturized microstrip branch-line coupler is 36% of that traditional microstrip branch-line coupler.%基于尺寸原因定向耦合器在微波电路中的应用受到限制，采用接终端开路短截线的方法使四分之一波长的微带线变成T型结构，并进一步对短截线等效成为十字型微带线的方法，设计出小型化的定向耦合器。

通过使用ADS软件进行仿真。

设计的定向耦合器各项性能指标符合要求，面积为常规微带分支线定向耦合器的36％。

【期刊名称】《电子设计工程》【年(卷),期】2011(019)024【总页数】3页(P108-110)【关键词】微带线;T型结构;S参数;小型化【作者】陈佳;张绍洲【作者单位】新疆大学电气工程学院,新疆乌鲁木齐830047;宁波大红鹰学院机电学院,浙江宁波315175【正文语种】中文【中图分类】TN622定向耦合器在微波技术中有着广泛的应用，如用来检测功率、频率和频谱；把功率进行分配和合成；构成雷达天线的收发开关；是平衡混频器和测量电桥中重要组成部分；还可以利用定向耦合器来测量反射系数和功率等。