大功率IGBT直流特性和动态特性的PSPICE仿真

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PSpice直流仿真(一)

PSpice直流仿真实践（1）使用PSpice软件最终目的就是对各种电路进行仿真分析。

本章列举了各种模拟电路PSpice仿真实践的例子，读者通过这些例子，可以进一步体会PSpice 的应用特点和强大的电路分析能力。

PSpice可以对以下类型的电路进行仿真分析：直流电路、交流电路、电路的暂态、模拟电子电路、模拟电路、数模混合电路。

一、直流工作点分析语句此语句规定计算并打印出电路的直流工作点(又称直流偏置点)。

这时电路中的电感按短路、电容按开路处理。

设置了该语句，输出文件可打印所有节点电压、所有电压源的电流及电路的直流功耗、所有晶体管各极的电流和电压、非线性受控源的小信号(线性化)参数。

注意：无论输入文件中有无.OP语句，程序在进行直流、交流和暂态分析时，都要自动进行直流偏置点分析。

只是没有.OP语句时，输出文件只打印所有节点电压和所有电压源的电流及电路的直流功耗三项内容。

二、直流扫描分析语句直流分析语句用于对电路作直流分析。

语句在执行过程中，对指定的变量在指定的范围内进行扫描，每给一个变量的扫描点，就对电路进行一次直流分析计算，计算内容是节点电压和支路电流。

直流分析语句可对如下变量进行扫描：●电源：任何独立电压源和独立电流源的电流、电压值均可设为扫描变量。

●模型参数：在.MODEL语句中描述的模型参数均可设为扫描变量。

●温度：设置TEMP作为扫描变量时，对每个扫描变量值，电路元器件的模型参数都要更新为当时温度下的值，所以执行该分析程序就是分析了扫描温度下的电路的直流特性。

●全程参数：扫描变量使用关键字PARAM，后跟参数名。

按照.PARAM的定义，该扫描变量就为全程参数。

说明：对哪个变量扫描，该变量就是自变量，即Probe输出图形的横坐标。

直流分析语句格式：分析语句对变量扫描时有四种扫描方式，它们是：LIN：线性扫描，每一个扫描点和它前后扫描点之间的距离是相等的。

每两个相邻扫描点间的距离为扫描增量。

PSPICE仿真流程

PSPICE仿真流程(2013-03—18 23：32：19)采用HSPICE 软件可以在直流到高于100MHz 的微波频率范围内对电路作精确的仿真、分析和优化。

在实际应用中，HSPICE能提供关键性的电路模拟和设计方案，并且应用HSPICE进行电路模拟时，其电路规模仅取决于用户计算机的实际存储器容量。

二、新建设计工程在对应的界面下打开新建工程：2）在出现的页面中要注意对应的选择3）在进行对应的选择后进入仿真电路的设计：将生成的对应的库放置在CADENCE常用的目录中,在仿真电路的工程中放置对应的库文件。

这个地方要注意放置的.olb库应该是PSPICE文件夹下面对应的文件,在该文件的上层中library 中的.olb中的文件是不能进行仿真的，因为这些元件只有.olb,而无网表。

lib。

4）放置对应的元件：对于项目设计中用到的有源器件,需要按照上面的操作方式放置对应的器件,对于电容，电阻电感等分离器件，可以在libraries中选中所有的库，然后在滤波器中键入对应的元件就可以选中对应的器件，点击后进行放置.对分离元件的修改直接在对应的元件上面进行修改:电阻的单位分别为:k m;电容的单位分别为：P n u ；电感的单位分别为：n 及上面的单位只写量级不写单位.5）放置对应的激励源:在LIBRARIES中选中所有的库，然后键入S就可以选中以S开头的库。

然后在对应的库中选中需要的激励源.激励源有两种一种是自己进行编辑、手工绘制的这个对应在库中选择：另外一种是不需要自己进行编辑：该参数的修改可以直接的在需要修改的数值上面就行修改，也可以选定电源然后点击右键后进行对应的修改。

6）放置地符号：地符号就是在对应的source里面选择0的对应的标号.7）直流电源的放置：电源的选择里面应该注意到选择source 然后再选定VDC或者是其它的对应的参考。

8）放置探头：点击对应的探头放置在感兴趣的位置处.6 对仿真进行配置：1）对放置的项目的名称进行设置,也就是设置仿真的名称.2)对仿真进行配置：对仿真的配置主要是对两个对应的选项进行操作,Analysis中的对应操作：这个里面主要对应analysis type 以及的操作，对应扫描频率，需要注意MEG的频率单位.在configuration Files里面要注意category 中应该选择library，在filename 中选择对应的IC的库文件,选定后再选择add as global 按键，然后点击确认就可以了。

基于PSPICE仿真的IGBT功耗计算

伺服技术・SERV O TECHN IQUE基于PSP I CE 仿真的IGBT 功耗计算收稿日期:2003-05-28曹永娟　李强　林明耀(东南大学电气工程系,南京　210096)摘　要:基于PSP I CE 仿真,结合无刷直流电机提出了IGBT 功率损耗的估算方法,研究了IGBT 功率损耗与开关频率和栅极电阻阻值之间的关系。

关键词:功耗;IGBT ;仿真;中图分类号:T P 211.5 文献标识码:A 文章编号:1001-6848(2004)06-0040-02M ethod of esti m a ti ng Power loss of IGBT ba sed on PSP I CE Si m ula tionCAO Yong -juan ,L I Q iang ,L I N M ing -yao(Southeast U niuersity ,N anjing 210096,Ch ina )Abstract :In th is paper .a m ethod of esti m ating pow er lo ss of IGBT based on PSP I CE si m ulati on is p ropo sed ,and the relati ons betw een pow er lo ss of IGBT and s w itch frequency ,gate resistance are p resented .Key words :pow er lo ss ;IGBT ;si m ulati on1　IGB T 电路仿真模型图1所示是三相星形连接的无刷直流电机的全控电路。

T 1,T 2…T 6为6只IGB T 绝缘栅晶体管,S 1,S 2…S 6为6个控制信号。

本电路采用两两导通六状态运行方式,同时PWM 调制采用上桥臂调制下桥臂恒通的调制方式,这样可以降低逆变器的开关损耗。

大功率IGBT的PSPICE仿真模型

大功率IGBT的PSPICE仿真模型
康劲松;陶生桂
【期刊名称】《电气传动自动化》
【年(卷),期】2002(024)004
【摘要】应用等效模拟的方法建立了大功率IGBT的PSPISE仿真用直流模型和动态模型，并对其进行了直流特性和动态特性的研究。

对照仿真值与器件的实际值相比，所有特性均较吻合，表明此PSPICE仿真模型适用于在工种应用中分析研究大功率IGBT的直流特性和动态特性。

【总页数】4页(P6-8,11)
【作者】康劲松;陶生桂
【作者单位】同济大学电气工程系上海 200331;同济大学电气工程系上海200331
【正文语种】中文
【中图分类】TM921.51;TP391.9
【相关文献】
1.大功率IGBT和IGCT在大功率逆变器中的应用比较 [J], 许鸿文
2.适用于电动汽车的SiC MOSFET PSpice仿真模型研究 [J], 赵波;周哲;徐艳明;李虹;郑琼林
3.大功率IGBT直流特性和动态特性的PSPICE仿真 [J], 康劲松;陶生桂;王新祺
4.基于电子电路仿真软件(PSPICE)的MOV全电流仿真模型的建立 [J], 刘艳;张其
林;李祥超
5.用于大功率逆变器的大功率IGBT与硬驱动GTO的比较 [J], S.Bernet;吴茂杉因版权原因，仅展示原文概要，查看原文内容请购买。

大功率IGBT直流特性和动态特性的PSPICE仿真

电力电子技术在电工学科中非常活跃，由于电力电子电路的非线性时变特性和网络拓扑结构复杂性，
计算机仿真对于电力电子装置的研究、发和应用有着重要意义．何商品化的电力电子仿真软件如开任ＰＰＣＳＩＥ或Ｓｂｒａｅ等都不可能包含所有的器件模型，尤其是新型的电力电子器件．于电力电子器件的建对
ＩＢ的特性分析．ＧＴ
关键词：缘栅双极型晶体管；特性；真；型绝仿模中图分类号：２４３Ｕ６．文献标识码：Ａ文章编号：２３—３４（０２０ —０１ —００５７Ｘ２０）６７０５
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摘要：应用等效模拟的方法建立了ＩＴＰＰＣＧＢＳＩＥ仿真的直流模型和动态模型，大功率ＩＢ的直流特性和动对ＧＴ态特性进行了仿真分析．型的仿真值与器件的实际值相比，有特性均较吻合，明仿真模型适用于大功率模所表
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大功率IGBT直流特性和动态特性的PSPICE仿真

人修正函数Ｆ１（ＶＧＥ）后，ＩＧＢＴ与ＭＯＳＦＥＴ在同一驱动电压特性下建立的相应关系式如下…：
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现代电力电子技术之igbt建模与仿真

现代电力电子技术IGBT建模与仿真一、IGBT结构及工作原理自上世纪80年代绝缘栅双极型晶体管（IGBT）问世以来，逐渐取代了晶闸管和功率MOSFET等器件，在中频、中等功率变流领域获得了广泛的应用。

IGBT 克服了功率MOSFET高通态损耗的特性，同时保持了MOSFET门极电压驱动的优点。

IGBT是一种PNPN四层结构的器件，其结构剖面图和等效电路如图（1）所示。

(a) 剖面图(b) 达林顿等效结构图(1)IGBT结构剖面图及等效电路由图（1）（b）可知，IGBT相当于一个MOSFET和一个BJT的混合电路。

当在其栅极施加一个足够大的正向电压时，MOSFET内部将形成沟道，为晶体管提供基极电流，从而使IGBT导通。

此时由于P+区的空穴注入到N-区产生电导调制效应，能够减小N-区的电阻，从而使IGBT具有较小的通态压降。

二、IGBT工作特性IGBT的工作特性分为静态特性和动态特性两种。

(1)静态特性静态特性描述了稳态情况下IGBT的电流与电压关系，最常用的是其伏安特性和转移特性。

伏安特性指的是在不同的Vge下，Ice与Vce之间的关系，如图（2）左图所示。

转移特性是指集电极电流Ic与栅射电压Uge之间的关系，如图（2）右图所示。

图(2)IGBT的静态特性(2)动态特性动态特性描述了开关过程中IGBT的电压电流随时间变化的关系，分为开通特性和关断特性。

(a)开通过程 (b)关断过程图(3)IGBT的动态特性在开通过程中有两点值得关注：一是电流Ic上升率较快时，快恢复二极管的反向恢复电流将导致Ic出现尖峰，这一尖峰会引起电磁干扰等问题；二是寄生电容Cgc导致Miller效应，使Vge出现Miller平台，增加开通损耗。

在关断过程中，Cgc的分流作用使得在Vce下降过程中同样会出现Miller 平台，增加关断损耗。

此外电流下降过程中，二极管偏置导通将引起电压过冲，导致电磁干扰问题。

由于MOSFET快速关断，PNP双极管中存储的电荷不能及时释放，关断过程中还会有一个较长的拖尾电流，也增加了关断损耗。

完整word版,【20140421】IGBT Spice Model导入与验证

如何在Pspice导入IGBT Spice Model并仿真验证2014-04-20 1 IGBT模型文件导入已下载IGBT Spice Model文件sgxxxn60.lib，导入到Pspice的步骤如下：1）打开Pspice Model Editor Student，进行模型编辑，如图1所示。

图 1 打开模型编辑2）在模型编辑器的File菜单中选择New。

3）将要导入的模型放在Pspice下的路径C:\Program Files\OrCAD_Demo\Capture\Library\Pspice，如图2，下一步，在Model菜单中选择Import并找到模型文件sgxxxn60.lib，打开。

②①图 2 Import模型文件4）然后创建Capture的元件符号。

模型文件（.lib ）处于打开状态时，选择File>Create Capture Parts。

③①②图 3 创建Capture元件符号弹出对话框如图3所示，在Enter Input Model Library点击Browser，选择模型文件，输出元件库会自动出现，但是文件名的扩展名为.olb。

5）点击OK按钮，一个.err文件窗口将打开，显示创建库的状态，检查有无错误提示，在状态窗口点击OK，完成符号文件创建。

2 编辑IGBT器件Pspice符号本节说明怎样用模型文件为模型文件中的器件创建相应的元件符号。

1）打开OrCAD Capture，如图4所示。

图4 启动Capture2）下拉菜单File>Open>Library，浏览创建的符号文件（SGXXXN60.OLB），点击打开，出现PCB 窗口，能看到sgxxxn60库中包含不同的器件，如图5所示。

图5 浏览符号文件.olb3）双击其中某个器件，以SGP02N60为例，出现如图6所示窗口，即为器件的原始生成符号，器件符号上的红线对应管脚，其中GATE为IGBT的栅极，ANO（anode阳极）为IGBT的集电极，KAT（kathode 阴极）为IGBT的发射极。

Pspice仿真类型及不同电源参数

图1-8 VSFFM属性设置框
VSFFM属性设置框中各项参数的含义及单位见表1-3。
表1-3 VSFFM的属性参数
参数
含义
单位
VOFF
直流偏移电压
伏特
VAMPL
振幅
伏特
FC
载波频率
赫兹
FM
调制频率
赫兹
MOD
调制因子
无
按图1-15设置参数的VSFFM波形如图1-16所示。
图1-9 VSFFM波形
e)指数信号（VEXP、IEXP）
设置完毕，点击确定按钮。
图1-10 Simulation Settings
3．进行电路仿真
（1）执行菜单命令PSpice/Run，或点击工具按钮，调用PSpice A/D软件对该电路图进行仿真模拟。
（2）依次点击工具按钮、、，则电路图上相应位置依次显示节点电压、支路电流及各元器件上的功率损耗。如图1-29所示。
以上各项填完之后，按确定按钮，即可完成仿真分析类型及分析参数的设置。
另外，如果要修改电路的分析类型或分析参数，可执行菜单命令PSpice/Edit Simulation Profile，或点击工具按钮，在弹出的对话框中作相应修改。
（3）电路的模拟仿真
a）PSpice A/D视窗的启动
执行菜单命令PSpice/Run，或点击工具按钮，即可启动PSpice A/D视窗执行电路的仿真模拟，并且系统可自动调用Probe模块，对模拟结果进行后处理，屏幕显示如图1-5所示。
图1-11 VEXP波形
l瞬态分析的应用
现在通过举例，来说明瞬态分析的应用方法。
例：图1-19所示电路的电压源为分段线性源，其波形如图1-20所示。试对该电路进行瞬态分析。

基于PSpice的IGBT擎住效应的仿真教学分析

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【教程】PSpice地4种基本仿真分析报告详解

【教程】PSpice的4种基本仿真分析详解PSpice A/D将直流工作点分析、直流扫描分析、交流扫描分析和瞬态TRAN分析作为4种基本分析类型，每一种电路的模拟分析只能包括上述4种基本分析类型中的一种，但可以同时包括参数分析、蒙特卡罗分析、及温度特性分析等其他类型的分析，现对4种基本分析类型简介如下。

1. 直流扫描分析（DC Sweep）直流扫描分析的适用范围：当电路中某一参数（可定义为自变量）在一定范围内变化时，对应自变量的每一个取值，计算出电路中的各直流偏压值（可定义为输出变量），并可以应用Probe功能观察输出变量的特性曲线。

例对图1所示电路作直流扫描分析图1（1）绘图应用OrCAD/Capture软件绘制好的电路图如图2所示。

图2（2）确定分析类型及设置分析参数a) Simulation Setting（分析类型及参数设置对话框）的进入•执行菜单命令PSpice/New Simulation Profile，或点击工具按钮，屏幕上弹出New Simulation（新的仿真项目设置对话框）。

如图3所示。

图3•在Name文本框中键入该仿真项目的名字，点击Create按钮，即可进入Simulation Settings （分析类型及参数设置对话框），如图4所示。

图4b）仿真分析类型分析参数的设置图2所示直流分压电路的仿真类型及参数设置如下（见图4）：•Analysis type下拉菜单选中“DC Sweep”；•Options下拉菜单选中“Primary Sweep”；•Sweep variable项选中“Voltage source”，并在Name栏键入“V1”；•Sweep type项选中“Linear”，并在Start栏键入“0”、End栏键入“10”及Increment栏键入“1”。

以上各项填完之后，按确定按钮，即可完成仿真分析类型及分析参数的设置。

另外，如果要修改电路的分析类型或分析参数，可执行菜单命令PSpice/Edit Simulation Profile，或点击工具按钮，在弹出的对话框中作相应修改。

PSPICE仿真

PSPICE仿真目录介绍： (3)新建PSpice仿真 (5)新建项目 (5)放置元器件并连接 (5)生成网表 (9)指定分析和仿真类型 (9)Simulation Profile设置： (11)开始仿真 (12)参量扫描 (14)Pspice模型相关 (18)PSpice模型选择 (18)查看PSpice模型 (18)PSpice模型的建立 (20)介绍：PSpice是一种强大的通用模拟混合模式电路仿真器，可以用于验证电路设计并且预知电路行为，这对于集成电路特别重要。

PSpice可以进行各种类型的电路分析。

最重要的有：●非线性直流分析：计算直流传递曲线。

●非线性瞬态和傅里叶分析：在打信号时计算作为时间函数的电压和电流；傅里叶分析给出频谱。

●线性交流分析：计算作为频率函数的输出，并产生波特图。

●噪声分析●参量分析●蒙特卡洛分析PSpice有标准元件的模拟和数字电路库（例如：NAND，NOR，触发器，多选器，FPGA，PLDs 和许多数字元件）分析都可以在不同温度下进行。

默认温度为300K电路可以包含下面的元件：●Independent and dependent voltage andcurrent sources 独立和非独立的电压、电流源●Resistors 电阻●Capacitors 电容●Inductors 电感●Mutual inductors 互感器●Transmission lines 传输线●Operational amplifiers 运算放大器●Switches 开关●Diodes 二极管●Bipolar transistors 双极型晶体管●MOS transistors 金属氧化物场效应晶体管●JFET 结型场效应晶体管●MESFET 金属半导体场效应晶体管●Digital gates 数字门●其他元件 (见用户手册)。

新建PSpice仿真新建项目如图1所示，打开OrCAD Capture CIS Lite Edition，创建新项目：File > New > project。

IGBT Model in SPICE

The concept of IGBT modellingand the evaluation of the PSPICE® IGBT modelMaster ThesisbyJoakim Karlssonconducted atALSTOM Power, Växjö2002AbstractThe ability to calculate power losses in the Insulated Gate Bipolar Transistors (IGBTs) embedded in a series resonant converter is of great interest at ALSTOM Power, Växjö. Correctly estimated power losses are valuable in various situations, e.g. when evaluating new types of IGBTs or different control strategies. Therefore, it is desirable to investigate and evaluate the existing computer based IGBT models, where the majority is implemented in various circuit simulators.In this thesis, the MicroSim® PSPICE® IGBT model is evaluated with respect to the application of the series resonant converter. Particularly, this involves the issues of conduction losses and switching losses both at hard and soft switching.To be able to characterise a specific IGBT, a major parameter extraction procedure is performed, adapted from Hefner [7]. The parameters are subsequently used in the simulations and the result is compared with reference values of the measured power losses in the series resonant converter. This procedure is carried out for three different IGBTs from different manufacturers (Eupec, Semikron and Toshiba).The evaluated model failed to comply with the stated specifications, mainly owing to the poor performance at soft switching with a capacitive snubber, and consequently, some possible future alternatives are given.AcknowledgementsFist of all, I sincerely would like to thank my supervisor at the department of Industrial Electrical Engineering and Automation (IEA), Per Karlsson, for his support and advice during this thesis.Secondly, I am also very grateful to Per Ranstad, supervisor at ALSTOM Power, Växjö, for the possibility of conducting my master thesis at ALSTOM and for the continuous assistance and encouragement during my work.I also would like to thank the rest of the staff at ALSTOM for making my time at the company very enjoyable.Contents1 INTRODUCTION (3)1.1 B ACKGROUND (3)1.2 W ORKING PROCESS (3)1.3 T HESIS OUTLINE (4)2 CONVERTER TOPOLOGY (5)2.1 S NUBBER CIRCUITS (5)Hard switching (5)The capacitive snubber (6)The charge-discharge RCD snubber (7)2.2 T HE SERIES RESONANT CONVERTER (7)3 POWER SEMICONDUCTORS (10)3.1 P OWER DIODES (10)Basic structure and i-v characteristics (10)Non-punch-through and punch-through diodes (11)On-state losses and conductivity modulation (11)Switching characteristics (11)Important features of bipolar devices (12)Schottky diodes (12)3.2 B IPOLAR J UNCTION T RANSISTORS (BJT S) (13)Basic structure and i-v characteristics (13)Gain mechanism (14)Switching characteristics (15)Second breakdown (17)Safe Operating Area (SOA) (17)3.3 M ETAL O XIDE S EMICONDUCTOR F IELD E FFECT T RANSISTORS (MOSFET S) (18)Basic structure (18)i-v characteristics (18)Inversion layer and field effect (19)Switching characteristics (20)Conduction losses and breakdown voltages (22)Safe operating Area (SOA) (22)3.4 I NSULATED G ATE B IPOLAR T RANSISTORS (IGBT S) (22)On-state operation (23)Switching characteristics (23)Latchup (24)4 IGBT MODELS (26)4.1 IGBT MODEL CATEGORIES (26)A. Behavioural models (26)B. Semimathematical models (26)C. Mathematical models (27)D. Seminumerical models (27)4.2 T HE M ICRO S IM®PSPICE® MODEL (27)5 PARAMETER EXTRACTION (29)5.1 P ARAMETER EXTRACTION PROCEDURE (29)Measurements (29)1:Tail Decay Rate (29)2: Tail size versus anode current for different anode voltages (30)3: Saturation current versus gate voltage (32)4: On-state voltage versus gate voltage (32)5: Gate and gate-drain charge (32)5.2 I MPLEMENTATION OF PARAMETER EXTRACTION PROCEDURE (34)The clamped inductive load circuit and the constant anode supply circuit (34)The ±15 V gate drive circuit (35)The current generator (35)5.3 R ESULTS OF THE PARAMETER EXTRACTION (35)5.4 D ISCUSSION (36)6 SIMULATION (37)6.1 S IMULATION IN PSPICE (37)6.2 S IMULATION OF THE SERIES RESONANT CONVERTER (37)6.3 R ESULTS (38)6.4 D ISCUSSION (39)7 CONCLUSIONS (40)7.1 R ESULTS (40)7.2 F UTURE WORK (40)REFERENCES (41)APPENDIX A (42)A.1 O PTIMAL VALUE OF SNUBBER CAPACITOR (42)A.2 U NDAMPED SERIES RESONANT CIRCUIT (42)APPENDIX B (43) (43)B.1 D ERIVATION OF βSSB.2 M EASUREMENT INSTRUMENTS (44)APPENDIX C (45)C.1 PSPICE SIMULATION MODEL OF THE SERIES RESONANT CONVERTER (45)C.2 S IMULATION OUTPUT - E UPEC (46)C.3 P OWER LOSSES - P ARAMETER SET: "O PTIMISED1” (47)C.4 P OWER LOSSES - P ARAMETER SET: "O PTIMISED2" (48)C.5P ARAMETER SETS FOR THE PSPICE SIMULATION (49)1 Introduction1.1 BackgroundALSTOM Power Environmental Systems produces and develops high frequency power converters (20 kHz, 100kW), for power supply in industrial applications. The converter includes a full-bridge inverter, Figure 1.1, where Insulated Gate Bipolar Transistors (IGBTs) constitute the switches.When replacing switching devices by different models, or when developing new systems and strategies, switching and conduction losses are essential to know for the evaluation of the system. At ALSTOM today, losses are partly manually measured, which is a very time consuming process. A more efficient way of estimating losses would be to use a computer-model for calculation or simulation, which may also provide a more detailed study of the various components constituting the total losses. The purpose of this thesis is to examine existing models and to evolve and evaluate an appropriate model.The model should be characterised by either information from datasheets or from measured characteristics. The dynamic characteristics and the performance at soft switching are of special interest. Moreover, the model should hold a margin of error of no more than 10 %.1.2 Working processThe starting point of this master thesis is the literary study, which starts with literature of power electronics to get basic background knowledge of converter topologies and semiconductors. To improve the understanding of converter design, the MicroSim® PSPICE® circuit simulator is used for simulations of fundamental converter topologies.The subsequent survey of IGBT models involves a major information retrieval, basically with the Internet and the IEEE database of scientific papers, IEEE Xplore™, as the principal source. After evaluation of the literary study, the conclusion is drawn that no simple model, for loss calculations only, is to be found and that a simulation model within a commercial simulator is the appropriate alternative. The final decision is made to evaluate the MicroSim® PSPICE® model since this is already available at ALSTOM and seems to have the potential of desired accuracy.The literary study also brought forward the importance of a parameter extraction procedure to characterise different devices correctly. Hefner [3]-[8] has done extensive research in the area of modelling semiconductors and since much of his work composes the basis of the PSPICE® model, also the parameter extraction procedure developed by him is adopted. Further simulations are executed to verify the procedures and circuits for the chosen set of parameters. Subsequently, design and construction of the extraction circuits as well as gate drive units are performed.The various measurements of the extraction procedure are logged on a computer, via an oscilloscope, to perform mathematical operations and curve fitting algorithms on the collected data. The processing of data is done in MATLAB® and the resulting parameters logged in an ordinary spreadsheet file.The final part of the evaluation process is to design a simulation model of the series resonant converter, including the IGBTs with adherent parameters, and to simulate the different scenarios in accordance with the reference measurements of the total losses.1.3 Thesis outlineIn Chapter 2 and 3, the series resonant converter, basic snubbers and relevant semiconductors are studied in order to present the fundamental facts to the reader. Some basic circuit- and semiconductor-knowledge is required, however, the important concepts are explained in general and the detailed studies are left to the interested reader.Chapter 4 involves the literary study made and the aspects of the evaluation. The parameter extraction procedure, as well as the designed circuits, is presented in the subsequent chapter, Chapter 5, along with results and comments. In Chapter 6 the simulation tool and the performed simulations are reviewed. Finally, the thesis is concluded in Chapter 7.2 Converter topologyMost electrical systems like computers, battery chargers, television sets etc. are operated at other voltage levels and functions than what is delivered by the line supply voltage. To adapt the voltage to the desired level,different kinds of converters are used and these can be categorised into ac/dc, dc/dc, ac/ac converters. Here ac/dc converters are assumed to have bi-directional flow of power and no distinction is made between conversion from ac to dc (rectifier ) or dc to ac (inverter). Not only the voltage level (amplitude) can be transformed with a converter; frequency and phase may also be altered arbitrarily.Depending on the requirements and prerequisites of the system, there are numerous different topologies for each specific converter category [19], [20]. Nevertheless, there are some factors that are common to all applications;the aim to minimise the losses during conversion and the building blocks of switches (semiconductor devices)and energy storage elements (inductors and capacitors). To enhance the first factor, additionally circuit elements may be added to the basic topologies to minimise overvoltages or the rate of rise of the voltage/current. In the first part of this section, the fundamentals of such elements, referred to as snubbers, are studied. In the subsequent section, the converter of special interest in this thesis, the series resonant converter, is investigated .For a more detailed understanding of different converter topologies and snubber circuits, refer to [19] – [22].2.1 Snubber circuitsSnubber circuits are used to reduce the stresses on semiconductor devices e.g. by minimising overvoltages during turn-off caused by stray inductances, limiting the rate of rise of the current during turn-on or enhance the switching trajectories to reduce the switching losses. There are multiple types of snubber circuits depending on the embedding circuit and type of semiconductor, as well as the purpose of the snubber. Consequently, diodes demand different snubbers than thyristors, turn-on snubbers differ from turn-off snubbers for transistors and additionally snubbers are needed in bridge applications [20].For the series resonant converter, as will be discussed later, the purely capacitive snubber is sufficient enough and is therefore, along with the RCD-snubber, the only snubber studied here.Hard switchingThe step down converter is a fundamental bridge configuration and is shown in Figure 2.1. The current source acts like an inductive load and the current is carried by the power transistor (IGBT in this case) during on-state,and by the freewheeling diode during off-state.Figure 2.1 The step down converter.In Figure 2.2, the collector current and collector-emitter voltage transients during switching of the step down converter are illustrated. The simulation values (in [21]) for the load current and the dc link voltage are I load =0.8I C and V dc = 0.6V CES , where I C is the rated maximum collector current and V CES is the rated maximum collector-emitter voltage. Note that the voltage and current are expressed in per unit of the normalisation values I C and V CES .During turn-on, the semiconductor is exposed to a high current peak as a consequence of the reverse recovery of the freewheeling diode (section 3.1). At the same time the collector-emitter voltage is still high, thus, causing high switching losses. During turn-off, the losses can be even higher due to the long collector current tail (section 3.4). The semiconductor is, under these conditions with simultaneously high current and voltage, said to be operating under hard switching.loadV dcFigure 2.2Normalised transistor current (black) and voltage (grey) during a) turn-on and b) turn-off of the power transistor in the stepdown converter [21].The capacitive snubberExtensive losses can cause critical junction temperatures or force the transistor to operate outside the Safe Operating Area and subsequently cause device failure. To avoid this, a snubber circuit is added to the converter,like in Figure 2.3.Figure 2.3 The step down converter with a capacitive snubber.The purpose of the capacitor, C s , added in parallel with the transistor, is to offer an alternative way for the load current at turn-off and simultaneously raise the collector-emitter voltage as in Figure 2.4b. This is not possible at hard switching, when the only alternative path for the current is through the freewheeling diode, which has to be forward biased to conduct, resulting in that approximately the entire dc link voltage, V dc , is applied across the transistor during the fall time of the current.Figure 2.4 Normalised transistor current (black) and voltage (grey) during a) turn-on and b) turn-off of the power transistor in the stepdown converter with a capacitive snubber. Note the collector current peak at turn-on, and the simultaneously change of thecollector current and collector-emitter voltage at turn-off [21].Switching with snubber circuit embedded is usually referred to as soft switching, or like in this case, zero voltage switching (ZVS), owing to the collector-emitter voltage level at the start of the turn-off switching sequence. Thisactually lowers the turn-off losses, and furthermore, can enhance the electromagnetic compatibility (EMC) byloadV dc C sreducing the derivative of the collector-emitter voltage. However, apparent from Figure 2.4a, the drawback of lower turn-off losses is the higher collector current peak at turn-on, adding stress to the transistor.At the instant of turn-on, the capacitor is fully charged and since the freewheeling diode is still conducting, the capacitor voltage equals the full dc link voltage. The diode becomes reversed biased at the peak of the reverse recovery current and at this moment the capacitor can be discharged through the transistor, causing the peak of the collector current during turn-on, Figure 2.4a.The value of the capacitor should be chosen in a way where the capacitor voltage reaches its off-state level at the same time as the current reaches zero [22], as in Figure 2.4b. If a lower capacitor value is applied, the capacitor voltage reaches the full dc link voltage before the current has reached zero, and if a higher value is chosen the discharge current at turn-on is increased. The mathematical expression for the optimal capacitor value can be found in Appendix A.1.The charge-discharge RCD snubberTo decrease the current stress of the transistor at turn-on, the discharge current has to be limited. This is done by a charge-discharge RCD snubber [21], further on only referred to as RCD snubber, shown in Figure 2.5.Figure 2.5 The step down converter with a RCD snubber.The diode is connected in parallel with the resistor to bypass the current at turn-off when the capacitive snubber is efficient, and the resistor only would cause higher losses. However, at turn-on, the capacitor is discharged through the resistor and the discharge current carried by the transistor is limited. During turn-on at hard switching, the transistor carries both the load current and the reverse recovery current, and at soft switching with a snubber, the transistor carries the sum of the load current and the discharge current when the freewheeling diode has recovered. Consequently, to not increase the stresses on the transistor, caused by the snubber compared to those caused by the freewheeling diode, the resistor value should be chosen in a manner to limit the discharge current to be lower than the reverse recovery current, I rr . This gives a resistor value ofHowever, the resistor value should not be chosen too high, since that would slow down the voltage decline and the capacitor should be fully discharged before the next turn-on to be efficient.2.2 The series resonant converterThe series resonant converter constitute one part of a group called load resonant converters [20]. As the name implies, the converter consist of a resonant part composed by inductors, capacitors and resistors and where usually the latter is part of the load. The purpose of this circuit is to offer soft switching (ZVS or ZCS),according to the resonant behaviour of the converter, resulting in lower switching losses, which in turn may facilitate a higher switching frequency.A series resonant DC to DC full-bridge converter is shown in Figure 2.6, where the resonant LC tank is connected to the load via a rectifier. Without the rectifier, the circuit is referred to as a series resonant DC to AC converter.A transformer may be included before the rectifier to provide an output voltage of desired magnitude as well as electrical isolation between in- and output.rrdcS I V R >V dc R s sFigure 2.6 A series resonant DC-DC full bridge converter.To simplify the analysis of the series resonant converter, only a half-bridge, Figure 2.7, is studied. The operation of the switches and the output waveforms are identical, with the addition of the operation of one switch in the half-bridge converter (e.g. T +) corresponding to the operation of two switches, in antiparallel (T A+ and T B-), in the full-bridge converter. The transistors in the full-bridge are in Figure 2.7a replaced by ideal switches, the resonant tank consists of L r and C r and the capacitance of the output capacitor, C f , is usually large to make the output ripple low.Figure 2.7 Series resonant converter: a) half-bridge b) equivalent circuit[20].The output waveform and the internal state of the converter is depending on the direction of the flow of the current and which device that is conducting. When i L is positive, it flows through the transistor, T +, if it is on and through the diode, D − , otherwise and the output voltage, V o , is reflected across the rectifier so that V B´B =V o .When the current reverses, it commutates to the opposite transistor and diode (T − and D +) and V B´B = -V o .This sequence of operation can be described by an equivalent circuit illustrated in Figure 2.7b, and the different states during one period are:od AB dAB o d AB dAB L o d AB dAB o d AB dAB L V V v V v conducting D V V v V v conducting T i for V V v V v conducting D V V v V v conducting T i for ++=+=+−=−=<−−=−=−+=+=>+−−+½½:½½:0½½:½½:0''''The derivation of the output waveforms for the basic undamped series-resonant circuit is given in Appendix A.2.The resonant converter has a resonance frequency equal toand the converter can be operated in different modes depending on whether the switching frequency is lower or higher than the resonance frequency. In this study, only continuos-conduction mode, with ωs > ω0, is considered.During this mode of operation, the switches turn-on at both zero voltage and zero current but are forced to turn off at a finite current. Starting at ω0t 0 in Figure 2.8, the transistor T + starts conducting the reversing positive current and when T + is gated off, at ω0t 1, T - is gated on. However, the current is still positive and therefore commutates to the diode, D − , but declines quickly towards zero because of the great negative voltage applied over the LC tank (- V d /2 – V o ). When the current becomes negative, it is carried by T − and an identical half-cycle as the first one is started.As mentioned earlier, during this mode of operation, the switches turn on at zero current and also (nearly) zero voltage (because the parallel freewheeling diode is conducting, hence, is forward biased, prior to turn-on). This means the reverse recovery characteristicof the freewheeling diode is not of importance since the current has reached zero at turn-off of the diode. Another advantage of the simultaneously ZVS and ZCS, is that lossless turn-off capacitors in parallel with the transistors can be used as turn-off snubbers.Figure 2.8 Series resonant converter, continuos conduction mode (ωs > ω0)[.20].rr C L f 1200==πω3 Power semiconductorsPower semiconductors are substantially different from ordinary logic-level semiconductors considering structure and characteristics. In this chapter the most important and basic features of the power diode, bipolar junction transistor (BJT), metal oxide semiconductor field effect transistor (MOSFET) and the insulated gate bipolar transistor (IGBT) are studied. The reader is assumed to have some basic knowledge of semiconductor physics and be familiar with common terms like the pn -junction, recombination, lifetime, drift and diffusion.The following subsections are a summary of the more comprehensive texts about semiconductors found in [20]and [22].3.1 Power diodesBasic structure and i-v characteristicsThe basic structure of the power diode is the simplest form of all semiconductors and the fundamental building block for all semiconductor devices. The cross-section of a power diode is illustrated in Figure 3.1a and it is realised by epitaxial growth of a lightly doped n - layer on top a heavily doped n + layer. The thickness of the n -layer, also referred to as the drift region, is dependent on the desired breakdown voltage. To form the pn -junction, a heavily doped p + region is diffused into the drift region.Figure 3.1 a) Cross section of power diode and b) circuit symbol.The feature that distinguishes the power diode from the signal level diode is the drift region, which is specific to all power semiconductors, and enables the depletion layer to grow into the drift region at reverse blocking voltages. Devices with high breakdown voltages, hence, thick drift regions, would logically have a high resistivity on account of its thick, lightly doped layer. However, the conductivity modulation (discussed later) of the drift region reduces this problem in all bipolar devices (i.e. devices utilising minority carrier injection).The typical i-v -characteristic of a power diode is shown in Figure 3.2. It resembles the characteristic of a signal level diode with the difference of an ohmic curvature, instead of an exponential, in forward bias. This is due to the ohmic resistance of the drift region appearing at the high current levels applied to power diodes. At reverse bias only a low leakage current is allowed to flow until the breakdown voltage, BV BD , is reached and avalanche breakdown occurs with device failure as a possible consequence.Figure 3.2 The i-vcharacteristic of the power diode.Anode(a)(b)d +-Non-punch-through and punch-through diodesThere are two different ways of designing the breakdown voltage rating of a diode. The first one is to make the drift region wide enough to maintain the entire depletion region, which is called non-punch-through diodes. The second approach is to have a more lightly doped drift region, where the depletion region extends all the way to the n+substrate, however, not much further into the layer because of the heavy doping in the substrate. Practically, this results in a punch-through diode with half the thickness of the drift region compared to a non-punch-through diode with the same rated breakdown voltage.On-state losses and conductivity modulationThe total losses of semiconductors can principally be divided into on-state losses (or conduction losses) and switching losses, ignoring off-state losses during reverse bias. For the power diode, at least for medium switching frequencies, the on-state losses are predominant and mostly dissipated in the drift region. For signal level diodes, the voltage across the diode under forward bias can be said to be constant and approximately equal to 0.7 V. If the current flowing through the diode is defined as I, the dissipated power would be given by P=0.7*I. Yet, this would lead to an underestimation of the losses in the power diode on account of the ohmic resistance in the drift region.Calculating this resistance on basis of geometry and equilibrium carrier densities, not considering the effect of the carrier injection into the drift region, also referred to as the conductivity modulation, would also lead to false results.When the pn-junction is forward biased, holes are injected from the anode into the drift region. At low-level injection these holes are in minority compared to the thermal equilibrium value of the electron concentration of the lightly doped n- drift region. However, at high-level injection the number of injected holes widely overcomes the quantity of electrons of the drift region, thus, creating a space charge of holes. This space charge attracts electrons from the n+substrate, which start to drift towards the injected holes while recombining along the way. This phenomenon is called double injection and results in a much lower ohmic resistance of the drift region than at thermal equilibrium.Switching characteristicsLosses during the switching period depend both upon the switching times as well as the curvature of the voltage and current during that time. Furthermore, these factors are affected by both the intrinsic properties of the device as well as the interaction with the surrounding circuit. An example of this is the time rate of change, di/dt, which often is controlled by circuit inductances or by the turn-off of a solid-state device where the diode serves as a freewheeling diode.In Figure 3.3 the switching period can be divided into five different time intervals, where t ri and t fr represent the turn-on transient and t fi and t rr the turn-off transient.Turn-on transientDuring t ri, the space charge within the depletion region, upheld by the high reverse bias, is removed and at the end of t ri, the depletion layer has reached its thermal equilibrium level. At the start of the second phase, t fr, the pn-junction is now forward biased and injection of minority carriers commences and the excess-carrier distribution grows towards the steady-state value supported by the current I0. The great voltage overshoot, built up during t ri, is due to the ohmic resistance in the drift region, where the conductivity modulation not yet is fully developed, in addition with the voltage contributed by stray inductances in the wafer and bonding wires. When the current has reached its maximum value, I0, di/dt approaches zero, hence, the voltage caused by stray inductances reduces along with the voltage drop over the drift region, which is shorted out by the vast amount of excess carriers. Nevertheless, the peak voltage during this period may reach several tens of volts and can affect the performance of the circuit.Turn-off transientThe turn-off sequence is basically the reverse of the turn-on sequence. Starting at t fi, the excess charge is removed from the drift region and this carrier sweep out is continued by the negative diode current during t rr1,in addition with the recombination process. When all the excess charge at the ends of the drift region is swept out, the junctions can become reversed biased (t rr2 ) and the depletion regions expand into the drift region from both。

IGBTPspice静态模型的仿真研究

［］胡波鸥， — —冷却联合装置［翁颐庆．振动流化干燥— １Ｃ］．第三届全国干燥会议论文集，１９８９．
，作者简介：李凤兰（女，高级工程师，主要从事产品质１９６２－）量检验管理工作。收稿日期：２０１０年６月５日
责任编辑李超
用Ｐ采用组合方ｓｉｃｅ仿真软件中基本的器件模型，ｐ法建立了Ｉ并对ＧＢＴ在Ｐｓｉｃｅ的ＣＡＤ静态模型，ｐ照实际器件进行了其特性仿真分析，验证了本文所建模型的正确性和实用性。
１ＩＧＢＴ的结构和特性
１．１ＩＧＢＴ的结构和基本功能绝缘栅双极型晶体管（ＩｎｓｕｌａｔｅｄＧａｔｅＢｉｏｌａｒｐ
《 · 数字技术与机械加工工艺装备２新技术新工艺》０１０年第８期
子、空穴浓度的平衡，出现了Ｎ－区电子浓度的增也就是在Ｎ－区产生了电导调制效应，使Ｉ加，ＧＢＴ具有较低的导通压降。因此本文在功率ＭＯＳＦＥＴ模型后加了电流控制电压源Ｈ来反应电导调制效应，它的电流是从集电极电流Ｉ受控源Ｃ上取样的，
集功率ＭＯＩＧＢＴ）Ｓ管绝缘栅双极型晶体管（和双极型晶体管Ｂ具有易驱动、低ＪＴ优点于一身，通态压降、较快开关速度等特点，因此在电力电子装置中作为开关元件而广泛应用于主回路中。然而，在实际应用中，常常因各种原因造成ＩＧＢＴ的损坏。应用建模与仿真技术研究Ｉ可以在ＧＢＴ功率器件，较短的时间内用较低的成本进一步优化器件的结构、工艺，研究器件的特性，提高器件的整体性能，从而提高功率器件ＩＧＢＴ的使用率。功率器件的建模一般有２种方法：一是根据功率半导体物理结构和方程建立模型，称为原始模型方法；二是采用存在的称为组合模型方法。器件模型进行组合，

IGBT的基本特性仿真研究

IGBT的基本特性仿真研究作者：郝保明许海峰李彪等来源：《赤峰学院学报·自然科学版》 2013年第15期郝保明，许海峰，李彪，陈欣欢（宿州学院机械与电子工程学院，安徽宿州 234000）摘要：本文首先从IGBT内部结构组成和工作原理两方面出发，阐述了一种新的简便的IGBT建模方法，利用此方法建立的模型参数数量大为减少，结构简单且参数容易提取，减小了模型计算量，对IGBT的基本特性进行了仿真研究.关键词：IGBT；模型；仿真研究中图分类号：TM769 文献标识码：A 文章编号：1673-260X（2013）08-0038-02PSpice自从问世以来，就迅速得到了广泛应用，并且具有快速、准确的仿真功能[1].虽然包括PSpice、Saber等在内的许多模拟软件具有很齐全的功能，能够进行准确模拟，但通常不能直接对IGBT和功率MOSFET器件性能进行仿真[2].原因是：第一，这些模拟软件通常并没有功率器件模型的库；第二，一般来说这些软件需要器件的初始条件，包括工艺数据等参数.1 IGBT的简化结构以NPN型IGBT为例，N-IGBT是三端器件，其基本结构是由N沟道的VDMOSFET与双极型晶体管BJT两部分所组成[3]，简化结构如图1所示：从简化结构可以看出IGBT相当于两个级：输入级和输出级.其中，输入级是MOSFET，输出级是PNP晶体管.输入控制着输出，是一种电压控制型器件.其开通和关断是由栅极和发射极间电压UGE所决定的.当给IGBT栅极一个正向电压降时，IGBT就能够自动导通；当给IGBT的栅极一个反向的负电压时，IGBT就会自动关断.流过MOSFET的电流为IGBT电流的主要部分，MOSFET的参数取值决定了IGBT的特性，包括其开关和存储时间等.2 IGBT的基本特性仿真2.1 静态特性IGBT的静态特性包括转移与输出共两种特性.所谓转移特性指的是IGBT的集电极电流IC 和IGBT的栅极与发射极间电压UGE的变化关系[4].仿真特性曲线电路如下图2所示，并对ZT1和ZT2两种状态的特性曲线分别作了对比.转移特性曲线(ZT1)如图3所示，横轴V_V2表示栅极与发射极间电压UGE，纵轴-I（R1）表示集电极电流IC.其中参数设置是：V_V2的变化范围是0~15V，其步长是1V.集电极与发射极间电压为30V，R1=0.05mΩ，仿真温度是25℃.阀值电压计算值为4.3V，与仿真波形显示相符.输出特性的曲线(ZT1)如图4所示，图中横坐标V_V1为IGBT的集电极与发射极间电压UGE，纵坐标-I(R1)是集电极电流IC.仿真参数设置是：V_V1的变化范围是0~20V，步长设置为1V.分别选取栅射极之间电压UGE为5V、8V、14V和15V共四个电压值，观察其输出特性.为增加模型仿真的可比性，需要考虑温度效应.取绝对温度T=368.15K，根据计算的模型参数带入仿真，可得在ZT2时的仿真特性曲线，如图5和图6所示.根据转移特性和输出特性曲线可以看出，温度变化影响着IGBT,温度效应影响着IGBT的内部参数.此方法适用于不同软件对IGBT的仿真，适用范围广.同时，结合PSpice软件的特点，可设定不同的仿真温度，从而根据不同温度下IGBT的输出特性曲线，重新取值计算模型参数.2.2 动态特性根据所示的仿真电路原理图，将各参数值代入元件模型进行仿真，其他参数采用默认值.栅极串联电阻Rg关系着开通和损耗的能量，取值越大，则其损耗就越大；但Rg对开关过程有着多方面的影响，若减小Rg，电流变化迅速也会导致反并联二极管反向峰值电流增大.因此，考虑到仿真实验的收敛性，取Rg为3Ω.激励信号采用脉冲源，周期选为100us，IGBT栅极驱动电压波形如图8所示.集射极间电压为300V，负载电感和电阻分别取0.035mH、50Ω.IGBT的输出电压以及电流如图9所示.其中，横坐标V(Cgc：1)指的是IGBT集射间电压UCE，纵坐标-I(L1)指的是负载电流，负号表示电流从L1的端点1流向端点2，仿真波形中导通压降在2V左右，与实际相符.3 总结本文首先从IGBT内部结构组成和工作原理两方面出发，阐述了一种新的简便的IGBT建模方法，运用PSpice软件结合厂商提供的特性曲线和特征参数，并考虑温度效应的影响，建立了IGBT仿真模型.参考文献：〔1〕梁中华,成燕,孙勇军，等.一种适合电路仿真的IGBT模型[J].沈阳工业大学学报,2001,23(11):18-21.〔2〕Bonsbaine,A., Trigkidis,G. and Benamrouche,N. An integrated electro thermal model of IGBT devices(experimental validation)[C].Proceedings of the 44th International Universities Power Engineering Conference(UPEC), Glasgow, 2009: 1-5.〔3〕Sigg,J.,Turkes,P.and Kraus,R. Parameter extraction methodology and validation for an electro thermal physics-based NPT IGBT model[C].Industry Applications Conference of IEEE(IAS), New Orleans,LA,1997,2:1166-1173.〔4〕康劲松,陶生桂,王新祺.大功率IGBT直流特性和动态特性的PSpice仿真[J].同济大学学报,2002,30(6):710-714.。

【20140421】IGBT Spice Model导入与验证

图1 打开模型编辑2）在模型编辑器的File菜单中选择New。

3）将要导入的模型放在Pspice下的路径C:\Program Files\OrCAD_Demo\Capture\Library\Pspice，如图2，下一步，在Model菜单中选择Import并找到模型文件sgxxxn60.lib，打开。

②①图2 Import模型文件4）然后创建Capture的元件符号。

模型文件（.lib ）处于打开状态时，选择File>Create Capture Parts。

③①②图3 创建Capture元件符号弹出对话框如图3所示，在Enter Input Model Library点击Browser，选择模型文件，输出元件库会自动出现，但是文件名的扩展名为.olb。

5）点击OK按钮，一个.err文件窗口将打开，显示创建库的状态，检查有无错误提示，在状态窗口点击OK，完成符号文件创建。

2 编辑IGBT器件Pspice符号本节说明怎样用模型文件为模型文件中的器件创建相应的元件符号。

1）打开OrCAD Capture，如图4所示。

图4 启动Capture2）下拉菜单File>Open>Library，浏览创建的符号文件（SGXXXN60.OLB），点击打开，出现PCB 窗口，能看到sgxxxn60库中包含不同的器件，如图5所示。

基于PSPICE仿真的IGBT功耗计算

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PSpice仿真(二)实验报告

PSpice仿真(二)实验报告课程名称：电路与模拟电子技术实验指导老师：张冶沁成绩：实验名称： PSpice的使用练习2 实验类型： EDA 同组学生姓名：一、实验目的和要求：1.熟悉ORCAD-PSPICE软件的使用方法。

2.加深对共射放大电路放大特性的理解。

3.学习共射放大电路的设计方法。

4.学习共射放大电路的仿真分析方法。

二、实验原理图：图1 三极管共射放大电路三、实验须知：1．静态工作点分析是指：答：求解静态工作点Q,在输入信号为零时,晶体管和场效应管各电极间的电流和电压就是Q 点。

可用估算法和图解法求解2．直流扫描分析是指：答：按照预定范围设置直流电压源变化值，观察电路的直流特性3．交流扫描分析是指：答：按照预定范围设置交流电压源变化值，观察电路的交流特性4．时域（瞬态）分析是指：答：控制系统在一定的输入下，根据输出量的时域表达式，分析系统的稳定性、瞬态和稳态性能5．参数扫描分析是指：答：在基本电路特性分析中，每个元器件的参数都取确定值，而在参数扫描分析中,将考虑由于参数变化引起的电路特性变化情况6．温度扫描分析是指：答：在电路参数固定的情况下，测试温度是对电路性能的影响大小7．写出PSpice仿真中调用元器件的模型库位置：答：在安装目录下的\tools\capture\library\pspice中，软件内使用place part可以调用8．PSpice仿真电路图中节点号为0（即接地）的参考节点的作用：为计算其他节点的电位值提供了计算标准。

参考节点通常取何种元器件：电源负极。

解决电路负载开路引起的悬浮节点的方法是：在开路节点和参考节点之间连接一个大阻值电阻。

9．电路图中设置节点别名的好处是：答：通过节点别名描述电路中各个元器件之间的连接关系，生成电连接网表文件；电路中不同位置的节点，只要节点名相同就表示在电学上是相连的；PSpice在模拟结束后，采用节点名表示电路特性分析的结果。

10．放置电源端子符号的好处是：答：放置端子的作用是把外部的输入信号通过端子引入到电路中,把电路上的输出信号通过端子引到外部的负载上。

高功率因数整流器的PSpice仿真

高功率因数整流器的PSpice仿真1 概述随着计算机辅助设计软件的涌现，越来越多的电子设计自动化（EDA）软件应运而生，成为设计人员的得力工具。

而PSpice 软件作为当今应用广泛的EDA 软件之一，是电子电路计算机辅助分析和设计中常用的一个通用软件，对电路模拟分析的精度较高，使设计人员不必搭焊实际电路，而直接进入计算机模拟分析阶段，非常方便。

兼之其PSpice 仿真计算中采用了精确的半导体器件模型，稀疏矩阵技术等，使得PSpice 仿真在数学、物理上的概念非常清晰且精度高、通用性好，可以模拟各种类型的电路。

近年来，电网的谐波污染和无功问题日益严重，主要原因是电力半导体器件及电力电子设备装置的广泛应用。

而特别针对于传统的开关电源，大量采用不控整流加电解电容滤波供电，其网侧电流波形为尖峰脉冲，输入电流波形严重畸变，功率因数非常低。

为满足要求，有必要采用功率因数校正技术。

本文研究的高功率因数整流器，主电路采用桥式整流，再级联以Boost 升压式斩波器作为功率因数校正环节，如高功率因数整流器的核心部分为开关管VT 的控制电路部分，本方案控制部分电路结构简单易懂，主要包括PI 调节器环节、乘法器环节、电压电流检测环节、开关管驱动环节以及滞环比较环节。

而其中的关键是滞环比较环节的设计，滞环比较环节电路原理滞环比较器是在电压比较器的基础上引入了正反馈，形成了滞环控制。

输出电压u 忆o 为+UZ 或-UZ，其中+UZ 和-UZ 为运算放大器的正负电源电3 高功率因数整流器的PSpice 仿真PSpice 通用电路模拟软件是能在微型计算机上运行的SPICE 程序。

SPICE 是Simulation Program with Intergrated Circuit Emphasis 的缩写，它由美国加州大学伯克利分校于20 世纪70 年代推出，主要用于集成电路的电路分析程序。

PSpice 除包含有SPICE 的仿真功能外，在计算的可靠性、收敛性及仿真速度等方面都有改进，并扩展了许多新功能。