A q-oscillator Green Function
实时时钟模块(I2C总线)PT7C4311说明书
Real-time Clock Module (I2C Bus)Features→→→→→→→→→→→→→→→→→。
DescriptionFunction BlockNotes:1. No purposely added lead. Fully EU Directive 2002/95/EC (RoHS), 2011/65/EU (RoHS 2) & 2015/863/EU (RoHS 3) compliant.2. See https:///quality/lead-free/ for more information about Diodes Incorporated’s definitions of Halogen - and Antimony-free, "Green" and Lead-free.3. Halogen- and Antimony-free "Green” products are defined as those which contain <900ppm bromine, <900ppm chlorine (<1500ppm total Br + Cl) and <1000ppmSOMaximum RatingsDC Electrical Characteristics(Unless otherwise specified, V= 1.5 ~ 5.5 V, T = -40 °C to +85 °C.)1.After switchover (V SO), V BAT (min) can be2.0V for crystal with R S=40kΩ.2.Switch-over and deselect point.3.Valid for Ambient Operating Temperature: T A = -40 to 85°C; V CC = 2.0 to 5.5V (except where noted). VCC fall time should not exceed 5mV/μs.4.All voltages referenced to GND.5.In 3.3V application, if initial battery voltage is ≥ 3.4V, it may be necessary to reduce battery voltage (i.e., through wave soldering thebattery) in order to avoid inadvertent switchover/reselection for VCC – 10% operation.6.For rechargeable backup, V BAT (max) may be considered to be V CC.AC Electrical CharacteristicsTiming DiagramRecommended Layout for Crystal1212 the equation as below:Cpar + [(C1+C G)*(C2+C D)]/ [(C1+C G)+(C2+C D)] =C LCpar is all parasitical capacitor between X1 and X2.C L is crystal’s load capacitance.Note: The crystal, traces and crystal input pinsshould be isolated from RF generating signals.Function DescriptionOverview of Functions1.Clock function2.Interface with CPU3.Oscillator enable/disable4.Calibration functionRegisters*1. PT7C4311 uses 6 bits for address. That is if write data to 41H, the data will be written to 01H address register.*2. Stop bit. When this bit is set to 1, oscillator and time count chain are both stopped.*3. CEB: Century Enable Bit. CB: Century Bit.*4. Control FT/OUT pin output DC level when 512Hz square wave is disabled.*5. Frequency Test. 512Hz square wave output is enabled at FT/OUT pin, which is using for frequency test.*6. Sign Bit. “1” indicates positive calibration; “0”indicates negative calibration.*7. Using for modifying count frequency. If 20ppm is wanted to slow down the count frequency, 10 (01010) should be loaded. *8. Initialize the control and status register to 10000000 if calibration function is not required.Clock calibrationCalibration:3.Time Counter∙∙∙* Note 2: Do not care.* Note 3: Century Enable Bit and Century Bit.4.Days of the week Counter5.Calendar Counter∙∙Communication1.I2C Bus Interfacea)Overview of I2C-BUSb)System ConfigurationFig.1 System configurationc)Starting and Stopping I2C Bus Communications∙∙∙d)Data Transfers and Acknowledge Responses during I2C-BUS Communication∙Data transfers*Note: with caution that if the SDA data is changed while the SCL line is at high level, it will be treated as a START, RESTART, or STOP condition.Fig.2 Starting and stopping on I2C busData acknowledge response (ACK signal)e)Slave Address2.I2C Bus’s Basic Transfer FormatSCL from Master1289SDA from transmitter(sending side)SDA from receiver(receiving side)Release SDALow activeACK signala)Write via I2C busb)Read via I2C bus∙Standard read∙Simplified readNote:1.The above steps are an example of transfers of one or two bytes only. There is no limit to the number of bytes transferredduring actual communications.2.49H, 4AH are used as test mode address. Customer should not use the addresses.Part MarkingW Package ZE PackagePackaging Mechanical 8- SOIC (W)8- TDFN (ZE)Ordering Information1.No purposely added lead. Fully EU Directive 2002/95/EC (RoHS), 2011/65/EU (RoHS 2) & 2015/863/EU (RoHS 3) compliant.2.See https:///quality/lead-free/ for more information about Diodes Incorporated’s definitions of Halogen- and Antimony-free, "Green" andLead-free.3.Halogen- and Antimony-free "Green” produ cts are defined as those which contain <900ppm bromine, <900ppm chlorine (<1500ppm total Br + Cl) and<1000ppm antimony compounds.4. 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Microscopic Description of the Breathing Mode and
For a spherical case the HF equations can be reduced to,
82hm2*(r)R"(r)l(lr21)R(r)ddr82hm2*(r)R' (r)
2(rA,A,A)
... ...
A A((rr21,, 1 2,,12))
...A(rA,A,A)
In the spherical case, the single-particle wave function i(r,,) is given in terms of the radial R (r), the spherical spin harmonic Yjlm(r,) , and the isospin m ( ) functions:
We consider the isoscalar breathing mode in which the neutrons and protons move in phase (∆T=0, ∆S=0).
o
(r,t)o( r,t)
In the scaling model, we have the matter density oscillates as
We use the scattering operator F
A
F f (ri ) i1
where f (r) 1 r2 for monopole excitation, to obtain the strength function 2
S (E ) 0 F n2(E E n ) 1 Im T (fr [ G f)] n
具有多项差分算子的三阶q-差分方程边值问题
具有多项差分算子的三阶q-差分方程边值问题杨小辉;李杰民【摘要】q-差分方程边值问题解的存在性已经引起国内外数学工作者的研究兴趣,并且得到许多有价值的结果.研究一类三阶q-差分方程边值问题,该问题是由一个三阶q-差分方程和3个具有多项q-差分算子为边界条件构成.这种边界条件可以看成是Sturm-Liouville边界条件的推广.利用Banach压缩映射原理和Krasnoselskii 不动点定理,获得了该类边值问题解的存在性和唯一性的充分条件.所得条件简洁,便于验证.结果推广和改进了已有文献中的定理.最后,举2个例子来演示所得结论的应用.【期刊名称】《四川师范大学学报(自然科学版)》【年(卷),期】2015(038)006【总页数】9页(P875-883)【关键词】q-差分方程;q-微分;q-积分;边值问题【作者】杨小辉;李杰民【作者单位】广东警官学院计算机系,广东广州510230;岭南师范学院数学与计算科学学院,广东湛江524048【正文语种】中文【中图分类】O175.7q-差分方程历史悠久[1-4],q-差分方程在多个学科中已得到应用[5-8].近年来q-差分方程解的存在性问题是数学工作者研究的中心问题之一[9-17].Sturm-Liouville型边值问题一直是大家关注的问题[18-21].B. Ahmad等[12]研究了三阶q-差分方程两点边值问题解的存在性,其中是标准三阶q-差分算子.C. L. Yu等[15]研究了三阶q-差分方程两点边值问题正解的存在性,其中,0<q<1,Iq={qn:n∈N}∪{0,1},f∈C([0,1]×[0,+∞),[0,+∞)),α,β≥0,α+β>0且(α-β)/(α+β)≤q,α、β、q都是常数,注意到边值问题(1)和(2)仅涉及到一个q-差分算子Dq,而涉及多项q-差分算子的三阶q-差分方程边值问题的研究较少.受到文献[12-13]的启发,本文研究具有4个q-差分算子的三阶q-差分方程两点边值问题其中,0<q<1,f∈C(I×R,R),I=[0,1],参数pi∈(0,1)(i=1,2,3),α、β、γ都是常数,且α,β,γ≥0,记首先介绍相关概念,然后给出2个引理.定义 2.1[8] 设0<q<1,g在R上有定义,g在t∈R点的q-差分为定义 2.2[8] 设0<q<1,g在R上有定义,g在t∈R点的高阶q-差分为定义 2.3[8] 设0<q<1,t>0,函数g(t):[0,t]→R在区间[0,t]上的q-积分记为Iqg(t),定义为). (该级数收敛)若g在[a,b]上有定义,函数g(t)定义在区间[a,b]上的q-积分定义为注意到IqDqg(t)=g(t)-g(0)(g(t)在t=0处连续).引理 2.4[8] q-差分算子有如下性质:r.引理 2.5 设y(t)∈C[0,1],则u为边值问题u(0)-αDp1u(0)=0,u(1)+βDp2u(1)=0,的解当且仅当β(1+p2)]y(s)dqs+(1-p2)dqs}.证明设u为(4)式的解.在[0,t]上对方程进行q-积分得到对(6)式在[0,t]上进行q-积分得到对(7)式在[0,t]上进行q-积分得到其中,a0、a1、a2是常数.当t≠0时,注意到}.又有还有同理此时,可知Dpiu(0)=a1.当t≠0时,于是有].类似(10)和(11)式可得利用(4)式的边值条件可以得到把(13)式代入(8)式,并令t=1得到由(9)式知所以利用u(1)+βDp2u(1)=0得到s.(15)式两边通乘以1+q,左边等于β(1+p2)+γβ(1+p3)]=a2Δ,右边等于β(1+p2)]y(s)dqs+s.整理得β(1+p2)]y(s)dqs+(1-p2)dqs}.把(16)式代入(13)式,可得a0和a1,把a0、a1和a2代入(8)式得到(5)式,所以u满足(5)式.反之,设u满足(5)式,容易验证u满足(4)式.证毕.为了进一步的分析,设X=C[I,R]表示从I到R的所有连续函数集合,定义范数‖X‖=sup{|x(t)|,t∈I}.这时X为Banach空间.记(1+p2)q](1-p2+β)}.定理 3.1 设f∈C(I×R,R),I=[0,1],且满足Lipschitz条件∀t∈I, u,v∈R,L为Lipschitz常数,则当LH<1时,(3)式有唯一解,其中H为(17)式定义.证明构造X上的非线性算子F为β(1+p2)]f(s,u(s))dqs+(1-p2)dqs}, u∈X.由f的连续性容易证明F:X→X是全连续算子,u为(18)式的解当且仅当u∈X为F 的不动点.设先取δ使LH≤δ<1,再取r使r≥MH/(1-δ).设Br={u∈X:‖u‖≤r},当u∈Br时,有|u(t)|≤r,t∈[0,1],所以β(1+p2)]f(s,u(s))dqs+(1-p2)dqs}|≤f(s,0)|+|f(s,0)|)dqs+(1+β)(1+q)qs+β(1+p2)]×(|f(s,u(s))-f(s,0)|+|f(s,0)|)dqs+(|f(s,u(s))-f(s,0)|+|f(s,0)|)dqs]}≤(1+β)(1+q)qs+β(1+p2)|dqs+β(1+p2)dqs+(1+p2)q](1-p2+β)]}≤(1+p2)q](1-p2+β)]}≤(Lr+M)H=LHr+MH≤LHr+(1-δ)r=(LH+1-δ)r≤r.(H为(17)式所定义.)这表明FBr⊂Br.设u,v∈X有[f(s,u(s))-f(s,v(s))]dqs+β(1+p2)][f(s,u(s))-f(s,v(s))]dqs+[f(s,u(s))-f(s,v(s))])dqs}|≤L‖u-v‖β(1+p2)|dqs+L‖u-v‖L‖u-v‖(1+p2)q](1-p2+β)]}=LH‖u-v‖.当LH<1时,F是压缩映射.由Banach压缩映射原理,F在Br内有唯一不动点u.利用引理2.5,u是(3)式的唯一解.引理 3.2[18](Krasnoselskii不动点定理) 假设K是Banach空间X的一个非空有界闭凸子集.若算子F1和F2是满足条件:(i) F1x+F2y∈K,x,y∈K;(ii) F1是全连续算子;(iii) F2是压缩算子,那么存在z∈K使得z=F1z+F2z.定理 3.3 设f∈C(I×R,R),且满足条件:(A1) |f(t,u)-f(t,v)|≤L|u-v|,L为Lipschitz常数;(A2) 存在φ∈C(I,R+)使得|f(t,u)|≤φ(t), ∀(t,u)∈I×R,若Lh<1,其中则(3)式至少有一解.证明设Banach空间X如第二节定义.算子F1和F2分别如下定义:(1+β)(1+q)qs+β(1+p2)]f(s,u(s))dqs+(1-p2)dqs}, u∈X.由引理3.2知u为(3)式的解当且仅当u满足u=F1u+F2u.设r≥‖φ‖H且固定,取K={u∈X:‖u‖≤r}.证明分3步完成.第1步:证当u,v∈K时,F1u+F2v∈K.β(1+p2)]f(s,v(s))dqs+β(1+p2)]|f(s,v(s))|d qs+(1-p2+β)|f(s,v(s))|/(1-p2)dqs}|≤‖φ‖(1+p2)q](1-p2+β)]}≤‖φ‖(1+p2)q](1-p2+β)]}≤‖φ‖H≤r.因此F1u+F2v∈K,这表明引理3.2的(i)成立.第2步:证F1是全连续算子.由条件(A2)知F1是连续,又K有界,于是可设∀t1,t2∈I,且t1<t2,u∈K有qs(t1-t2)]f(s,u(s))dqs|=→0, t1→t2.上式表明F1(K)是相对紧的.由Arzelá-Ascoli定理知F1在K上是紧的,所以F1是全连续算子.因此引理3.2的(ii)成立.第3步:证F2是压缩算子.设u,v∈K时有[f(s,u(s))-f(s,v(s))]dqs+[f(s,u(s))-f(s,v(s))]dqs}|≤|f(s,u(s))-f(s,v(s))|dqs+|f(s,u(s))-f(s,v(s))|dqs]}≤L‖u-v‖(1+p2)q](1-p2+β)]}≤Lh‖u-v‖.结合(19)式知F2是压缩算子.因此引理3.2的所有条件都成立.由引理3.2知存在u∈K满足u=F1u+F2u.所以(3)式至少有一解,即定理3.3成立.证毕.例 4.1 考查如下边值问题t∈[0,1], L>0,u(1)+D1/4u(1)=0,则当0<L<1/1.604 5时,(20)式有唯一解.证明与(3)式对应,易知q=1/2,p1=1/3,p2=1/4,p3=1/5,α=1/4,β=1,γ=1,容易验证Δ=99/20,H≈1.604 5,f=L[t3+cos t+1+sin u(t)],且|Lsin u-Lsin v|≤L|u-v|,当0<L<1/1.604 5时,有LH<1,所以定理3.1的条件完全满足,则(20)式有唯一解.证毕.例 4.2 考查如下边值问题t∈[0,1], L>0,u(1)+D1/4u(1)=0,则当0<L<3 213/4 070时,(21)式有唯一解.证明与(3)式对应,易知q=1/2,p1=1/3,p2=1/4,p3=1/5,α=1/4,β=1,γ=2,易算得Δ=153/20,h=4 070/3 213,且当0<L<3 213/4 070时,有Lh<1,所以定理3.3的条件完全满足,则(21)式有唯一解.证毕注 4.3 文献[12-13]中的定理不能应用到(20)和(21)式.致谢刘玉记教授对本文提供了指导,广东警官学院青年项目(2013-Q01)和湛江师范学院自然科学研究项目(QL1101)对本文给予了资助,谨致谢意.【相关文献】[1] Jackson F H. On q-difference equations[J]. Am J Math,1910,32(4):305-314.[2] Carmichael R D. The general theory of linear q-difference equations[J]. Am JMath,1912,34(2):147-168.[3] Mason T E. On properties of the solutions of linear q-difference equations with entire function coefficients[J]. Am J Math,1915,37(4):439-444.[4] Adams C R. On the linear ordinary q-difference equation[J]. Ann Math,1928,30(4):195-205.[5] Finkelstein R, Marcus E. Transformation theory of the q-oscillator[J]. J MathPhys,1995,36:2652-2672.[6] Finkelstein R.The q-Coulomb problem[J]. J Math Phys,1996,37:2628-2636.[7] Gasper G, Rahman M. Basic Hypergeometric Series[M]. Cambridge:Cambridge University Press,1990.[8] Kac V, Cheung P. Quantum Calculus[M]. New York:Springer-Verlag,2002:1-5.[9] Bangerezako G. Variational q-calculus[J]. J Math Anal Appl,2004,289(2):650-665.[10] Ahmad B, Ntouyas S K. Boundary value problems for q-difference inclusions[J]. Abst Appl Anal,2011,2011:15.[11] Ahmad B, Ahmed A, Ntouyas S K. A study of second-order q-difference equations with boundary conditions[J]. Adv Diff Eqns,2012,2012:35.[12] Ahmad B. Boundary-value problems for nonlinear third-order q-difference equations[J]. Electron J Diff Eqns,2011,94:1-7.[13] Wu G C. Variational iteration method for q-difference equations of second order[J]. J Appl Math,2012,2012:1-5.[14] Thiramanus P, Tariboon J. Nonlinear second-order q-difference equations with three-point boundary conditions[J]. Comput Appl Math,doi:10.1007/s40314-013-0067-x. [15] Yu C L, Wang J F. Eigenvalue of boundary value problem for nonlinear singular third-order q-difference equations[J]. Adv Diff Eqns,2014,2014:21.[16] Ntouyas S K, Tariboon J. Nonlocal boundary value problems for q-difference equations and inclusions[J]. Inter J Diff Eqns,2015,2015:1-12[17] Xu N, Zhong C P. Existence and properties of meromorphic solutions of some q-difference equations[J]. Adv Diff Eqns,2015,2015:16.[18] 赵亚明. 奇异Sturm-Liouville型方程与自然边界条件[J]. 四川师范大学学报:自然科学版,1987,15(3):37-42.[19] 罗卫华,吕晓亚,吴开腾. Sturm-Liouville边值问题的正解存在性及其界[J]. 四川师范大学学报:自然科学版,2011,34(1):55-58.[20] 倪黎,韦煜明,茹凯,等. 带p-Laplacian算子Sturm-Liouville型三点边值问题正解的存在性[J].广西师范学院学报:自然科学版,2013,30(1):26-28.[21] 王勇,张秋果,韦煜明,等. Sturm-Liouville边值问题三个正解的存在性[J]. 广西师范学院学报:自然科学版,2012,29(2):29-33.[22] 韦煜明,王勇,唐艳秋,等. 具p-Laplacian算子时滞微分方程边值问题解的存在唯一性[J]. 广西师范学院学报:自然科学版,2012,30(2):48-53.[23] 王媛. 二阶差分方程边值问题正解的存在性[J]. 西南大学学报:自然科学版,2009,31(7):58-62.[24] 顿调霞,李永祥. 一类三阶常微分方程的两点边值问题的正解[J]. 四川师范大学学报:自然科学版,2014,37(6):810-813.[25] Smart D R. Fixed Point Theorems[M]. Cambridge:Cambridge University Press,1980. 2010 MSC:39A13。
R7731A PWM 控制器商品说明书
R7731A-10 August 2011General DescriptionThe R7731A is a high-performance, low cost, low start-up current and current mode PWM controller with burst triple mode to support green mode power saving operation. The R7731A integrates functions of soft start, Under Voltage LockOut (UVLO), Leading Edge Blanking (LEB), Over Temperature Protection (OTP) and internal slope compensation. It provides the users a superior AC/DC power application of higher efficiency, low external component counts and lower cost solution.To protect the external power MOSFET from being damaged by supply over voltage, the R7731A output driver is clamped at 12V. Furthermore, R7731A features fruitful protections like Over Load Protection (OLP) and Over Voltage Protection (OVP) to eliminate the external protection circuits and provide reliable operation. R7731A is available in SOT-23-6 and DIP-8 packages.Burst Triple Mode PWM Flyback ControllerFeaturesz Very Low Start-up Current (<30μA)z 10/14V UVLOz Soft Start Function z Current Mode Controlz Jittering Switching Frequency z Internal Leading Edge Blanking z Built-in Slope Compensationz Burst Triple Mode PWM for Green-Mode z Cycle-by-Cycle Current Limit z Feedback Open Protection z Over Voltage Protectionz Over Temperature Protection z Over Load Protectionz Soft Driving for Reducing EMI z Driver Capability ±200mA z High Noise ImmunityzOpto-Coupler Short ProtectionzRoHS Compliant and Halogen FreeApplicationsz Adaptor and Battery Charger z ATX Standby Power z Set-Top Box (STB)z DVD and CD(R)z TV/Monitor Standby Power zPC PeripheralsOrdering InformationR7731AG : Green (Halogen Free and Pb Free)Note :Richtek products are :` RoHS compliant and compatible with the current require-ments of IPC/JEDEC J-STD-020.` Suitable for use in SnPb or Pb-free soldering processes.Marking InformationIDP= : Product Code W : Date CodeR7731AGN : Product Number YMDNN : Date CodeR7731AGERichTek R7731A GNYMDNNPin Configurations(TOP VIEW)SOT-23-6DIP-8GATE VDD NC CSRTNCCOMP GND VO+VO-* : See Application InformationR7731A-10 August 2011Function Block DiagramVDDRTGATEGNDAbsolute Maximum Ratings(Note 1)z Supply Input Voltage, V DD-------------------------------------------------------------------------------------------------−0.3V to 30V z GATE Pin----------------------------------------------------------------------------------------------------------------------−0.3V to 20V z RT, COMP, CS Pin----------------------------------------------------------------------------------------------------------−0.3V to 6.5V z I DD-------------------------------------------------------------------------------------------------------------------------------10mAz Power Dissipation, P D @ T A = 25°CSOT-23-6----------------------------------------------------------------------------------------------------------------------0.4WDIP-8---------------------------------------------------------------------------------------------------------------------------0.714Wz Package Thermal Resistance (Note 2)SOT-23-6, θJA-----------------------------------------------------------------------------------------------------------------250°C/WDIP-8, θJA----------------------------------------------------------------------------------------------------------------------140°C/Wz Junction T emperature-------------------------------------------------------------------------------------------------------150°Cz Lead Temperature (Soldering, 10 sec.)---------------------------------------------------------------------------------260°Cz Storage T emperature Range----------------------------------------------------------------------------------------------−65°C to 150°C z ESD Susceptibility (Note 3)HBM (Human Body Mode)------------------------------------------------------------------------------------------------4kVMM (Machine Mode)--------------------------------------------------------------------------------------------------------250V Recommended Operating Conditions (Note 4)z Supply Input Voltage, V DD-------------------------------------------------------------------------------------------------12V to 25Vz Operating Frequency-------------------------------------------------------------------------------------------------------50k to 130kHz z Junction T emperature Range----------------------------------------------------------------------------------------------−40°C to 125°C z Ambient T emperature Range----------------------------------------------------------------------------------------------−40°C to 85°CElectrical CharacteristicsTo be continuedNote 1. Stresses listed as the above "Absolute Maximum Ratings" may cause permanent damage to the device. These are for stress ratings. Functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may remain possibility to affect device reliability.Note 2. θJA is measured in the natural convection at T A = 25°C on a low effective single layer thermal conductivity test board of JEDEC 51-3 thermal measurement standard.Note 3. Devices are ESD sensitive. Handling precaution is recommended.Note 4. The device is not guaranteed to function outside its operating conditions.Note 5. Guaranteed by design.R7731A-10 August D MAX vs. Temperature7071727374757677787980-40-2020406080100120Temperature (°C)D M A X (%)Typical Operating CharacteristicsV TH vs. Temperature9101112131415-40-25-105203550658095110125Temperature (°C)V D D (V )I DD_ST vs. Temperature10121416182022242628-40-1510356085110135Temperature (°C)I D D _S T (µA)I DD_OP vs. Temperature1.251.301.351.401.451.501.55-40-1510356085110135Temperature (°C)I D D _O P (m A )f OSC vs. Temperature57585960616263-40-1510356085110135Temperature (°C)f O S C (k H z )V COMP vs. Temperature5.405.445.485.525.565.60-40-2020406080100120Temperature (°C)V C O M PR7731A-10 August 2011V CLAMP vs. V DD78910111213111213141516171819202122V DD (V)V C L A M P (V)V GATE_OFF vs. V DD400425450475500525550575600111213141516171819202122V DD (V)V G A T E _O F F (m V)I SUPPLY vs. Temperature0.300.350.400.450.50-40-2020406080100120Temperature (°C)I S U P P L Y (m A)I SUPPLY vs. V DD0.4080.4100.4120.4140.4160.4180.4200.4220.4240.426111213141516171819202122V DD (V)I S U P P L Y (m A)V CLAMP vs. Temperature10.010.511.011.512.012.513.0-40-1510356085110135Temperature (°C)V C L A M P (V)GATE vs. Temperature50100150200250300350-40-25-105203550658095110125Temperature (°C)G A T E (n s )V CSTH vs. Temperature0.8000.8150.8300.8450.8600.8750.890-40-25-105203550658095110125Temperature (°C)V C S T H (V )R7731A-10 August 2011Application InformationUVLOUnder Voltage LockOut (UVLO) block is to ensure V DD has reached proper operation voltage before we enable the whole IC blocks. To provide better temperature coefficient and precise UVLO threshold voltage, the reference voltage of hysteresis voltage (10V / 14V) is from band-gap block directly. By this way, R7731A can operate more reliable in different environments.The maximum start-up current (30μA) is only for leakage current of IC at UVLO(on)-0.1V. The external al-capacitor on VDD may have 5 to 6μA extra leakage current. So designed start-up current of the system should exceed 36μA or more and IC can start up normally. In addition,designed start-up current of system should be less than 380μA, and IC can work normally at hiccup mode.Jittering OscillatorFor better EMI performance, R7731A will operate the system with ±6% frequency deviation around setting frequency.To guarantee precise frequency, it is trimmed to 5%tolerance. It also generates slope compensation saw-tooth,75% maximum duty cycle pulse and overload protection slope. By adjusting resistor of RT pin according to the following formula :Figure 1. CompetitorV CSV OUTV CS(500mV/Div)V OUT (2V/Div))(k R 6500(kHz)f T OSCΩ=It can typically operate between 50kHz to 130kHz. Note that RT pin can 't be short or open otherwise oscillator will not operate.Built-in Slope CompensationTo reduce component counts, slope compensation is implemented by internal built-in saw-tooth. Since it 's built-in, it 's compromised between loop gain and sub-harmonic reduction. In general design, it can cancel sub-harmonic to 90Vac.Leading Edge Blanking (LEB)MOSFET C OSS , secondary rectifier reverse recovery current and gate driver sourcing current comprise initial current spike. The spike will seriously disturb current mode operation especially at light load and high line. R7731A provides built-in 420ns LEB to guarantee proper operation in diverse design.Noise ImmunityCurrent mode controller is very sensitive to noise. R7731A takes the advantages of Richtek long term experience in designing high noise immunity current mode circuit and layout. Also, we amplify current sense signal to compare with feedback signal instead of dividing feedback signal.All the effort is to provide clean and reliable current mode operation.Soft-StartDuring initial power on, especially at high line, current spike is kind of unlimited by current limit. Therefore,besides cycle-by-cycle current limiting, R7731A still provides soft-start function. It effectively suppresses the start-up current spike. As shown in the Figure 1 and Figure 2, the start-up V CS is about 0.3V lower than competitor. The typical soft-start duration is 4ms (R T =100k Ω). Again, this will provide more reliable operation and possibility to use smaller current rating power MOSFET .Gate DriverA totem pole gate driver is fine tuned to meet both EMI and efficiency requirement in low power application. An internal pull low circuit is activated after pretty low V DD to prevent external MOSFET from accidentally turning on during UVLO.Burst Triple ModeTo fulfill green mode requirement, there are 3 operation modes in R7731A. Please also refer to Figure 3 for details.`PWM ModeFor most of load condition, the circuit will run at traditional PWM current mode.`Burst ModeDuring light load, switching loss will dominate the power efficien cy calculation. This mode is to cut switching loss. As shown in Figure 3, when the output load gets light, feedback signal drops and touches V BURL (Typical value is 1.75V). Clock signal will be blanked and system ceases to switching. After V OUT drops and feedback signal goes back to V BURH (1.8V, typically), switching will be resumed. Burst mode so far is widely used in low power application because it 's simple, reliable and will not have any patent infringement issue.`VDD Holdup ModeWhen the V DD drops down to V DD turn off threshold voltage, the system will be shut down. During shut down period, controller does nothing to any load change and might cause V OUT down. To avoid this, when V DD drops to a setting threshold, 11V, the hysteresis comparator will bypass PWM and burst mode loop and force switching at a very lo w level to supply energy to VDD pin. The designed value is 11.25V with 0.5V hysteresis band.Furthermore, VDD holdup mode is only designed to prevent V DD from touching turn off threshold voltage under light load or load transient moment. Relative to burst mode, switching loss will increase on the system at VDD holdup mode, so it is highly recommended that the system should avoid operating at this mode during light load or no load condition, normally.Figure 3. Burst Triple ModeFigure 2. R7731AV CSV OUTV CS(500mV/Div)V OUT (2V/Div)V V V V V V VR7731A-10 August 2011ProtectionR7731A provides fruitful protection functions that intend to protect system from being damaged. All the protection functions can be listed as below :`Cycle-by-Cycle Current LimitThis is a basic but very useful function and it can be implemented easily in current mode controller.`Over Load ProtectionLong time cycle-by-cycle current limit will lead to system thermal stress. To further protect system, system will be shut down after about 4096 clock cycles. it 's about 60ms delay in 67kHz operation. After shutdown, system will resume and behave as hiccup. By proper start-up resistor design, thermal will be averaged to an acceptable level over the ON/OFF cycle of IC. This will last until fault is removed. #It's highly recommended to add a resistor in parallel with the opto-coupler. T o provide sufficient bias current to make TL-431 regulate properly,1.2k Ω resistor is suggested.`Brownout ProtectionDuring heavy load, this will trigger 60ms protection and shut down the system. If it 's in light load condition,system will be shut down after V DD is running low and triggers UVLO.` OVPOutput voltage can be roughly sensed by VDD pin.If the sensed voltage reaches 27V threshold, system will be shut down after 20μs deglitch delay.`Feedback Open and Opto-Coupler ShortThis will trigger OVP or 60ms delay protection. It depends on which one occurs first.`OTPInternal OTP function will protect the controller itself from suffering thermal stress and permanent damage. It stops the system from switching until the temperature is under threshold level. Meanwhile, if V DD reaches V DD turn off threshold voltage, system will hiccup till over temperature condition is gone.Figure 4. R-C Filter on CS PinNegative Voltage Spike on Each PinNegative voltage (< −0.3V) on each pin will cause substrate injection. It leads to controller damage or circuit false trigger. Generally, it happens at CS pin due to negative spike because of improper layout or inductive current sense resistor. Therefore, it is highly recommended to add a R-C filter to avoid CS pin damage, as shown in Figure 4. Proper layout and careful circuit design should be done to guarantee yield rate in mass production.Auxiliary Ground (c)ICGround (d)MOSFET Ground (b)Figure 5. PCB Layout GuideLayout ConsiderationA proper PCB layout can abate unknown noise interference and EMI issue in the switching power supply. Please refer to the guidelines when you want to design PCB layout for switching power supply:The current path (1) from bulk capacitor, transformer,MOSFET, Rcs return to bulk capacitor is a huge high frequency current loop. It must be as short as possible to decrease noise coupling and kept a space to other low voltage traces, such as IC control circuit paths, especially.Besides, the path (2) from RCD snubber circuit to MOSFET is also a high switching loop, too. So keep it as small as possible.It is good for reducing noise, output ripple and EMI issue to separate ground traces of bulk capacitor (a), MOSFET (b), auxiliary winding (c) and IC control circuit (d). Finally,connect them together on bulk capacitor ground (a). The areas of these ground traces should be kept large.Placing bypass capacitor for abating noise on IC is highly recommended. The bypass capacitor should be placed as close to controller as possible.To minimize reflected trace inductance and EMI minimize the area of the loop connecting the secondary winding,the output diode, and the output filter capacitor. In addition,provide sufficient copper area at the anode and cathode terminal of the diode for heatsinking. Provide a larger area at the quiet cathode terminal. A large anode area can increase high-frequency radiated EMI.R7731A-10 August 2011Outline DimensionA1HSOT-23-6 Surface Mount PackageRichtek Technology CorporationHeadquarter5F, No. 20, Taiyuen Street, Chupei City Hsinchu, Taiwan, R.O.C.Tel: (8863)5526789 Fax: (8863)5526611Information that is provided by Richtek Technology Corporation is believed to be accurate and reliable. Richtek reserves the right to make any change in circuit design,specification or other related things if necessary without notice at any time. No third party intellectual property infringement of the applications should be guaranteed by users when integrating Richtek products into any application. No legal responsibility for any said applications is assumed by Richtek.Richtek Technology CorporationTaipei Office (Marketing)5F, No. 95, Minchiuan Road, Hsintien City Taipei County, Taiwan, R.O.C.Tel: (8862)86672399 Fax: (8862)86672377Email:*********************8-Lead DIP Plastic Package。
Short Course for Qualcomm -- Analysis and Design of LC VCOs -- by B Razavi 2010 [good]
Analysis and Design of LC VCOs
April 19, 2010
Behzad Razavi Electrical Engineering Department University of California, Los Angeles
33
Analysis of Phase Noise
Need to answer these questions: • • How does noise injected by a device corrupts the phase? How much noise does each device inject? Tens of papers have been published on phase noise in oscillators. Many mechanisms result in phase noise. No single approach has been sufficient to give insight into all mechanisms. We follow two approaches here: - Approach I: based on time averages (a) the average spectrum of noise of a device while the noise spectrum varies with time. (b) the “average resistance” - Approach II: based on phase response of an oscillator to an injected impulse in the time domain [Hajimiri & Lee, JSSC, Feb. 98].
卫星导航相关术语速查表
Bureau International des Poids et Mesures (BIPM)
Butterworth filter
双相移键控(BPSK) 比特移位寄存器
BlockI,II,IIA,IIR,IIR-M,IIF(型)卫星 BOC(m,n)码 boxcar 滤波器
国际时间局
巴特沃思(Butterworth)滤波器
粗捕获(C/A)码 载噪比(C/N0) 载波相位测量 载波跟踪环
载波消除 铯原子钟 特征方程(式)
码片 码率 片式原子钟(CSAC) 园误差概率 钟噪声 闭环转移函数 码-载波发散 码钟 码分多址(CDMA) 码发生器 码相位测量 码转换 码消除 相干信号跟踪 冷启动 梳状函数 共视时间传递 复指数 结构干涉 辐射(方向)图可控天线 控制段 控制段误差 传统惯性参考系统 传统陆地参考系统
带宽,零点至零点 带宽扩展 基带 基带采样 基线 基本函数 北斗
双相偏置载频(BOC)
binary phase shift keying (BPSK) bit shift register
Block I, II, IIA, IIR, IIR-M, IIF satellites BOC(m,n) codes boxcar filter
量子力学索引英汉对照
21-centimeter line, 21厘米线AAbsorption, 吸收Addition of angular momenta, 角动量叠加Adiabatic approximation, 绝热近似Adiabatic process, 绝热过程Adjoint, 自伴的Agnostic position, 不可知论立场Aharonov-Bohm effect, 阿哈罗诺夫-玻姆效应Airy equation, 艾里方程;Airy function, 艾里函数Allowed energy, 允许能量Allowed transition, 允许跃迁Alpha decay, 衰变;Alpha particle, 粒子Angular equation, 角向方程Angular momentum, 角动量Anomalous magnetic moment, 反常磁矩Antibonding, 反键Anti-hermitian operator, 反厄米算符Associated Laguerre polynomial, 连带拉盖尔多项式Associated Legendre function, 连带勒让德多项式Atoms, 原子Average value, 平均值Azimuthal angle, 方位角Azimuthal quantum number, 角量子数BBalmer series, 巴尔末线系Band structure, 能带结构Baryon, 重子Berry's phase, 贝利相位Bessel functions, 贝塞尔函数Binding energy, 束缚能Binomial coefficient, 二项式系数Biot-Savart law, 毕奥-沙法尔定律Blackbody spectrum, 黑体谱Bloch's theorem, 布洛赫定理Bohr energies, 玻尔能量;Bohr magneton, 玻尔磁子;Bohr radius, 玻尔半径Boltzmann constant, 玻尔兹曼常数Bond, 化学键Born approximation, 玻恩近似Born's statistical interpretation, 玻恩统计诠释Bose condensation, 玻色凝聚Bose-Einstein distribution, 玻色-爱因斯坦分布Boson, 玻色子Bound state, 束缚态Boundary conditions, 边界条件Bra, 左矢Bulk modulus, 体积模量CCanonical commutation relations, 正则对易关系Canonical momentum, 正则动量Cauchy's integral formula, 柯西积分公式Centrifugal term, 离心项Chandrasekhar limit, 钱德拉赛卡极限Chemical potential, 化学势Classical electron radius, 经典电子半径Clebsch-Gordan coefficients, 克-高系数Coherent States, 相干态Collapse of wave function, 波函数塌缩Commutator, 对易子Compatible observables, 对易的可观测量Complete inner product space, 完备内积空间Completeness, 完备性Conductor, 导体Configuration, 位形Connection formulas, 连接公式Conservation, 守恒Conservative systems, 保守系Continuity equation, 连续性方程Continuous spectrum, 连续谱Continuous variables, 连续变量Contour integral, 围道积分Copenhagen interpretation, 哥本哈根诠释Coulomb barrier, 库仑势垒Coulomb potential, 库仑势Covalent bond, 共价键Critical temperature, 临界温度Cross-section, 截面Crystal, 晶体Cubic symmetry, 立方对称性Cyclotron motion, 螺旋运动DDarwin term, 达尔文项de Broglie formula, 德布罗意公式de Broglie wavelength, 德布罗意波长Decay mode, 衰变模式Degeneracy, 简并度Degeneracy pressure, 简并压Degenerate perturbation theory, 简并微扰论Degenerate states, 简并态Degrees of freedom, 自由度Delta-function barrier, 势垒Delta-function well, 势阱Derivative operator, 求导算符Determinant, 行列式Determinate state, 确定的态Deuterium, 氘Deuteron, 氘核Diagonal matrix, 对角矩阵Diagonalizable matrix, 对角化Differential cross-section, 微分截面Dipole moment, 偶极矩Dirac delta function, 狄拉克函数Dirac equation, 狄拉克方程Dirac notation, 狄拉克记号Dirac orthonormality, 狄拉克正交归一性Direct integral, 直接积分Discrete spectrum, 分立谱Discrete variable, 离散变量Dispersion relation, 色散关系Displacement operator, 位移算符Distinguishable particles, 可分辨粒子Distribution, 分布Doping, 掺杂Double well, 双势阱Dual space, 对偶空间Dynamic phase, 动力学相位EEffective nuclear charge, 有效核电荷Effective potential, 有效势Ehrenfest's theorem, 厄伦费斯特定理Eigenfunction, 本征函数Eigenvalue, 本征值Eigenvector, 本征矢Einstein's A and B coefficients, 爱因斯坦A,B系数;Einstein's mass-energy formula, 爱因斯坦质能公式Electric dipole, 电偶极Electric dipole moment, 电偶极矩Electric dipole radiation, 电偶极辐射Electric dipole transition, 电偶极跃迁Electric quadrupole transition, 电四极跃迁Electric field, 电场Electromagnetic wave, 电磁波Electron, 电子Emission, 发射Energy, 能量Energy-time uncertainty principle, 能量-时间不确定性关系Ensemble, 系综Equilibrium, 平衡Equipartition theorem, 配分函数Euler's formula, 欧拉公式Even function, 偶函数Exchange force, 交换力Exchange integral, 交换积分Exchange operator, 交换算符Excited state, 激发态Exclusion principle, 不相容原理Expectation value, 期待值FFermi-Dirac distribution, 费米-狄拉克分布Fermi energy, 费米能Fermi surface, 费米面Fermi temperature, 费米温度Fermi's golden rule, 费米黄金规则Fermion, 费米子Feynman diagram, 费曼图Feynman-Hellman theorem, 费曼-海尔曼定理Fine structure, 精细结构Fine structure constant, 精细结构常数Finite square well, 有限深方势阱First-order correction, 一级修正Flux quantization, 磁通量子化Forbidden transition, 禁戒跃迁Foucault pendulum, 傅科摆Fourier series, 傅里叶级数Fourier transform, 傅里叶变换Free electron, 自由电子Free electron density, 自由电子密度Free electron gas, 自由电子气Free particle, 自由粒子Function space, 函数空间Fusion, 聚变Gg-factor, g-因子Gamma function, 函数Gap, 能隙Gauge invariance, 规范不变性Gauge transformation, 规范变换Gaussian wave packet, 高斯波包Generalized function, 广义函数Generating function, 生成函数Generator, 生成元Geometric phase, 几何相位Geometric series, 几何级数Golden rule, 黄金规则"Good" quantum number, "好"量子数"Good" states, "好"的态Gradient, 梯度Gram-Schmidt orthogonalization, 格莱姆-施密特正交化法Graphical solution, 图解法Green's function, 格林函数Ground state, 基态Group theory, 群论Group velocity, 群速Gyromagnetic railo, 回转磁比值HHalf-integer angular momentum, 半整数角动量Half-life, 半衰期Hamiltonian, 哈密顿量Hankel functions, 汉克尔函数Hannay's angle, 哈内角Hard-sphere scattering, 硬球散射Harmonic oscillator, 谐振子Heisenberg picture, 海森堡绘景Heisenberg uncertainty principle, 海森堡不确定性关系Helium, 氦Helmholtz equation, 亥姆霍兹方程Hermite polynomials, 厄米多项式Hermitian conjugate, 厄米共轭Hermitian matrix, 厄米矩阵Hidden variables, 隐变量Hilbert space, 希尔伯特空间Hole, 空穴Hooke's law, 胡克定律Hund's rules, 洪特规则Hydrogen atom, 氢原子Hydrogen ion, 氢离子Hydrogen molecule, 氢分子Hydrogen molecule ion, 氢分子离子Hydrogenic atom, 类氢原子Hyperfine splitting, 超精细分裂IIdea gas, 理想气体Idempotent operaror, 幂等算符Identical particles, 全同粒子Identity operator, 恒等算符Impact parameter, 碰撞参数Impulse approximation, 脉冲近似Incident wave, 入射波Incoherent perturbation, 非相干微扰Incompatible observables, 不对易的可观测量Incompleteness, 不完备性Indeterminacy, 非确定性Indistinguishable particles, 不可分辨粒子Infinite spherical well, 无限深球势阱Infinite square well, 无限深方势阱Inner product, 内积Insulator, 绝缘体Integration by parts, 分部积分Intrinsic angular momentum, 内禀角动量Inverse beta decay, 逆衰变Inverse Fourier transform, 傅里叶逆变换KKet, 右矢Kinetic energy, 动能Kramers' relation, 克莱默斯关系Kronecker delta, 克劳尼克LLCAO technique, 原子轨道线性组合法Ladder operators, 阶梯算符Lagrange multiplier, 拉格朗日乘子Laguerre polynomial, 拉盖尔多项式Lamb shift, 兰姆移动Lande g-factor, 朗德g-因子Laplacian, 拉普拉斯的Larmor formula, 拉摩公式Larmor frequency, 拉摩频率Larmor precession, 拉摩进动Laser, 激光Legendre polynomial, 勒让德多项式Levi-Civita symbol, 列维-西维塔符号Lifetime, 寿命Linear algebra, 线性代数Linear combination, 线性组合Linear combination of atomic orbitals, 原子轨道的线性组合Linear operator, 线性算符Linear transformation, 线性变换Lorentz force law, 洛伦兹力定律Lowering operator, 下降算符Luminoscity, 照度Lyman series, 赖曼线系MMagnetic dipole, 磁偶极Magnetic dipole moment, 磁偶极矩Magnetic dipole transition, 磁偶极跃迁Magnetic field, 磁场Magnetic flux, 磁通量Magnetic quantum number, 磁量子数Magnetic resonance, 磁共振Many worlds interpretation, 多世界诠释Matrix, 矩阵;Matrix element, 矩阵元Maxwell-Boltzmann distribution, 麦克斯韦-玻尔兹曼分布Maxwell's equations, 麦克斯韦方程Mean value, 平均值Measurement, 测量Median value, 中位值Meson, 介子Metastable state, 亚稳态Minimum-uncertainty wave packet, 最小不确定度波包Molecule, 分子Momentum, 动量Momentum operator, 动量算符Momentum space wave function, 动量空间波函数Momentum transfer, 动量转移Most probable value, 最可几值Muon, 子Muon-catalysed fusion, 子催化的聚变Muonic hydrogen, 原子Muonium, 子素NNeumann function, 纽曼函数Neutrino oscillations, 中微子振荡Neutron star, 中子星Node, 节点Nomenclature, 术语Nondegenerate perturbationtheory, 非简并微扰论Non-normalizable function, 不可归一化的函数Normalization, 归一化Nuclear lifetime, 核寿命Nuclear magnetic resonance, 核磁共振Null vector, 零矢量OObservable, 可观测量Observer, 观测者Occupation number, 占有数Odd function, 奇函数Operator, 算符Optical theorem, 光学定理Orbital, 轨道的Orbital angular momentum, 轨道角动量Orthodox position, 正统立场Orthogonality, 正交性Orthogonalization, 正交化Orthohelium, 正氦Orthonormality, 正交归一性Orthorhombic symmetry, 斜方对称Overlap integral, 交叠积分PParahelium, 仲氦Partial wave amplitude, 分波幅Partial wave analysis, 分波法Paschen series, 帕邢线系Pauli exclusion principle, 泡利不相容原理Pauli spin matrices, 泡利自旋矩阵Periodic table, 周期表Perturbation theory, 微扰论Phase, 相位Phase shift, 相移Phase velocity, 相速Photon, 光子Planck's blackbody formula, 普朗克黑体辐射公式Planck's constant, 普朗克常数Polar angle, 极角Polarization, 极化Population inversion, 粒子数反转Position, 位置;Position operator, 位置算符Position-momentum uncertainty principles, 位置-动量不确定性关系Position space wave function, 坐标空间波函数Positronium, 电子偶素Potential energy, 势能Potential well, 势阱Power law potential, 幂律势Power series expansion, 幂级数展开Principal quantum number, 主量子数Probability, 几率Probability current, 几率流Probability density, 几率密度Projection operator, 投影算符Propagator, 传播子Proton, 质子QQuantum dynamics, 量子动力学Quantum electrodynamics, 量子电动力学Quantum number, 量子数Quantum statics, 量子统计Quantum statistical mechanics, 量子统计力学Quark, 夸克RRabi flopping frequency, 拉比翻转频率Radial equation, 径向方程Radial wave function, 径向波函数Radiation, 辐射Radius, 半径Raising operator, 上升算符Rayleigh's formula, 瑞利公式Realist position, 实在论立场Recursion formula, 递推公式Reduced mass, 约化质量Reflected wave, 反射波Reflection coefficient, 反射系数Relativistic correction, 相对论修正Rigid rotor, 刚性转子Rodrigues formula, 罗德里格斯公式Rotating wave approximation, 旋转波近似Rutherford scattering, 卢瑟福散射Rydberg constant, 里德堡常数Rydberg formula, 里德堡公式SScalar potential, 标势Scattering, 散射Scattering amplitude, 散射幅Scattering angle, 散射角Scattering matrix, 散射矩阵Scattering state, 散射态Schrodinger equation, 薛定谔方程Schrodinger picture, 薛定谔绘景Schwarz inequality, 施瓦兹不等式Screening, 屏蔽Second-order correction, 二级修正Selection rules, 选择定则Semiconductor, 半导体Separable solutions, 分离变量解Separation of variables, 变量分离Shell, 壳Simple harmonic oscillator, 简谐振子Simultaneous diagonalization, 同时对角化Singlet state, 单态Slater determinant, 斯拉特行列式Soft-sphere scattering, 软球散射Solenoid, 螺线管Solids, 固体Spectral decomposition, 谱分解Spectrum, 谱Spherical Bessel functions, 球贝塞尔函数Spherical coordinates, 球坐标Spherical Hankel functions, 球汉克尔函数Spherical harmonics, 球谐函数Spherical Neumann functions, 球纽曼函数Spin, 自旋Spin matrices, 自旋矩阵Spin-orbit coupling, 自旋-轨道耦合Spin-orbit interaction, 自旋-轨道相互作用Spinor, 旋量Spin-spin coupling, 自旋-自旋耦合Spontaneous emission, 自发辐射Square-integrable function, 平方可积函数Square well, 方势阱Standard deviation, 标准偏差Stark effect, 斯塔克效应Stationary state, 定态Statistical interpretation, 统计诠释Statistical mechanics, 统计力学Stefan-Boltzmann law, 斯特番-玻尔兹曼定律Step function, 阶跃函数Stem-Gerlach experiment, 斯特恩-盖拉赫实验Stimulated emission, 受激辐射Stirling's approximation, 斯特林近似Superconductor, 超导体Symmetrization, 对称化Symmetry, 对称TTaylor series, 泰勒级数Temperature, 温度Tetragonal symmetry, 正方对称Thermal equilibrium, 热平衡Thomas precession, 托马斯进动Time-dependent perturbation theory, 含时微扰论Time-dependent Schrodinger equation, 含时薛定谔方程Time-independent perturbation theory, 定态微扰论Time-independent Schrodinger equation, 定态薛定谔方程Total cross-section, 总截面Transfer matrix, 转移矩阵Transformation, 变换Transition, 跃迁;Transition probability, 跃迁几率Transition rate, 跃迁速率Translation,平移Transmission coefficient, 透射系数Transmitted wave, 透射波Trial wave function, 试探波函数Triplet state, 三重态Tunneling, 隧穿Turning points, 回转点Two-fold degeneracy , 二重简并Two-level systems, 二能级体系UUncertainty principle, 不确定性关系Unstable particles, 不稳定粒子VValence electron, 价电子Van der Waals interaction, 范德瓦尔斯相互作用Variables, 变量Variance, 方差Variational principle, 变分原理Vector, 矢量Vector potential, 矢势Velocity, 速度Vertex factor, 顶角因子Virial theorem, 维里定理WWave function, 波函数Wavelength, 波长Wave number, 波数Wave packet, 波包Wave vector, 波矢White dwarf, 白矮星Wien's displacement law, 维恩位移定律YYukawa potential, 汤川势ZZeeman effect, 塞曼效应。
绿光泵浦的黄光波段可调谐窄线宽光学参量振荡器
第49卷第11期V ol.49N o.ll红外与激光工程Infrared and Laser Engineering2020年11月Nov. 2020绿光泵浦的黄光波段可调谐窄线宽光学参量振荡器张鹏泉\项铁铭”,史屹君2(1.杭州电子科技大学电子信息学院,浙江杭州310018;2.天津可宏振星科技有限公司,天津300192)摘要:为实现波长可调谐的窄线宽黄光波段激光输出,设计搭建了以倍频声光调QNd:YAG激光器 的532 nm脉冲绿光输出为泵浦源、以II类相位匹配磷酸钛氧钾(KTP)晶体为非线性介质的折叠腔光 学参量振荡器(OPO)。
首先产生腔内振荡的近红外可调谐闲频光,在此基础上基于LBO晶体I类非 临界相位匹配方式对OPO的闲频光进行内腔倍频,得到波长调谐范围587.2〜595.2 nm的黄光波段输 出。
为改善OPO光谱特性,在OPO闲频光谐振腔内插入熔融石英标准具,有效压缩了 OPO输出黄光 的光谱线宽。
绿光泵浦源脉冲重复频率10 kHz、平均功率24.0 W下在波长591.2 nm处获得了最高黄 光输出功率2.89 W,光束质量因子A/2=3.4,从532 nm泵浦光到黄光输出的转换效率为12.0%,脉冲宽 度37 ns,对应峰值功率7.8 kW。
此时黄光光谱半高全宽为0.15 nm,相比未在OPO腔内插入标准具自 由运转状态下的光谱得到明显改善。
关键词:光学参量振荡器;可调谐激光;窄线宽激光中图分类号:TN248.1 文献标志码:A DOI:10.3788/IRLA20200275Green pumped yellow wavelength tunable narrow linewidthoptical parametric oscillatorZhang Pengquan1,Xiang Tieming1*,Shi Yijun2(1. School of Electronics and Information, Hangzhou Dianzi University, Hangzhou 310018, China;2. Tianjin Bright Star Technology Co., LTD, Tianjin 300192, China)Abstract:A pulsed optical parametric oscillator(OPO)was demonstrated for the purpose of wavelength-tunable yellow output with narrow spectral line width.The OPO pumped by the green output of an acousto-optic Q-switched Nd:YAG used a type II phase-matched KTi0P04(KTP)crystal as the nonlinear gain medium and a folded cavity arrangement.The OPO was designed to have the idler wave tunable in near infrared oscillated in the cavity,which was further frequency doubled to generate the wavelength-tunable yellow output by using a LiB305(LBO)crystal with type I non-critical phase matching scheme.A fused silica etalon was inserted in the idler wave cavity to narrow the idler wave and the resultant yellow spectral line width.The wavelength of the yellow output obtained could be tuned over587.2-595.2 nm,within which the maximum average output power of 2.89 W was obtained at 591.2 nm,under an incident average green pump power of24.0 W.The beam quality factor M2was 3.4.The conversion efficiency from the green pump to the yellow output was 12.0%. The pulse width at the maximum output power was37 ns,and the peak power was 7.8 kW.The spectral line width of the yellow output was0.15 nm,which was narrowed effectively compared with that without etalon in the OPO cavity. Key words:optical parametric oscillator;tunable laser;narrow linewidth laser收稿日期:2020-06-15;修订日期:2020-07-11作者简介:张鹏泉(1976-),男,正高级工程师,硕士,主要从事光电信号探测和对抗方向的研究工作。
单片机红外遥控外文翻译
Infrared Remote And Chips Are IntroducedPeople's eyes can see the visible wavelength from long to short according to the arrangement, in order to red, orange, yellow, green, green, blue, violet. One of the red wavelengths for 0.62 ~ 0.76 mount, Purple is 0.38 wavelength range ~ mount. Purple is shorter than the wavelength of light called ultraviolet ray, red wavelengths of light is longer than that of infrared light. Infrared remote control is to use wavelength for 0.76 ~ 1.5 mount between the near infrared to transfer control signal.Commonly used infrared remote control system of general points transmit and receive two parts. The main component part for the launch of infrared light emitting diode. It is actually a special light emitting diode, due to its internal material differs from ordinary light emitting diode, resulting in its ends on certain voltage, it is a rather infrared light. Use of infrared light emitting diode the infrared wavelengths, for 940nm appearance and ordinary, just the same light emitting diode five different colors. Infrared light emitting diode generally have black and blue, transparent three colors. Judgment of infrared light emitting diode and judgment method, using a millimeter to ordinary diode electric block measure of infrared light emitting diode, reverse resistance. The infrared light emitting diode luminescence efficiency to use special instrument to measure precise, and use only spare conditions to pull away from roughly judgment. Receiving part of infrared receiving tube is a photosensitive diode.In actual application of it receiving diode to reverse bias, it can work normally, i.e., the infrared receiving circuit application in diode is used to reverse, higher sensitivity. Infrared receiving diode usually have two round and rectangular. Due to the power of infrared light emitting diode (or lesscommonly 100mW), so ir receiving diode received signals is weak, so will increase high-gain ones.the amplifier circuit.In common CX20106A, etc. PC1373H moon infrared receiving special amplifier circuit. In recent years both amateur or formal products, mostly using infrared receiving head finished. The head of infrared receiving product packages generally has two kinds: one kind USES sheet shielding, A kind of plastic packaging. There are three pin, namely the power is (VDD), power negative (GND) and data output (VO or OUT). Infrared receiving head foot arrangement for types varied, manufacturer's instructions. Finished the advantages of infrared receiving head is not in need of sophisticated debugging and shell screen, use rise as a transistor, very convenient. But when used in the infrared receiving attention finished first carrier frequency.Infrared remote common carrier frequency for 38kHz, this is transmitted by using 455kHz Tao Zhen to decide. At the launch of crystals were integer frequency, frequency coefficients, so commonly 12, so 455kHz ÷12 hundredth kHz 38kHz hundredth 379,000. Some remote control system adopts 36kHz, 56kHz, etc. general 40kHz launched by the crystals of oscillation frequency to decide.Infrared remote characteristic is not influence the surrounding environment and does not interfere with other electric equipment. Due to its cannot penetrate walls, so the room can use common household appliance of remote control without mutual interference, Circuit testing is simple, as long as given circuit connection, generally does not need any commissioning can work, Decoding easily, can undertake multiple remote control. Because each manufacturer produces a great deal of infrared remote application-specific integrated circuit, when need press diagram so jip. Therefore, the infrared remote now in household appliances, indoor close (less than 10 meters) in the remote control is widely used.Multiple infrared remote control system of infrared emission controlbuttons, there are many parts general representative of different control function. When pressed a button, correspondingly in the receiver with different output.Receiving the output state can be roughly divided into pulse, level, self- locking and interlock, data five forms. "The pulse output is according to launch" when the button, the receiver output terminals output corresponding "effective", a pulse width 100ms in general. "Level" refers to the output launch press button, the receiver output corresponding output level ", "effective transmit to loosen the receiver" level "disappears. This "effective pulse" and "effective", may be of high level is low, and may also depend on the output corresponding static state, such as feet for low, static "high" for effective, As for the static, "low" high effective. In most cases, "high" for effective. "Since the lock" refers to launch the output of each time you press the button, a receiver output corresponding change, namely originally a state for high level into a low level, originally for low level into high level. The output power switch and mute as control etc. Sometimes also called the output form for "invert". "The interlock" refers to multiple outputs each output, at the same time only one output. The TV sets of this case is selected, the other is like the light and sound input speed, etc."Data" refers to launch the output some key, use a few output form a binary number, to represent different keystroke.Normally, the receiver except a few data output, but also a "valid" output data, so the timely to collect data. This output form with single-chip microcomputer or are commonly used interface. In addition to the above output form outside, still have a "latch" and "temporary" two forms. The so- called "latch" refers to launch the output signal of each hair, the receiver output corresponding ", "new store until you receive signals. "Temporary" output and the introduction of"level"output is similar.Remote distance (Remote Control effect of RF Remote Control distance)are the major factors as follows:unched in power transmission power: while distance, but great power consumption, easy to generate interference.2.and receiving the receiver sensitivity, receiving, remote distance increased sensitivity to improve, but easy to cause disturbance maloperation or abuse.3.antenna, using linear antenna, and parallel, remote distance, but occupies a large space, in use the antenna spin, pull can increase the remote distance.4.and the higher height: antenna, remote farther, but by objective conditions.5.and stop: current use of wireless remote use of UHF band stipulated by the state, the propagation characteristics of approximate linear transmission, light, small, transmitters and receivers diffraction between such as walls are blocking will greatly discounted remote distance, if is reinforced concrete walls, due to the absorption effect conductor, radio waves.Considering the design of hardware volume small to be embedded in the remote control, so we chose 20 foot single-chip chip AT89C2051. Below is the introduction of the function.(1)AT89C2051 internal structure and performanceAT89C2051 is a byte flash 2K with programmable read-only memory can be erased EEPROM (low voltage, high performance of eight CMOS microcomputer. It adopts ATMEL of high-density non-volatile storage technology manufacturing and industrial standard MCS - 51 instruction set and lead. Through the combination of single chip in general CPL1 and flash memory, is a strong ATMEL AT89C2051 microcomputer, its application in many embedded control provides a highly flexible and low cost solutions. The compatible with 8051 AT89C2051 is CHMOS micro controller, the Flash memory capacity for 2KB. And CHMOS 80C51 process, have two kinds ofleisure and power saving operation mode. The performance is as follows.5.CUP, 2KB Flash memory,Working voltage range 2.7-6V, 128KB data storage.The static working way: 0-24MHz, 15 root input/output line.A programmable serial, 2 a 16-bit timing/counters.There is a slice of inside precision simulation comparator, 5the interrupt sources,2priority.Programmable serial UART channel, Directly LED driver output,The internal structure of AT89C2051 is shown in figure 1.Figure 1 AT89C2051 interior structure(2)AT89C2051 chip pin and functionIn order to adapt to the requirement of intelligent instrument, embedded in the chip foot AT89C2051 simplified configuration, as shown in figure b. The major changes to: (1) the lead foot from 20 to 40 wires, (2) increased a simulated comparator.=Diagram b AT89C2051 foot figure.AT89C2051 pin function:1.the VCC:voltage.1.to GND.1.P1 mouth: P1 mouth is an 8-bit two-way I/O port. P1.2 ~ P1.7 mouth pin the internal resistance provides. P1.0 and P1.1 requirements on the external pull-up resistors. P1.0 and P1.1 also separately as piece inside precision simulation comparator with input (AIN0) and reversed-phase input (AIN1). Output buffer can absorb the P1 mouth 20mA current and can directly LED display driver. When P1 mouth pin into a "1", can make its input. When the pin P1.2 ~ P1.7 as input and external down, they will be for the internal resistance and flow current (IIL). In flash P1 mouth during the procedure and program code data receiving calibration.2.P3: the P3.0 ~ P3.5 P3, P3.7 is the internal resistance with seven two- way I / 0 lead. P3.6 for fixed inputs piece inside the comparator output signal and it as a general I/O foot and inaccessible. P3 mouth buffer can absorb20mA current. When P3 mouth pin into "1", they are the internal resistance can push and input. As input, and the low external P3 mouth pin pull-up resistors and will use current (IIL) outflow. P3 mouth still used to implement the various functions, such as AT89C2051 shown in table P3 mouth still receive some for flash memory programming and calibration of program control signals.5.RST: reset input. RST once, all into high level I/O foot will reset to "1". When the oscillator is running, continuous gives RST pin two machine cycle of high level can finish reset. Each machine cycle to 12 oscillator or clock cycle.6.XTAL1: as the oscillator amplifier input and inverse internal clock generator input.7.XTAL2: as the oscillator reversed-phase the amplifier's output.P3 mouth function as is shown in table 1.Table1P3 mouth pin FunctionP3.0 RXD(Serial input)P3.1 TXD(Serial output port)P3.2 INT0(External interruption0)P3.3 INT1(External interruption1)P3.4 TO(The timer 0 external input)P3.5 T1(The timer1 external input)(3)the software and hardware constraints.AT89C2051Due to the foot of the chip AT89C2051, no set limits of external storage interface,so, for external memory read/write instructions as MOVX etc.Due to 2KB ROM, so, the space to jump instruction should pay attention to the destination address range (transfer 000H - 7FFH), beyond the range ofaddresses, will not meet wrong results. The scope of data storage is 00H (7FH -- when stack manipulation), also should be noticed.The input signal is simulated by the original P3.6 foot into the microcontroller,so the original P3.6 foot.Unable to external use. Simulation comparator can compare two simulation, if the size of the voltage external A D/A converter and its output as A comparator analog input, and by simulating the comparator another input voltage to be measured, through the introduction of the software method can realize the A/D conversion.8.the Flash memory AT89C2051)Provide a 2KB of single-chip AT89C2051 in Flash memory chips, which allows the online program to modify or use special programming programming.(1)Flash memory encryptionAT89C2051 SCM has 2 encryption, can programming (P) or programming (U) to obtain different encryption functionality. Encryption functionality table as shown in table1-1.Encrypt a content erased only through chips to erase operation.(2)Flash memory programming and procedures,the piece inside chip AT89C2051 Flash memory programming.Note:1.the counters RESET at an EPROM inside the rising edge, and 000HRESET to XTAL1by foot is executed, pulse count.2.pieces of 10ms to erase PROG pulse.3.during the programming P3.1 pulled low RDY/BSY instructions. (3).AT89C2051 SCM in Flash memory chips programming steps are as follows:1.in the sequence is the VCC GND pin, add working voltage, XTAL1 pin RESET,receiving GND pin, other than the above time,waiting for10ms.2.In P3.2 pin RESET,heightening level.3.In P3.3, P3.4, P3.5, P3.7 pin;add model multilevel.4.P1.0 P1.7 -- for the 000H unit add data bytes.5.RESET to increase the 12V activation programming.6.P3.2 jump to a one byte programming or encryption.7.calibration has been programming, data from 12V to RESET logic level "H" and set P3.3 P3.7 -- for the correct level, and can output data in P1 mouth.8.For the next addresses) in the unit XTAL1 byte programming, a pulse, make address counter add 1, in mouth add programming data.9.programming and calibration circuit figure c, d.Figure c programming circuit Figure d calibration circuit Explanation:(1)P3.1 during programming instructions to be low RDY/BSY,(2)single erasing the PROG10ms need,(3)internal EEPROM address counter on the rising edge RESET, and 000H RESET to XTAL1 by foot pulses are executed.Along with the rapid development of science and technology, human society has undergone earth-shaking changes. Make our life more colorful. In these changes, the remote control technology has been widely permeates TV, aerospace, military, sports and other production, all aspects of life. From the broad sense, all equipped with electric locomotive facility or electrical switches, if feel some necessary, can consider to improve existing with remote control device, the operation fixed switch to realize the remote operation of theoriginal equipment,stop, the variable,etc. Function.switch, for example, can be used to control the electric control switch the light switch, We design the infrared remote control system to realize the opponent switch quantity control. Infrared remote characteristic is not influence the surrounding environment and does not interfere with other electric equipment. Due to its cannot penetrate walls, so the room can use common household appliance of remote control without mutual interference, Circuit testing is simple, as long as given circuit connection, generally does not need any commissioning can work, Decoding easily, can undertake multiple remote control.红外遥控人的眼睛能看到的可见光按波长从长到短排列,依次为红、橙、黄、绿、青、蓝、紫。
光学专业英语词汇总结
Vocabulary 2
Ultraviolet 紫外的 visible 可见的 infrared 红外的 scalar function 标量函数 vector function 矢量函数 wavelength 波长 frequency 频率 Angular frequency 角频率 Radian 弧度
Vocabulary 9
frequency conversion 频率转换 Down conversion 下转换 Parametric process 参量过程 Nonparametric process 非参量过程 Spontaneous Parametric Down conversion 自发参量下转换 quasi-phase match 准相位匹配 Phase mismatch 相位失配
Hologram 全息图 holography 全息术 holographic reconstruction 全息再现 holographic recording 全息记录 volume holography 体全息术 reference wave 参考波 object wave 物波 coherent light 相干光
Hologram vocabulary
Emulsion
感光乳剂 slit 缝
Orthogonal
正交的 monochromatic 单色的
Exposure
曝光
bragg condition
布拉格(布喇格)条件 conjugate 共扼 rainbow hologram 彩虹全息图
Vibrate 振动 Apparatus 器械,仪器 Minimal 最小的 Fluctuation 波动,起伏 illuminate 照明 Transparency 透明物 Planar 平面的 Three-dimensinal 三维的
LC circuit
The two-element LC circuit described above is the simplest type of inductor-capacitor network (or LC network). It is also referred to as a second order LC circuit to distinguish it from more complicated (higher order) LC networks with more inductors and capacitors. Such LC networks with more than two reactances may have more than one resonant frequency.The order of the network is the order of the rational function describing the network in the complex frequency variable s. Generally, the order is equal to the number of L and C elements in the circuit and in any event cannot exceed this number.Animated diagram showing the operation of a tuned circuit (LC circuit). The capacitor C stores energy in its electric field E and the inductor L stores energy in its magnetic field B (green). This jerky animation shows "snapshots" of the circuit at progressive points in the oscillation. The oscillations are slowed down; in an actual tuned circuit the charge oscillates back and forth tens of thousands to billions of times per second.An LC circuit, oscillating at its natural resonant frequency, can store electrical energy. See the animation at right. A capacitor stores energy in the electric field (E) between its plates, depending on the voltage across it, and an inductor stores energy in its magnetic field (B), depending on the current through it.If an inductor is connected across a charged capacitor, current will start to flow through the inductor, building up a magnetic field around it and reducing the voltage on the capacitor. Eventually all the charge on the capacitor will be gone and the voltage across it will reach zero. However, the current will continue, because inductors resist changes in current. The current will begin to charge the capacitor with a voltage of opposite polarity to its original charge. Due to Faraday's law,the EMF which drives the current is caused by a decrease in the magnetic field, thus the energy required to charge the capacitor is extracted from the magnetic field. When the magnetic field is completely dissipated the current will stop and the charge will again be stored in the capacitor, with the opposite polarity as before. Then the cycle will begin again, with the current flowing in the opposite direction through the inductor.The charge flows back and forth between the plates of the capacitor, through the inductor. The energy oscillates back and forth between the capacitor and the inductor until (if not replenished from an external circuit) internal resistance makes the oscillations die out. In most applications the tuned circuit is part of a larger circuit which applies alternating current to it, driving continuous oscillations. If these are at the natural oscillatory frequency (Natural frequency),resonance will occur. The tuned circuit's action, known mathematically as a harmonic oscillator, is similar to a pendulum swinging back and forth, or water sloshing back and forth in a tank; for this reason the circuit is also called a tank circuit.[1] The natural frequency (that is, the frequency at which it will oscillate when isolated from any other system, as described above) is determined by the capacitance and inductance values. In typical tuned circuits in electronic equipment the oscillations are very fast, thousands to billions of times per second.Resonance occurs when an LC circuit is driven from an external source at an angular frequency at which the inductive and capacitive reactances are equal in magnitude. The frequency at which this equality holds for the particular circuit is called the resonant frequency. The resonant frequency of the LC circuit iswhere L is the inductance in henries, and C is the capacitance in farads. The angular frequency has unitsof radians per second.The equivalent frequency in units of hertz isLC circuits are often used as filters; the L/C ratio is one of the factors that determines their "Q" and so selectivity.For a series resonant circuit with a given resistance, the higher the inductance and the lower the capacitance, the narrower the filter bandwidth. For a parallel resonant circuit the opposite applies. Positive feedback aroundthe tuned circuit ("regeneration") can also increase selectivity (see Q multiplier and Regenerative circuit).Stagger tuning can provide an acceptably wide audio bandwidth, yet good selectivity.History[edit]one-turn coil with a spark gap. When a high voltage from an induction coil was applied to one tuned circuit, creating sparks and thus oscillating currents, sparks were excited in the other tuned circuit only when the circuits were adjusted to resonance. Lodge and some English scientists preferred the term "syntony" for this effect, but the term "resonance" eventually stuck.[2] The first practical use for LC circuits was in the 1890s in spark-gap radio transmitters to allow the receiver and transmitter to be tuned to the same frequency. The first patent for a radio system that allowed tuning was filed by Lodge in 1897, although the first practical systems were invented in 1900 by Italian radio pioneer Guglielmo。
医院术语大全
医院术语大全Neurology神经科Neurosurgery神经外科Obstetrics-Gynecology妇产科Oncology癌症专科Ophthalmology眼科Optometry验光科Orthopedic Surgery骨外科Osteopathy整骨疗科Otolaryngology (ENT)耳鼻喉科Pathology病理科Pediatrics小儿科Plastic surgery整形外科Podiatry足科Psychiatry精神治疗科Physiatry物理康复科Physical Medicine and Rehabilitation物理疗法及恢复正常生活护理Pulmonary Medicine肺科Radiation Oncology癌症放射疗科RadiologyX光科Urology泌尿科Vascular Surgery血管外科Other Health Care Professionals其它医疗专业人员Audiologist听觉学专家Dental Assistant牙医助理Dietitian饮食指导员Genetic Counselor遗传病辅导员Health Technician健康技员Laboratory Technician化验技员Medical Assistant医务助理Medical Technologist医学技师Home Visiting Nurse家访护士Nutritionist营养专家Pharmacist药剂师Pharmacologist药理学专家Physical Therapist物理治疗员Physician's Assistant医生助手Psychologist心理学专家Psychologic Counselor心理辅导员ABSS 自动(磁带)空白部分扫描ABL(automatic bright limiting)自动亮度限制ABL ON OFF 自动黑电平开/关ABL SW ON 自动黑电平开关接通ABO 自动电子束最佳化ABO ADJ 自动电子束最佳化调整ABO VIDEO 自动电子束最佳化视频ABO VIDEO ADJ 自动电子束最佳化视频调整ABO VIDEO IN 自动电子束最佳化视频输入AC (alternating current)交流电AC IN 交流输入AC MOTOR 交流电机AC MOTOR SWAC 交流电机开关AC 自动色(饱和度)控制AC mains input 交流电输入ACC AMP ACC放大ACC AMP (REC) ACC放大录制ACC/APC BURST FLAG 自动色度控制/自动相位控制旗脉冲ACC (automatic chrominance control) 自动色度控制ACC AMP (automatic chrominance control amplifier) 自动色度控制ACC BF PHASE 自动控制旗脉冲相位ACC LEVEL 自动色度控制电平ACC LEVEL SW 自动色度控制电平开关ACC BURST GATE ACC色同步选通门ACC DC AMP ACC直流放大ACC DET 自动消色放大ACTION 作用ADAPTOR适配器ADC(automatic degaussing circuit)自动消磁电路ADD CIRUIT 相加电路ADD RESSING 寻址ADJ (ADJUSTMENT) 调整ADV (一桢一桢)步进AERIAL 天线AFC (automatic frequency control) 自动频率控制AFC BALANCN 自动频率控制平衡调节AFC CENTER AFC中心AFC DC 自动频率控制(AFC)直流AFC DC BIAS AFC直流偏置AFC (DC) OUT 自动频率控制(DC)输出AFC DRIVE 自动频率控制推动AFC ERROR 自动频率控制误差信号AFC ERROR BUFFER AFC误差缓冲AFC FH TUNINGAFC行频调谐AFC FH TUNING AMP AFC行频调谐放大AFC GAIN AFC增益AFC GATE 自动频率控制门AFC IN 自动频率控制输入AFC OUT AFC输出AFC PULSE AMP 自动频率控制脉冲放大AFC SET 自动频率控制设定AFC VCO AFC压控振荡器AFC VCO FREQ AFC压控振荡器频率AFPC (automatic frequency phase control) 自动频率相位控制AFS(automatic frequency stabilization)自动频率稳定AFTER CLOCK 时钟后AFTER CLOCK PULSE 时钟脉冲之后AGC 自动增益控制AGC AMP AGC放大AGC DETECTOR 自动增益控制检测AGC ERROR BUFFER 自动增益控制误差缓冲器AGC PROT AGC保护AH(AUDIO/CTLHEAD) AH(音频控制磁头)ALARM TONE BURS T 告警音频缓冲ALT 行交替ALT PULSE 行交替脉冲ALTERNA TEDSC 交替的副载波ALU 运算器AMP(amplifier)放大器AMPLIFIER DETECTOR 放大器/检波器AMPLITUED LIMIER 限幅器ANALOG SWITCH 模拟开关ANODE 阳极ANC 自动消噪电路ANTENNA 天线APC(automatic phase control)自动相位控制APC BF INV APC 旗脉冲倒相APC 自动相位控制(检波)ARC(automatic resolution control)自动清晰度控制AT(Ampere turns)安(培)匝数ATT (A TTENUA TOR) 衰减器AUTOMA TIC地自动B(blue)蓝色B(brightness)亮度BA(buffer amplifier)缓冲放大器BALANCE平衡BALUN 平衡-不平衡转换器BRIGHT 亮度BRIGHTNESS 亮度调节BLLE OUT OFF蓝枪截止调节BLUE OUT 蓝色输出BURST 色同步信号BURST GATE 色同步选通电路BURST PHASE 色同步信号相位CURRENT LIMITTER 电流限制器CEN 中心CHROMA 色度CHROMA AMP 色度放大器CHROMA BURST AMP 色度、色同步信号放大器CHROMA BOARD 色通道板CHROMA FIL TER 色度滤波器CHROMINANCE 色度通道CLAMPER 钳位器CMOS (complementary metal -oxide-semiconductor) 互补型金属-氧化物半导体COLOUR CONT(color controller) 彩色控制器COLOUR DIFFERENCE 色差COLOUR SYNC 彩色同步调节COLORKILLER 消色器COLORTONE 色调CONT 对比度、控制CONTRAST 对比度CONTROL 控制CONSOLE-控制柜CPT (color picture tube) 彩色显像管CPT BOARD 彩色显像管座板CRT (cathode - ray tube) 阴极射线管(显像管)CRT DRIVE BOARD 显像管激励电路板DC (direct current)直流电DAMPER 阻尼器DGC (degaussing coil) 消磁线圈DL (delay line ) 延时线DRIVE 激励、推动DRIVE TRANSF 推动变压器DY (deflection yoke)偏转线圈EHT (extra -high tension) 极高压EMERGENCY-急停装置ERROR AMP (error amplifier) 误差电压放大器E-W CORRECTION(east - west correction) 东西向校正FBT (fly back transformer) 逆程变压器FILTER 滤波器FLIP FLOP 双稳态触发器FIYEACK BLANKING 回扫消隐FOCUS 焦点FOCUS VR (focus variable rheostat) 聚焦电位器f.(fuse) (fuse) 保险丝GANTRY-机架G (green) 绿色的GND (ground) 接地GREEN CUT OFF 绿枪截止调节GREEN OUT 绿色输出GREY 灰度G - Y MATRIX (G - Y )矩阵H. BLK (horizontal blanking) 行消隐H.DY (horizontal deflection yoke) 行偏转线圈HFC (high frequency choke) 高频扼流圈H.HOLD (horizontal hold) 行同步调节H (L).DRIVE (horizontal driver) 行推动放大器HLIN (horizontal linearity) 行线性H(L)OUT BOARD 行输出板H .M(module)厚膜电路HOR AFC (horizontal automatic frequency control) 行自动频率控制HOR DRIVE TRANS 行激励变压器HORIZONTAL 行(水平)扫描部分HORIZ O/P (horizontal out put ) 行脉冲输出H.OSC(horizontal oscillator) 行振荡器H.PHASE (horizontal phase) 行(同步)相位调节H.SIZE 水平幅度调节器HV (high voltage)高压IC (integrated circuit)集成电路INPUT 输入KC (kilohertz) 千周KHz(kilohertz) 千赫KILLER AMP (killer amplifier)消色放大器LINE-FILTER-滤波器LOW 低的LPF(low-pass filter)低通滤波器MAIN BOARD主电路板MC (megacycles per second)兆赫MF (ceramic filter) 陶瓷滤波器MFD;mfd(microfarad) 微法ms (millisecond) 毫秒mV (mill volt) 毫伏ON/OFF 开/关operating point 工作点OSC (oscillator) 振荡器OUT 输出端OVERLOAD -FUSE 过载保险丝OVERLOAD -PROTECTION 过载保护OVER-VOLTGE PROTECTION 过压保护POSITIVE THERMISTOR 正温度系数热敏电阻POWER BOARD 电源板POWER CORD 电源线POWER DRIVE 功率激励POWER RECT(power rectifier) 电源整流器POWER REG(power regulation) 功率调整POWER REG RM (power regulation reluctance) 功率调整管散热片POWER SUPPLY 电源POWER TRANS (power transformer) 电源变压器PROTECTOR-保护装置PEDESTAL CLAMP 消隐脉冲钳位PEDESTAL CLAMPER 消隐脉冲钳位电路PF (Pico farad) 微微法PHASE CONT (phase controller)相位控制器R (red) 红色的R.GBL( red background) 红色背景(暗平衡)调节RECT (rectifier) 整流器R.DRIVE (red drive) 红色驱动(白平衡)调节RED CUT OFF 红色截止调节RED OUT 红色输出REFERENCE VOLTAGE 基准电压REGULA TOR 稳压器SIDE-PINAMP 左校正放大器STACK-硅堆SUB BRIGHT 副亮度调节SYNC (synchonization) 同步(信号)SAW (surface acoustic wave) 声表面滤波器SCREEN 帘栅极(加速器)电压调节SYNC (synchronous separator)同步信号分离器TF (temperature fuse) 温度保险丝THERMO-SENSOR 温度探头TO CPT BOARD 接到显像管印刷板VOUT SWITCH 垂直泵浦开关V.BLK (vertical blanking) 场消隐V.DY (vertical deflection yoke) 场消隐线圈VERT CENT (vertical center) 场中心调节VERT DRIVE AMP 场推动放大器VERT OSC (vertical oscillator) 场振荡器VERT SIZE (vertical size) 垂直幅度调节VERT TRIGG (vertical trigger) 场触发V HOLD 场同步调节VIDEO 视频放大VIDEO& CHROMA BOARD 视频与色度印制板VOLTAGE DIVIDER 分压器X (crystal) 石英晶体谐振器Y AMP 亮度放大器yoke 偏转线圈;P Urine Analyzer 尿液分析仪blood sugar(glucose )analyzer血糖分析仪test strip 测试条reagent 试剂Semi-automatic Biochemical Analyzer半自动生化分析仪Automatic Blood Cell Analyzer全自动血细胞分析仪Urine sediments analyzer尿沉渣Bio-safety Cabinet 生物安全柜Incubator培养箱High Frequency Electrotome 高频电刀shadowless lamp无影灯High speed refrigerated centrifuge高速冷冻离心机hot air sterilizer热空气消毒箱microbiological incubator微生物培养箱Halogen light 卤素灯disposable sterile injector 一次性无菌注射针injection set注射器disposable venous infusion needle一次性静脉输液针disposable infusion set 一次性使用输液器blood transfusion set输血器infusion bag液袋urine drainage bag集尿袋blood bag血袋medical catheter医用导管stainless steel needle不锈钢医用针管blood taking needle采血针needle destroyer针头销毁器automatic packer自动纸塑包装机scalp vein set头皮针uniprocessor version单机版network version网络版macromolecule-solvent 高分子溶解的macromolecule cold accumulation 高分子蓄冷cold treatment冷疗法ice pack冰袋eyeshade 眼罩Medical injection pump医用灌注泵lithotrite 碎石机extracorporeal shock wave lithotrite体外冲击碎石机Ballistic intracroporeal lithotrite 气压冲击体内碎石机Laparoscope 腹腔镜Urology 泌尿外科kidney stones 肾结石Multi-parameter monitor, 多参数监护仪maternal monitor/fetal monitor母亲/胎儿监护仪ICU monitor 重症监护仪anesthetic equipment 麻醉机respirator呼吸机electronic colposcope 电子阴道镜smog absorber烟雾吸收器digital film room 数字胶片室Permanent Magnet Open Magnetic Resonance system 永磁开放式磁共振系统Ultrasonic Color Doppler Diagnostic system彩色超声多普勒诊断系统Mobile CT system 移动CT系统X-ray Mammary Machine 乳腺X线机Mammography乳腺high precision Stereotaxic 高精度脑立体定向仪portable Type-B ultrasonic 便携式B超Sterilization and Disinfection Equipment消毒灭菌设备Radiotherapeutic equipment.放射疗法设备pharmaceutical equipments.制药设备horizontal pressurized steam sterilizer普通卧式压力蒸汽灭菌器medical electronic linear accelerator医用电子直线加速器high frequency X-rays diagnostic machine高频X射线诊断机simulated positioner模拟定位机high frequency mobile X-rays machine高频移动X射线机内科系统Medicine Systems外科系统Surgery Systems医技科室Medical Laboratory血液病科Hematology Department普外(肝胆)General Surgery临床检验Clinical Laboratory输血科Blood Bank内分泌科Endocrinology Department胸外科Thoracic surgery病理科Pathology Deparment脑电图室ECG Laboratory消化内科Digestive System Department心外科Cardial Surgery传统放射科Traditional Radiology Department肺功能室Lung Function Laboratory心血管内科V asculocardiology Deparment泌尿外科Urology SurgeryMR室MR Laboratory胃镜室Dndoscope Laboratory神经内科Neurology Department肿瘤外科Oncological SurgerySCT室SCt Laboratory人工肾室Hemodialyses Room介入科Invasive Technology Department神经外科Neurological Surgery超声诊断科UItrasonic Diagnosis DeparmentDSA室DSA Room呼吸科Pneumology Department骨科Orthopedics Department超声多谱勒室UItrasonic Doppler Laboratory血液净化室Laminar Airflow (LAF) Room肾内科Urology Department小儿外科Pediatric Surgery核医学科Isotopic Laboratory高压氧仓室Hyperbaric Chamber小儿科Pediatrics Department整形科Plastic SurgeryECT 室ECT Laboratory院内感染监控室Nosocomial Infection Monitory中医科Traditional Chinese Medicine Department 烧伤科Department of Burn供应室Supply House血液成份分离室Cytopheresis Laboratory高干病房Senior Officials inpatient Ward妇产科Obstetric and Gynecologic Department营养室Nutrition House体外反搏室Counter Extropulsative Room华侨病房Overseas Chinese Ward口腔科Stomatological Department康复科Rehabilitation Department保健科Medical Care Department for personnel眼科Ophthalmologic Department针灸科Acupuncture and Moxibustion De-parment 耳鼻喉科Otorhinolaryngologic Department理疗科Physiotherapy Deparment痔疮科Hemorrhoids Deparment按摩科Massage Department皮肤科Dermatology Department麻醉科Anesthesia Department省级重点学科Key Subjects at the Provincial Level血液病、内分泌疾病、肝胆外科、胸心外科Hematology, Endocrinology, Genneral Surgery and Cardio-Thoracic Surgery 省级医疗领先特色专业The Leading Subjects of Medicine at the Provincial Level心内科、烧伤科、儿科心理学Cardiology, Department of B urn , Pediatric Psychology医院特色专科Characteristic Professional Subjects of Union Hospital消化内科、普外、肿瘤、泌尿、神经、整形、耳鼻喉科、介入、影像Digestive System Diseases, GeneralSurgery, Oncology, Urology, Neurology, Plastic Surgery,Otorhinolary, Invasive Department and Medical Imagery。
振荡器osc
H(jω) ≈ H(jω0) + ∆ωd-d---Hω--- ω = ω0
(12)
根据 Barkhausen 条件在振荡频率点 ω0 处
H(jω0) = 1,
∆ω
d----H--dω
ω = ω0
«1
因此振荡频率附近的噪声功率的传输函数
T(ω0 + ∆ω) 2 ≈
------1------- 2 ∆ω d-d---Hω---
(8)
因此,相位噪声谱被调制或搬移到了载波 ωc 处。
jitter phase noise
ωc
11 of 20
射频集成电路设计基础 > 振荡器 > 振荡器中的干扰和相位噪声
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• 振荡器的 Q 值与相位稳定性
若正反馈系统的开环传递函数为 H(jω),并且其选 频网络为 LC 谐振电路 (LC tank),其 Q 值为
gmc = ω02C1C2rs
(1)
为了保证起振,必须满足 R > rs ,即
gm > gmc = ω02C1C2
(2)
起振时,流经管子的平均电流可能不等于偏置电流,其中的一部分对 C1, C2 充 放电,导致管子栅极偏压 VB 发生偏移, gm 逐渐靠近 gmc。在很多情况下稳定工 作时的 VB 将低于 Vth,即管子工作在 Class-C 状态。在这种状态下,管子的平均 电流等于偏置电流,而其中的交流分量则通过 LC 网络后在栅极产生正弦信号。
Qiuting Huang 对共漏极 ( 跟随器结构 ) CMOS Colpitts 振荡器的工作原理给出了很 好的分析 [6]。首先,在小信号条件下,振 荡器的等效电路为
Waves CODEX 用户指南说明书
WAVES CODEX USER GUIDETABLE OF CONTENTSCHA PTER 1 – INTRODUCTION (3)1.1 Welcome (3)1.2 Product Overview (3)1.3 Concepts and Terminology (3)1.4 Components (4)CHA PTER 2 – QUICK STA RT GUIDE (5)CHA PTER 3 – INTERFA CE A ND CONTROLS (6)3.1 Interface (6)3.2 Controls (7)CHA PTER 4 – STA NDA LONE A PPLICA TION (21)CHA PTER 5 – THE WAVESYSTEM (23)CHAPTER 1 – INTRODUCTION1.1 WelcomeThank you for choosing Waves! In order to get the most out of your Waves processor, please take the time to read through this manual.In conjunction, we also suggest that you become familiar with . There you will find an extensive A nswer Base, the latest Tech Specs, detailed Installation guides, new Software Updates, and current information on Licensing and Registration.By signing up at , you will receive personalized information on your registered products, reminders when updates are available, and information on your licensing status.1.2 Product OverviewWaves CODEX is an advanced polyphonic granular wavetable synthesizer designed to provide complex evolving sounds that can range from acoustic and analog emulations, to completely new, previously unheard sounds. CODEX’s controls are highly inviting, combining tradition with intuitive interaction in a way that lets users easily customize existing presets and build custom patches from the ground up. The unique wavetable oscillators are embedded in a subtractive synthesis engine that allows further shaping of the sound using traditional analog subtractive components and layout.1.3 Concepts and TerminologyCODEX is powered by Virtual Voltage™ technology, which connects its various generators and transformation filters, envelopes and modulators. CODEX therefore uses many of the same terms used by its hardware forerunners: VCF (Voltage-Controlled Filter), VCA (Voltage-Controlled Amplifier), and so on.While anyone acquainted with synthesis should feel right at home with the subtractivesection of CODEX, the advanced oscillators might at first seem daunting. Please browse through the oscillator section of this manual to become better acquainted with the unique features and vast possibilities of CODEX’s oscillators. The learning curve will pay off when encountering other synthesizers, whether software or hardware, analog or digital.All users can partake in the instant gratification of CODEX’s vast selection of factory presets. CODEX’s team of experienced preset developers created hundreds of presets, sorted by category, so you can quickly find the exact sound you need—leads, pads, basses, sound effects, sequences, gated rhythmic lines, motions, and more. Or just browse around until something catches your ear and captures your imagination.Whether in the studio or live on stage, many musicians like to control their synth parameters in real time for enhanced creativity and expression. While CODEX supports the basic automation features of plugin hosting technologies such as VST, it also supports MIDI Learn. Assigning a CODEX control to a knob on your MIDI controller is as easy as right-click > Learn > knob turn, done!Granular wavetable synthesis, analog modeling, and supreme sound quality do come at a price. CODEX can be rather CPU- and RAM-hungry compared to other software synthesizers. During the days of analog hardware, less expensive oscillators were often considered “dirty,” while accurate oscillators were costlier than was practical for most musicians. Ironically, in today’s digital world, creating a pristine oscillator is relatively easy; it’s the “dirtiness” that takes more CPU calculations to recreate. To conserve CPU power, CODEX lets you select the maximal number of simultaneous voices.1.4 ComponentsCODEX has one component: CODEX Stereo.CODEX is a virtual instrument plugin and will appear under the related selection menus for virtual instruments under all supported DAW host applications.Waves CODEX also works as a standalone application, using ASIO (Windows) or Core Audio (Mac) drivers to play through your audio device of choice. CODEX receives MIDI data to trigger notes and control changes.CHAPTER 2 – QUICK START GUIDEOpen CODEX on an instrument track in your DAW of choice, or launch the CODEX standalone application.1. Select a preset from CODEX’s factory presets.2. Play!Use the next/previous preset arrow controls on the toolbar to scroll through presets. If you’re looking for a certain type of sound, click the load button to reveal the factory presets, sorted by category.CHAPTER 3 – INTERFACE AND CONTROLS 3.1 Interface3.2 ControlsThe CODEX interface is arranged into four sections grouped according to function:•Voice generation and subtractive elements (blue)•Modulation sources and patches (purple)•Effects, EQ, global and output sections (green)•Arpeggiator/sequencer (red)3.2.1 OscillatorsCODEX has two wavetable oscillators which are labeled1 and 2.OCTA VE determines the pitch range.Range: 32, 16, 8, 4, 2 (from lowest to highest pitch)TUNE determines the pitch.Range: -12 to +12 semitonesFINE fine-tunes the pitch.Range: -100 to +100 centsRESOLUTION determines the resolution of the oscillator.Range:0 to 100TA BLE selects one of 64 internal or imported wavetables.Range: 1 to 64WA VE selects a specific section of the wavetable to be used as the oscillator.Range: 0 to 10FORMA NT controls the formant of the oscillator by altering the spectral peaks of the sound spectrum.Range: -100 to 100STA RT determines the start point of the wavetable scan.Range: 1 to 64MID determines the midpoint of the wavetable scan.Range: 1 to 64END determines the endpoint of the wavetable scan.Range: 1 to 64SPEED determines the wavetable scanning speed.Range: 0 to 100 (free-running), 4/1 to 1/32 (synched)SCA N SYNC determines whether the wavetable scanning is synchronized to the host clock.LOOP – When enabled, wavetable scanning loops continuously between ‘MID’ and‘END’ points.IMPORT allows the integration of user-selectable WAV files. Imported WAV files are automatically converted into wavetables by CODEX. These new wavetables are saved along with presets. CODEX supports WAV files of any length, bit-depth and sampling frequency. For optimal results, samples of between one and five seconds are recommended.FC enables CODEX’s advanced formant correction algorithms.When enabled, formant can be controlled using the ‘FRMNT’ knob.OSC SYNC synchronizes the triggering of the OSC 2 waveform to the rate of OSC 1. When activated, OSC 2 pitch controls affect only the timbre of OSC 2, not its pitch.Range: On/OffFM (Frequency Modulation) controls the amount by which the frequency of OSC 2 is modulated by OSC 1. (OSC 2 only)Range: 0 to 1003.2.2 Additional Oscillators & Mix SectionThis section is used to combine OSC 1 and OSC 2, and to add noise, sub-oscillation and ring modulation.SUB mixes in a triangle wave one octave below OSC 1.Range: 0 to 100NOISE mixes in white noise.Range: 0 to 100RING controls the ring modulation of OSC 1 and OSC 2.Range: 0 to 100OSC 1 / OSC 2 activate each oscillator.Range: In/OutMIX balances the mix between OSC 1 and OSC 2.Range: -50 to +50MONO toggles between monophonic and polyphonic modes.Range: On (mono) / Off (polyphonic)RTRG controls envelope re-triggering. When activated, every note restarts the envelopes. (Mono mode only)Range: On/OffUNISON activates a doubling effect which creates a richer sound.Range: On/OffPORT determines the glide time (portamento) between notes.Range: 0.5 to 2500 ms (0 to 2.5 seconds)A LWA YS/LEGA TO determines whether glide will occur always, or only when the previous note is still held.Range: Legato, Always3.2.3 Voltage Controlled Filter (VCF)The VCF section includes selectable filter slope, frequency and resonance parameters as well as a filter envelope which determines the filter movement on each trigger.TYPE determines the filter type.Range: High Pass, Low Pass, Band Pass, Band RejectSLOPE toggles between two types of pole filters.Range: 2-pole/12 dB per octave, 4-pole/24 dB per octaveCUTOFF controls the VCF cutoff frequency.Range: 0 to 100 (20 Hz to 20 kHz)RES controls the amount of filter resonance.Range: 0 to 100ENV determines the envelope’s cutoff modulation depth.Range: -100 to 100KBD controls keyboard tracking using C3 as its reference point.Range: 0 to 100FM controls the amount of frequency modulation on filter cutoff by OSC 1.Range: 0 – 100A DSR determines the filter’s envelope cutoff behavior after a note is triggered:A(Attack): 1 to 10,000 (1 millisecond to 10 seconds)D (Decay): 1 to 10,000 (1 millisecond to 10 seconds)S (Sustain): 0% to 100% amplitudeR (Release): 1 to 10,000 (1 millisecond to 10 seconds)At 0, time constants are linear; at positive values, the envelope slopes become more concave (below, in black), for a punchier response. Negative values result in a more convex slope shape, for smoother response (below, in red.)VEL sets the VCF cutoff in relation to the Note On velocity.Range: 0 to 100SHA PE determines the contour of the envelope time constants.Range: -50 to 503.2.4 Voltage Controlled Amplifier (VCA)The VCA envelope controls the note level from trigger to release:A(Attack): 1 to 10,000 (1 millisecond to 10 seconds)D (Decay): 1 to 10,000 (1 millisecond to 10 seconds)S (Sustain): 0% to 100% amplitudeR (Release): 1 to 10,000 (1 millisecond to 10 seconds)VEL sets the envelope depth in relation to the Note On velocity.Range: 0 to 100SHA PE determines the contour of the envelope time constants.Range: -50 to 50PUNCH controls the dynamic transient enhancer which makes for a “snappier’ attack.Range: On/Off3.2.5 Low-Frequency Oscillators (LFOS)CODEX features four LFOs: two free and two synchronized. The free LFOs have a continuous cycle time control, while the synched LFOs are voice-triggered and use musical note values (based on the host BPM) to determine the oscillation rate.TYPE controls the LFO waveform shape.Range: Sine, Saw Down, Saw Up, Triangle, Pulse, S&H (random)RA TE controls the frequency of the free LFO.Range: 0.1 to 100 HzTIME controls the rate of the synched LFO, locked to the project’s BPM.Range: 4/1, 3/1, 2/1, 1/1, 1/2, 3/8, 1/3, 1/4, 3/16, 1/6, 1/8, 3/32, 1/12, 1/16, 3/64, 1/24, 1/32LED pulsates at the same rate as the LFO.3.2.6 Envelope 3In addition to the filter and the VCA envelopes, CODEX includes an envelope generator that can be freely assigned to selectable destinations via the modulation matrix.A(Attack): 1 to 10,000 (1 millisecond to 10 seconds)D (Decay): 1 to 10,000 (1 millisecond to 10 seconds)S (Sustain): 0% to 100% amplitudeR (Release): 1 to 10,000 (1 millisecond to 10 seconds)VEL sets the envelope depth in relation to the Note On velocity.Range: 0 to 100SHA PE determines the contour of the envelope time constants.Range: -50 to 503.2.7 Modulation MatrixThe Modulation Matrix allows the patching of modulation sources to selectable destinations.PHA SE inverts the phase, per assignment. Depending on the phase setting, this will change the direction of the modulation.Range: On/OffSRC determines the modulation source.Range: LFO1, LFO2, LFO3, LFO4, Env3, Modwheel, Keyboard, Velocity,Aftertouch, Bender, VCF Envelope, SequencerDEST determines the destination of the modulation source.Range: Global Pitch, OSC 1 Freq, OSC 1 Resolution, OSC 1 Formant, OSC 1Table, Scan 1 Start, Scan 1 Mid, Scan 1 End, Scan 1 Speed, OSC 2 Freq, OSC2 Resolution, OSC 2 Formant, OSC 2 Table, Scan 2 Start, Scan 2 Mid, Scan 2End, Scan 2 Speed, OSC 2 FM, Sub, Noise, Ring, OSC Mix, VCF Cutoff, VCFRes, VCF FM, VCF Env, VCF A, VCF D, VCF R, VCF Crv, VCA, VCA A, VCA D, VCA R, VCA Crv, Pan, HP, LP, Dist, Crusher, Chorus, Delay Mix, Delay Rate,Reverb, Porta, LFO 1 Rate, LFO 2 Rate, LFO 3 Rate, LFO 4 Rate, Env3 A, Env3 D, Env3 S, Env3 R, Env3 Crv, Arp/Seq Rate, Arp\Seq Gate, Mod 1, Mod 2, Mod 3, Mod 4, Mod 5, Mod 6MOD sets the degree to which the modulation source affects the destination.Range: 0% to 100%3.2.8 Arpeggiator/Sequencer (ARP/SEQ)The ARP/SEQ section functions both as a traditional arpeggiator and as a 16-step sequencer. Each sequencer step has an In/Out toggle as well as a pitch control which may be set +/-24 semitones from the currently held note.MODE determines the operational mode.Range: Off, (Arp) Up, (Arp) Down, (Arp) Up/Down, (Arp) Random, SequenceOCT determines the range, in octaves, of the arpeggiator.Range: 1, 2, 3, 4RA TE sets the arpeggiator rate. The drop-down menu activates the host BPM sync function and sets the rate using note values. When the rate is set to Free, the host BPM sync is off and the rate is set manually using the knob on the right.Range: Free (host BPM sync off; range 1 to 50 Hz),1/2, 3/8, 1/3, 1/4, 3/16, 1/6, 1/8, 3/32, 1/12, 1/16, 3/64, 1/24, 1/32GA TE determines the length of each sequencer step as a percentage of its note length.Range: 5% to 100%STEPS determines the number of steps in the current sequence.Range: 1 to 16HOLD enables latching of arpeggiator and sequencer notes after the keys are released.RTRG enables re-triggering of arpeggiator and sequencer notes when new notes are played.SWING pushes even-numbered notes/steps toward the next odd-numbered note/step, to create a shuffle/swing feel.Range: 0 to 1003.2.9 Effects (FX)DIST controls the amount of distortion effect. Distortion is applied per voice, eliminating IMD (inter-modulation distortion.) Settings below 50% create a warm, saturated drive effect; settings above 50% result in a more aggressive, “crunchier” sound.Range: 0% to 100%PRE VCF is a toggle control which determines the placement of the distortion effect in the signal path, either pre-VCF or post-VCF.Range: Pre/PostCRSHR is a distortion-like effect which simulates a reduction in the sample-rate and resolution of the sound.Range: 0 to 100DELA Y is a stereo delay which allows separate delay times for left and right channels, using note values.▪LEFT sets the delay time for the left channel.Range: 1/2, 3/8, 1/3, 1/4, 3/16, 1/6, 1/8, 3/32, 1/12, 1/16, 3/64, 1/24, 1/32▪RIGHT sets the delay time for the right channel.Range: 1/2, 3/8, 1/3, 1/4, 3/16, 1/6, 1/8, 3/32, 1/12, 1/16, 3/64, 1/24, 1/32▪MIX determines the amount of delay effect in the mix.Range: 0 to 100▪FEEDBA CK determines the amount of gain fed back to the delay input.Range: 0 to 100REVERB controls both the amount and size of the virtual plate reverb sound. In addition to increasing the amount of reverb, higher values also increase the reverb size and time.Range: 0 to 100CHORUS determines the amount of chorus modulation.Range: 0 to 1003.2.10 EQThe EQ section is a 4-band fixed-frequency graphic equalizer with high-pass and low-pass filters. At high boost levels, the EQ saturates with different tonal qualities than the Distortion module.HiPA SS controls the high-pass filter frequency.Range: 20 to 20,000 HzLoPA SS controls the low-pass filter frequency.Range: 20 to 20,000 Hz100 controls equalization at 100 Hz.Range: -30 to +30 dB600 controls equalization at 600 Hz.Range: -30 to +30 dB1500 controls equalization at 1500 Hz.Range: -30 to +30 dB9000 controls equalization at 9000 Hz.Range: -30 to +30 dB3.2.11 Global and Out SectionsTEMPO displays the current tempo.Range: 1 to 300SOURCE determines the clock source.Range: Host, InternalBEND RA NGE determines the range of the pitch bender.Range: 0 to 12VOICES determines the number of voices that can be played simultaneously.Range: 1 to 32GA IN sets CODEX’s overall output volume, after all generators and processors.Range: -80 to 0 dBFSMETER displays CODEX’s overall output energy.CHAPTER 4 – STANDALONE APPLICATIONThe CODEX application can be used as a standalone instrument. It requires ASIO drivers for Windows or Core Audio for macOS. Codex.exe (Win) or codex.app (Mac) loads the CODEX instrument and configuration preferences dialogs. Set up the standalone application from its File menu:• All Notes Off Sends an All-Notes-Off MIDI command to CODEX. This isuseful in cases of “stuck” sustaining notes.• Preferences Displays the Preferences dialog for the Audio, MIDI, and User Choices configurations.P REFERENCESOutput displays the audio devices available on the system.Test plays a sound if the outputs are configured correctly.Active Output Channels allows selection of audio outputs from the selected device. Sample Rate displays and sets the sample rate.*Audio Buffer Size displays and sets the buffer size, which influences latency.* *In Windows, sample rate and buffer size cannot bechanged from this panel. To modify these settings: closethe application, adjust sample rate and buffer size withWindows Onlyyour driver's control panel (link shown below), and thenrelaunch.Active MIDI Inputs displays a list of available MIDI input devices on the current system. Select the MIDI device for receiving MIDI data.Tempo: Sets the tempo for all relevant plugins. By default, tempo-based Waves plugins are in a “tempo listen” state and will fix to this value.CHAPTER 5 – THE WAVESYSTEMUse the bar at the top of the plugin to save and load presets, compare settings, undo and redo steps, and resize the plugin. To learn more, click the icon at the upper-right corner of the window and open the WaveSystem Guide.。
NUVOTON NuMicro Nano100系列微控制器数据手册说明书
ApplicationsNuvoton NuMicro TMFamilyNuMicro TMNano100 Base SeriesContactus:*******************Ultra-Low Power ARM ® Cortex ®-M0 MCU forWearable Device Portable Medical DeviceMobile Payment Smart Card Reader Wireless Audio, Motion Gaming IPTV Remote Control Smart Home Appliances Alarm and Security Monitoring Zigbee Smart Energy AMR Portable GPS Data Logger Electronic Shelf Label Electronic Toll Collection IoT DeviceNano100Features of NuMicro TM Nano100 Base Series◆ Core– ARM® Cortex®-M0 core running up to 42 MHz– 24-bit system tick timer– Single-cycle 32-bit hardware multiplier– NVIC for 32 interrupt inputs, each with four levels of priority– Support Serial Wire Debug (SWD) interface and two watch points/fourbreakpoints◆ Ultra-low Power– Single power supply: 1.8V ~ 3.6V– Normal mode: 200 uA/MHz at 12 MHz– Idle mode: 75 uA/MHz at 12 MHz– Power-down mode:2.5 uA (RTC on, RAM retention)1 uA (RTC off, RAM retention)– Wake-up time: less than 3.5 us◆Memory– 32/64/123 Kbytes Flash memory for program memory (APROM)– 4 Kbytes Flash memory for loader memory (LDROM)– 512 bytes page erase for flash memory– 8/16 Kbytes embedded SRAM– Configurable Data Flash (shared with APROM)◆ Clock Control– 12 MHz internal RC oscillator±2% deviation at - 40°C ~ +85°C, 1.8V ~ 3.6V±0.25% deviation at- 40°C ~ +85°C, 1.8V ~ 3.6V by 32.768 kHz oscillator auto calibration– 10 kHz internal RC oscillator for Watchdog timer and Wake-up– 4 ~ 24 MHz external crystal oscillator input for precise timing operation– 32.768 kHz external crystal oscillator input for RTC function and low power system operation– On-chip PLL, up to 120 MHz for high performance system operation◆ Timers– Four sets of 32-bit timers with 24-bit up-timer and one 8-bit pre-scale counter – Counter auto reload– Watchdog timer with 8-bit selectable time-out period– Supports event counter and pulse width capture mode◆ Peripheral DMA– 8 channels PDMA for peripheral timer, UARTs, SPIs, I²C, ADC, DAC, VDMA, CRC ◆ CRC– Supports CRC-CCITT, CRC-8, CRC-16, and CRC-32◆ RTC– Supports software compensation– RTC counter, calendar counter and alarm– 80 bytes backup register with snoop pin detection◆ PWM/Capture– 8 channels 16-bit PWM and 16-bit digital capture timers– Dead-zone generator for complementary paired PWM ◆ ADC/DAC– 12 channels 12-bit SAR ADC up to 2 MSPS– 2 channels 12-bit DAC up to 400 KSPS– 1.8V/2.5V internal voltage reference– On-chip temperature sensor◆Communication Interfaces– Five UARTs, (two UARTs up to 1 Mbit/s with flow control)– Three SPIs, up to 32 MHz (Master), 16 MHz (Salve)– Two I²C, up to 1 Mbit/s– Three ISO7816-3 (Smart card interface) with UARTs function– RS485, LIN, IrDA (SIR) function◆I2S– Interface with external audio CODEC– Master and Slave mode– Capable of handling 8, 16, 24 and 32-bit word sizes– Mono and stereo audio data◆ Wake-up Sources– RTC, WDT, I²C, Timer, UARTs, SPIs, BOD, GPIOs◆EBI Bus– Accessible space: 64 Kbytes in 8-bit mode or 128 Kbytes in 16-bit mode– 8/16-bit data width◆ Brown-out Detector– Three levels: 1.7V/2.0V/2.5V– Brown-out interrupt and reset option◆GPIOs– Up to 86 general-purpose GPIO pins– Three I/O modes: Push-Pull output, Open-Drain output, Input only with high impendence– All inputs with Schmitt trigger– All I/O pins can be configured as interrupt source with edge/level setting– Input 5V toleranceBuilt-in LDO for Wide Operating Voltage Range– 1.8V to 3.6VOperating Temperature– - 40°C ~ +85°CReliability– ESD HBM pass 8kV, EFT > ± 4kVCode Security and Series number– 96-bit unique ID– 128-bit unique customer ID◆Packages (RoHS)– QFN48 (7x7mm)– LQFP48 (7x7mm)– LQFP64 (7x7mm)– LQFP128 (14x14mm)。
相噪指标
Phase Noise.This is perhaps the most important parameter in many oscillators and it deserves an in-depth discussion on what it is, how it affects a system and how it can be minimised in an oscillator design.1 Phase Noise An oscillator can be considered as a filtered noise generator and therefore noise will surround the carrier, equivalent to random FM and AM modulations on the ideal RF sine wave - this additional noise is known as Phase Noise. If we consider the addition of a noise voltage to a sinusoidal voltage, we must take into account the phase relationship. A phasor diagram below can be used to explain the effect.2πft (radians)Figure 1 Phase noise phasor diagram. A phasor with amplitude A can have any value of phasefrom 0 to 360 degrees as represented by the phasor rotating around the origin. Including the phase component gives a phasor of value Asin(2πft) .Noise contains components at many frequencies, so its phase with respect to the main carrier is random, and its amplitude is also random. Noise can only be described in statistical terms because its voltage is constantly and randomly changing, but it does have an average amplitude that can be expressed in RMS volts. Figure 2 shows noise added to the carrier phasor, with the noise represented as a fuzzy, uncertain region in which the sum phasor wanders randomly.Random amplitude variationdue to noise pk-pkFigure 2 Phase noise added to a carrier. The phasor of figure 1 can also have a smaller phasoradded to it due to noise. This additional random noise phasor will cause a ‘circle’ of random values which is phase noise added to the carrier.The phase of the noise is uniformly random – no direction is more likely than any other – but the instantaneous magnitude of the noise obeys a probability distribution as shown.2 How phase noise effects a system.In transmitters local oscillator noise is amplified by the subsequent amplifier stages and is eventually fed to the antenna together with the wanted signal. The wanted signal is therefore surrounded by a band of noise originating from the phase noise of the local oscillator. Therefore the noise generated can spread over several kHz masking nearby lower power stations as shown in figure 3.W anted signal Strong,clean local signalW anted signal m asked by noise Strong,dirty local signalFigure 3 Transmitter spectrum for a clean and noisy local oscillator source. The lower diagram shows how a noisey local oscillator can raise the noise floor, swamping low powersignals close to carrier.The situation is more complicated with receivers and results in reciprocal mixing in the mixer. If we modulate a RF signal and mix it with a clean LO source a modulated IF signal will be the result. If, on the over hand, we mix a clean RF signal with a modulated LO source then again a modulated IF will be the result. To the listener the modulation will appear to be the same, as indeed it is. The effect can be explained by suggesting that the noise components are additional LO’s that are offset from the main carrier. Each of them mixes other signals that are offset from the LO by the receivers IF. Noise is the sum of a infinite number of infinitesimal components spread over a range of frequencies, so the signal it mixes into the IF are spread into an infinite number of small replicas, all at different frequencies.This amounts to a scrambling of these other weaker frequencies into the noise. It is for the reasons given that phase noise is a key design parameter for such applications as satellite repeaters, sensitive communication receivers and mobile phone base stations.3 Limits on phase noise performance - Leeson’s Oscillator Model An oscillator can be considered as an amplifier with positive feedback and initially the contribution of the amplifier noise specified by its Noise factor can be considered.Noise factor F is defined as follows:-()()()()178dB- = 10dBm - dB 6+ dBm 174- = S -:gives 6dB of figure noise a and input the at power 10dBm + a with amplifier an example, an As 1)= (B dBm/Hz 174 - = kTBUsing bandwidth Hz -1 a for 1 = B Where FkTB/P S becomes noise phase of densisty spectral The P FkTbecomesdeviation phase total the , - at exists relation phase random correlated a Since P FkT21; P FkT V V -:by given deviation phase a produces carrier the from + frequency any at bandwidth Hz -1 a in noise phase input The amplifier. free -noise a to power noise input total the is N where kTB N FGkTB = N GkTBN = G N N = S/N S/N= avs RMS 2avstotal 1RMS avs1RMS avs RMS1avs RMS1n peak in in out outin out c m m in f f f fm fo fm fo out F >=∆==∆=∆==∆=ϑϑϑϑϑϑThe phase-noise can be modelled by a noise-free amplifier and a phase detector at the input, as shown below in figure 4.f o /2Q L→ offset f mS θNoise-free amplifierS θ(f m )Phase Detector 1/f flickerFigure 4 Representation of oscillator noise. The close to carrier noise with a slope of 9dB/octaveis due to the flicker noise of the active device and has a cut-off at the flicker corner frequency of 1/f. The 6dB/octave section is due to phase noise according to Leeson’s equation and is a function of loaded Q, noise factor, power, temperature. Above carrier offsets of f o /(2QL) noise is broad-band noise as defined by FkTB/(P avs ).In reality the spectral purity of the carrier is affected by the device generated flicker noise at frequencies close to the carrier and shows a 1/f component with a corner frequency known as fc. The spectral phase noise can be given as:-constant.Boltzman = kfactor noise = F (Hz) offset frequency = fm (Hz)frequency corner flicker =fc (W) resonator in power Average = Pavs Kelvin)(degrees e temperatur = T (Hz)bandwidth frequency = B 1= B 1P FkTB= )( avs ⎟⎠⎞⎜⎝⎛+fm fc f Sm mThe phase noise at the input to the amplifier is likely to be bandwidth limited and in the case of an oscillator this is determined by the Q of the resonator and can be modelled as an amplifier with feedback as shown below in figure 5.S θ(f m ) resonatorOutputS θout (f m ),L(f m )Figure 5 Model of an oscillator for noise analysis. The main components of the system are theresonator, a noise-free amplifier and a noise source (phase modulator).The tank circuit or band pass resonator has a low-pass transfer function: -()resonator. the of bandwidth -half the is B/2 2/ (rad/s)frequency centre =o ; (rad/s) offset carrier = m ; Q loaded =QL Where/2Q j +11= )( o L =L o m m Q L ωωωωωωThese equations describe the amplitude response of a band pass resonator.The assumption is that phase noise is transferred, without attenuation, through the resonator up to the half bandwidth. The closed loop response of the phase feedback is given by: -⎟⎠⎞⎜⎝⎛+⎥⎥⎦⎤⎢⎢⎣⎡⎟⎟⎠⎞⎜⎜⎝⎛⎥⎥⎦⎤⎢⎢⎣⎡⎟⎟⎠⎞⎜⎜⎝⎛⎟⎠⎞⎜⎝⎛+=⎥⎥⎦⎤⎢⎢⎣⎡⎟⎟⎠⎞⎜⎜⎝⎛∆⎟⎟⎠⎞⎜⎜⎝⎛+=∆fm fc f f f f f fm fc f f f f f j f o m m in o m m in m in o m out m in m o m out 1P FkT Q 2f 1+121 = )(L )(S Q 2f 1+121 = )(L Finally 1P FkTB)(S where )(S Q 2f 1+1 = )(S densityspectral phase the becomes transfer power The )(Q 21)(avs 2L 2m 2L 2m avs 2L 2m L ϑϑϑϑϑωωϑwhere L(fm) = Phase noise (dBc/Hz) Q L = loaded Qf m = carrier offset frequency (Hz) f o = carrier centre frequency (Hz)f c = flicker corner frequency of the active device (Hz). T = temperature (°K).Pavs = Average power through the resonator (W). F = Noise factor of the active device. k = Boltzman constant1Q 2f +12P FkT = )(L -:density noise phase sided -single for equation Leeson the us gives finally This Q 4Q 4+12P FkT= )(L 4Q f 14Q f 1+12P FkT= )(L -: expression the out g Multiplyin2L m avs 2L 322L 22avs2L 22m 2L 22m avs ⎥⎥⎦⎤⎢⎢⎣⎡⎟⎠⎞⎜⎝⎛+⎟⎟⎠⎞⎜⎜⎝⎛+⎥⎥⎦⎤⎢⎢⎣⎡++⎥⎥⎦⎤⎢⎢⎣⎡++fm fc f fm fc f f fc f f f fm fc f f fm fc f fm fc f o m m o m o m o omFlicker effect Resonator Q Phase perturbationUsually the phase noise is specified in dBc/Hz ie :-dBc/Hz 1Q 2f +12P FkT10Log = )(L 2L m avs 10⎪⎭⎪⎬⎫⎪⎩⎪⎨⎧⎥⎥⎦⎤⎢⎢⎣⎡⎟⎠⎞⎜⎝⎛+⎟⎟⎠⎞⎜⎜⎝⎛+fm fc f fm fc f omThe Leeson equation identifies the most significant causes of phase noise in oscillators. Therefore it is possible to highlight the main causes in order to be able to minimise them.The relationship between loaded Q, noise factor and centre frequency can be used to derive the single-sideband phase noise performance, for a given frequency offset in the form of the nomograph shown in figure 5.Figure 5 Nomograph for calculating the phase noise of an oscillator. The nomograph is valid for offset frequencies 1/fc to fo/(2Q L), where fc = flicker corner frequency of the activedevice and QL = loaded Q of the resonator.The Leeson equation was evaluated using an ADS model for phase noise byentering, temperature, output power, noise figure and loaded Q. The ADS schematic then uses a phase noise demodulator to produce the predicted phase noise of the oscillator. Figure 7 shows the ADS test bench setup for calculating the phase noise.Figure 8 shows (a) a bipolar L-C oscillator and (b) a FET coaxial resonator oscillator.HB1NoiseNode[1]="PNoise_OL"NLNoiseStop=1 MHzNLNoiseStart=100 Hz NLNoiseMode=yes Order[1]=7Freq[1]=fcentrePhaseNoiseMod MOD2QL=15NF=3 dB Fcorner=10 kHz Rout=50 Ohm Fnom=fcentre MeasEqn meas1PNoise_OLout=real(PNoise_OL[0])VCO_OLout=VCO_OL[2]Figure 6 ADS circuit schematic for predicting the phase noise performance of an oscillator giventhe NF, loaded Q and flicker corner frequency. The P_1Tone block specifies the frequency (fcentre) and the oscillator power (10dBm). The Phase Noise modulator block simulates the noise generating from the oscillator based on NF, loaded Q and flicker corner frequency. The final block de-modulates the noise. The harmonic balance test set is set to measure non-linear noise. The NLNoise start and stop specify the Phase noise sweep range offset from the carrier and the noise node defines where to make the phase noise measurement (ie at the output).Sheet10 of14EqnPhaseNoise=10*log(0.5*VCO_phasenoise..PNoise_OL.noise**2)Figure 7 Phase noise prediction for a Bipolar Colpitts oscillator. The frequency is set to 1GHz and the loaded Q of the resonator is ~ 15. Note that the flicker noise is set to 10KHzwhich, is typical for a bipolar transistor.Eqn PhaseNoise=10*log(0.5*VCO_phasenoise..PNoise_OL.noise**2)Figure 8 Phase noise prediction for a FET Reflection oscillator. The frequency is set again to 1GHz and the loaded Q of the resonator is ~ 15. Note that the flicker noise is set to10MHz which, is typical for a MESFet transistor and dominates the phase noise of thisoscillator.The ADS simulations show how the phase noise is degraded close to the carrier by the addition of flicker noise especially for the GaAs FET device. We would therefore wish to maximise the loaded Q by using a coaxial or dielectric resonator. However this is all very well for a fixed frequency oscillator where we are able to maximise the Q, we generally require a variable frequency oscillator, (VCO) for use in a phase locked loop, to cover a band of frequencies. Such VCO’s require a method of converting the PLL control voltage to frequency and this is normally done with a varactor diode (Vari-capacitance diode).Unfortunately any noise on the PLL control voltage and any internally generated noise will modulate the carrier, increasing the overall phase noise performance. The equivalent noise voltage modulating the varactor is given by Nyquist’s equation:-Hz volts/root TR 4V enr n k =The peak phase deviation in a 1 Hz bandwidth which results from the varactor noise resistance is:-()() 2220L ie 2d20L -: is dBc/Hz in noise phase resulting e Hz/volt.Th in constant gain VCO the is K where 2= d v mnv m m mnv f V K Logf Logf f V K ==ϑϑTherefore, the total single-sideband phase noise will be the power sum of the oscillator phase noise given by the Leeson equation added to the varactor phase noise just given.Varactor modulation noise is most significant in broadband high-frequency VCO’s, because the VCO gain constant is large. The two following spreadsheet calculations show the addition of a varactor with a 10MHz/volt & 100MHz/volt varactor tuning range.Figure 9 shows the phase noise prediction for a 1GHz VCO (NF=10dB P=10dBm and loaded Q = 50) with a varactor tuning range 10MHz/volt, while figure 10 shows a 1GHz VCO (NF=10dB;P=10dBm and loaded Q = 50) with a varactor tuning range 100MHz/Volt.-180.00-160.00-140.00-120.00-100.00-80.00-60.00-40.00-20.000.0010100100010000100000100000010000000Frequency (Hz)P h a s e N o i s e (d B /H z)Figure 9 Phase noise prediction of a VCO with a varactor tuning range of 10MHz/volt. The VCO has the following parameters of noise figure =10dB, output power=10dBm and loaded Q = 50.Figure 10 Phase noise prediction of a VCO with a varactor tuning range of 100MHz/volt. The VCO has the following parameters of noise figure =10dB, output power=10dBm and loaded Q = 50.The previous examples show in the extreme varactor noise and flicker noise can dominate the main cause of noise in an oscillator – that generated by the resonator and specified by it’s loaded Q.In summary, in order to minimise the phase noise of an oscillator we therefore need to ensure the following:-(1) Maximise the Q.(2) Maximise the power. This will require a high RF voltage across the resonator and will be limited by the breakdown voltages of the active devices in the circuit.(3) Limit compression. If the active device is driven well into compression, then almost certainly the noise Figure of the device will be degraded. It is normal to employ some form of AGC circuitry on the active device front end to clip and hence limit the RF power input.(4) Use an active device with a low noise figure.(5) Phase perturbation can be minimised by using high impedance devices such as GaAs Fet’s and HEMT’s, where the signal-to-noise ratio or the signal voltage relative to the equivalent noise voltage can be very high.(6) Reduce flicker noise. The intrinsic noise sources in a GaAs FET are the thermally generated channel noise and the induced noise at the gate. There is no shot noise in a GaAs FET, however the flicker noise (1/f noise) is significant below 10 to 50MHz. Therefore it is preferable to use bipolar devices for low-noise oscillators due to their much lower flicker noise, for example a 2N5829 Si Bipolar transistor, has a flicker corner frequency of approximately 5KHz with a typical value of 6MHz for a GaAs FET device. The effect of flicker noise can be reduced by RF feedback, eg an un-bypassed emitter resistor of 10 to 30 ohms in a bipolar circuit can improve flicker noise by as much as 40dB.(7) The energy should be coupled from the resonator rather than another point of the active device. This will limit the bandwidth as the resonator will also act as a band pass filter.。
Oscillator_Resonators
Oscillator Resonator Design TutorialJ P SilverE-mail: john@1ABSTRACTThis paper discusses the design of various types of resonator that form the heart of any oscillator de-sign.The first section describes the different resonator types including lumped, coaxial, microstrip and dielectric. The following section deals with varactor diodes, including design equations, temperature & loaded Q performance. In the final section the defi-nitions of loaded and unloaded Q are described with a worked example and design techniques on Q transformations.2INTRODUCTIONThe resonator is key to the design of an oscillator. The loaded Q determines the phase noise performance of the oscillator. The oscillator frequency will determine to some degree the type of resonator eg At microwave frequencies resonators can be coaxial or microstrip and at low frequencies the resonators are almost always made up of lumped components.This tutorial gives design data for various types of resonator.3RESONATORS [1]The resonator is the core component of the oscillator, in that it is the frequency selective component and itsQ is the dominating factor for the phase noise per-formance of the oscillator.This section discusses the range of resonators, that can be used for an oscillator covering, dielectric, cavity, transmission line, lumped element and coaxial resona-tors.3.1LUMPED ELEMENTLumped element resonators can be configured to form either a low, high or band pass filter, and the given number of elements is directly related to the Q and loss of the resonator. The simplest resonators can consist of just two elements an inductor and a capacitor ie:-3.2TWO ELEMENT RESONATOR CIRCUITS Figure 1 shows a schematic diagram of a two-element resonator. This circuit is seldom used in oscillators as the loaded Q will be very low as the source and load impedances will directly load the tuned cir-cuit.Q =LRω..2Q =2.R.LωFigure 1 Schematic of a two element, lumped resona-tor, together with loaded Q equations.At resonance the transmission phase is zero and the network is loss less (except for the resistance of the inductor). The series resonator impedes signal trans-mission while the parallel network allows signal trans-mission. The main problem with such a simple resona-tor is achieving a required Q, for example if we want a Q of 30 we would need the following series inductor & capacitor at 1GHz:-0.05pF=947791*21=f21=C477nH=1E9*230*50*2=2.R.Q=L22−⎟⎠⎞⎜⎝⎛⎟⎠⎞⎜⎝⎛EELπππωAlthough the inductor is a realised value the capacitor could not be realised except in perhaps inter-digital form. This could be used if the oscillator is designed for fixed frequency but the value is impracticable as a varactor in a voltage controlled oscillator.The situation can be improved by using more than two elements eg 3 or 4 as described in the next section.3.3 THREE ELEMENT RESONATOR CIRCUITS The diagram below shows a range of three element lumped resonators - Figure 2.LQ R X X X L LC L==2.C2C 2LC X 2X R RX= +=Q R X X X C CL C==2.X L & X CL C ==212ππ....f fFigure 2 Schematic diagram of a range of three ele-ment resonators together with equations to calculate the reactive components and loaded Q.3.4 FOUR ELEMENT RESONATOR CIRCUITS Four element resonators are used most commonly in oscillators as the loaded Q of the resonator can be set independently of the resonant circuit so that sensiblecomponent values can be calculated. Figure 3 shows a four element lumped resonator and Figure 4 shows an alternative configuration.C seriesLFigure 3 Schematic diagram of a four element lumped resonator()()Qunloaded L the is Q where Q 1Q 11= Q where12R = X -:ely approximat is Q loaded given a for reactance The .C of function a is Q Loaded 1= L -:by given is f at resonate to inductance Required resistanceload t input/oupu = R1R R 211C -:is L inductor series the with resonates which e capacitanc Effective u uL e 2/1o cshunt shunt 2series o o 2o 2o e series −⎟⎟⎠⎞⎜⎜⎝⎛−++=−L e o eo shunt o o shunt seriesX Q R C C C C ωωωC seriesFigure 4 Schematic diagram of the alternative four element lumped resonator()resistanceload t input/oupu = R1R 2C -:is L inductor shunt the with resonates which e capacitanc Effective L.f 21= Ce -: inductor shunt resonate to e Capacitanc admittance inductor shunt given a is B & Qunloaded L the is Q whereQ 1Q 11= Q where X ..21C 12R = X o 2o shunt series 2L u uL e cseriesseries 2/1o cseries +−=⎟⎠⎞⎜⎝⎛−=∴⎟⎟⎠⎞⎜⎜⎝⎛−series o seriesL e o C C Ce f B Q R ωππ3.5 COAXIAL CABLE RESONATOR [2] A quarter-wave coaxial resonator is formed by short-ing the centre conductor of a coaxial line to its shield at one end, leaving the other end open-circuited. The physical length of the resonator is equal to one quarter the wavelength (90 degrees electrical length) in the medium filling the resonator. A diagram of a coaxial resonator is shown below in Figure 5.λ/4Figure 5 Schematic diagram of a coaxial cable resonator showing the critical dimensions .coaxcoax 41length Resonator = ;f2.99E8= λελλλ==rair airThe unloaded Q of the resonator is a function of theconductor losses, the dielectric losses and the physical dimensions of the coaxial cable ie:1-12-r r D C DC U Fm 8.854x10=y;permitivit relative ;1ie dielectric of ty conductivi = ..f.2= factor) ssipation Tangent/Di (Loss tan. = Q bygiven is conductors the separates that dielectric the from on contributi Q The conductorsthe of ty conductivi = and ty permeabili = whereb1a 1a b Ln....2.= Q bygiven is and conductors the in flow current to due lost energyto due is conductor from on contributi Q The Dielectric = D & Conductor = C e wher Q 1 Q 1Q 1o of εερσεεπσδσμσμπ=++=3.6 DESIGN EXAMPLE OF A COAXIAL CABLERESONATORThe following example is for the design of a coaxial resonator to operate in an oscillator at 1GHz. The reso-nator is made from semi-rigid coaxial cable that con-tains a dielectric of PTFE, which has a relative permit-tivity of ~ 2.2 and a tan δ of 0.0004.5.04cm= 36090.2.21E92.99E8= length Resonator3.7 CALCULATION OF RESONATOR QFACTORThe Q factor of the resonator determines the phase noise performance of the oscillator. Loss in the coax-ial cable from the conductivity of the sheath and the loss tangent of the dielectric will set the Q of the reso-nator. Most coaxial cables especially semi-rigid cables use copper as the conductor, therefore the equation forthe Q contribution for the conductor ie Qcc is given by:The dielectric of the cable also effects the Q of the resonator and is given by:92.95Q (0.000358) 3.58mm = b example above For 3.58mmor 0.141" is cable rigid -semi typicalof diameter Overall f 8.398.b. = Q unloaded to on contributi Conductor = Q cc cc =∴The dielectric of the cable also effects the Q of the resonator and is given by:6.98 2500192.951= Q 1+ Q 1= Q 1unloaded Total 25000.00041Q10GHz @ 0.0004 ~ PTFE for tan material dielectric of tangent loss tan.1= Q unloaded to on contributi loss Dielectric = Q d cc d d =+==∴δδNote the Q cc term dominates the overall Q factor of the resonator at this frequency.The table below shows (Table 1) design data for a range of common materials used in the construction of coaxial cables:-Material εr ρ tan δ Copper - 1.56E-8Ω.m - Gold - 2.04E-8Ω.m - Silver - 1.63 E-8Ω.m- Nylon 3.0 109-1011Ω.m 0.012@3GHz PTFE 2-2.1 1E-16 0.0004@10GHzPolythene HD 2.25 >1014Ω.m 0.0004@10GHz PVC flexi 4.5 109-1012Ω.mTable 1 Design data for a range of materials com-monly used in the construction of coaxial cables. The parameters shown are relative permittivity (εr), resistivity ρ (1/ρ = conductivity) and tan delta (tan δ).3.8 COAXIAL RESONATOR [3]A quarter-wave coaxial resonator is formed, by plating a piece of dielectric material with a high relative per-mittivity using a highly conductive metal.A cylindrical hole is formed along the axis of a cylin-der of high relative permittivity dielectric material. All surfaces, apart from the end surface, are coated with a good conductor to form the coaxial resonator. The physical length of the resonator is equal to one quarter the wavelength (90 degrees electrical length) in the medium filling the resonator. The diagram (Figure 6) below shows the key dimensions of a coaxial resona-tor.W≡Figure 6 Schematic diagram of a coaxial resonator showing the key dimensions. Note the resonator is plated with silver except for one end to allow it to be grounded.The expression for the unloaded Q of such a resonator is()⎟⎠⎞⎜⎝⎛⎟⎠⎞⎜⎝⎛+⎟⎠⎞⎜⎝⎛d W .079.1.60=Z Impedance Input 88.5 of with dielectric sivered a for 200 = 38.6 of with dielectric silvered a for 240 = k mmin diameter inside = d mm, in diameter outside = where d 1W 14.25d W .079.1Ln .o k. = r in L Ln W f r r εεεπεπε4.Zo.Q= Resistance .103*2*4.25.= e Capacitanc mmin length Physical = 103.4.25.8.Zo.= Inductance 8r82r Zox x l l lBelow resonance, such short-circuited coaxial lineelements simulate high-Q, temperature stable ‘ideal’inductors. They will only realise an ‘ideal’ inductor over a narrow range as shown in the diagram Figure 7.S elf R eson an t F req u en cyFigure 7 Frequency response of a coaxial resonator. The first region shows an area of inductance followed by a point of resonance followed by a region of ca-pacitance. The resonator is usually used below the self-resonant frequency so that in a VCO the varactor can be used to resonate with the coaxial resonator.In order to use the coaxial resonator as a ‘ideal’ induc-tor the resonator must be used below the self-resonant frequency.3.9 DESIGN EXAMPLE OF A COAXIALRESONATOR [4,5,6]The following section describes the design of a coaxial resonator to be used in a varactor controlled oscillator at 900MHz. We need therefore to select a suitable resonator that is inductive at 900MHz.Assume an ‘ideal’ starting inductance of 4nH at 900MHz.The material chosen is a silver-plated ceramic resona-tor with a relative permittivity of 38.6 from Transtech. It has a tab inductance of 1nH, a W/h ratio of 2.57, a width of 6mm and a characteristic impedance of 9.4Ω.9.74mm = 9.415.1tan .26036.0= Z Z tan .2 = resonator of Length .900MHz at 15.1 is reactance whose 3nH = 1-4 ie inductance required the from inductance tab the subtract We 60.36mm= 6.3800E 8/3E= c/= Wavelength 1o input 1g 68ro⎟⎠⎞⎜⎝⎛⎟⎟⎠⎞⎜⎜⎝⎛Ω−−ππλεflong0.161 = 0.60360.0973is line coaxial the Therefore 1241MHz= 0973.01.4800*6036.0=MHz 1.4.=Frequency Resonant Self 415.7= 0.0024610.00614.250.002460.006.079.1Ln .6E 800240. = Q= d 1W 14.25d W .079.1Ln .o k. = Q g g λλlo f f ⎟⎠⎞⎜⎝⎛+⎟⎠⎞⎜⎝⎛⎟⎠⎞⎜⎝⎛+⎟⎠⎞⎜⎝⎛The part resonance could be tested to ensure that it occurs at the self-resonant frequency of 1.241GHz.3.10 DIELECTRIC RESONATOR [7]At lower frequencies the length of W/d ratio of a coax-ial resonator becomes too big to realise so a dielectric ‘puck’ is used instead. The dielectric resonator is often made from the same material as the coaxial resonators except that they are not plated with a low-loss metal. In addition they are mounted on planer circuits as shown below (figure 35) and are coupled to a trans-mission line without a direct connection. As with other resonators, standing TE waves will be set up within the resonator, which will be dependent on the physical dimensions of the cylinder.The diagram of a dielectric resonator is shown below in Figure 8abFigure 8 Schematic diagram of a dielectric reso-nator showing the key dimensions.The most common resonant mode in dielectric resona-tors is the TE 01δ mode and when the relative dielectric constant is around 40, more than 95% of the stored energy are located within the resonator. For an ap-proximate estimation of the resonant frequency in TE 01δ mode of an isolated dielectric resonator, the fol-lowing simple formula can be used:⎟⎠⎞⎜⎝⎛+=45.3L a ..a 34F (mm)GHzr εThe above equation is accurate to about 2% in therange0.5 < a/L < 2 and 30 < εr < 50The approximate Q factor of the resonator is directly related to the dielectric loss ie tan δ.()r o εεωσδδ..= tan tan 1 Q unloaded =3.11 DESIGN EXAMPLE OF A DIELECTRICRESONATORThe following section describes the design of a dielec-tric resonator for a frequency of ~ 7GHz. A manufac-turer of dielectric resonators – Transtech can supply two relative permittivities of 30 and 38. The Trans-Tech D8733-0305-137 puck was selected with the following parameters, εr = 30, Diameter = 7.75mm, Height = 3.48mm, the resonant frequency can be esti-mated using:7.313GHz = 45.33.4793.8735.30.8735.33445.3L a ..a 34F (mm)GHz ⎟⎠⎞⎜⎝⎛+=⎟⎠⎞⎜⎝⎛+=r εThis calculated figure assumes that the resonator is in free-space. If the resonator is mounted on a substrate in a cavity then this will significantly alter the resonant frequency. A more accurate model to take into ac-count cavity and substrate is the Itoh and Rudokas model [7] which, is shown below in Figure 9:Region 6Figure 9 Itoh & Rudokas model of a dielectric resonator inside a metallic shielded cavityThis model can be simplified to the numerical solution of a pair of transcendental equations:()()⎟⎟⎠⎞⎜⎜⎝⎛+−−=000101201462o 0GHz)((mm)o y 291.0y 2.43+12.4048y +2.4048=a k 2.4048be to taken is x x a k y L height the calculate to entered is frequency initial An.a .150= a k ρεεπr r f()()[]2221111121620220212120211L .coth tan L .coth tan 1=L Length Resonator k .k = -: is 6 and 4 regions to common constant n propagatio The .k k .k k -: are 2 and 1 regions in constants n attenuatio The 1ααβεβεαεαβαβαρρρ−−+−−=−=r r r3.12 COUPLING OF RESONATOR TOMICROSTRIP LINE [8]For analysis of the resonator coupled to a micro-strip line, the transformation shown in the Figure 10 below is used. β (coupling coefficient) is used to provide an equivalent series resistance for the resonator:-R≡Figure 10 Dielectric resonator coupled to a micro-strip line and the corresponding circuit diagram. The resistor L simulates the coupling of the L-C resonant circuit of the dielectric resonator.Calculation of loaded Q:()βββπ = 1Q Q +1Q Q *Zo *2 = R 21= LC LU UL 2−⎟⎟⎠⎞⎜⎜⎝⎛=fWith the above equations it is possible to design VCO for a given Q for example if we want a minimum Q of 1000:Ω−−⎟⎟⎠⎞⎜⎜⎝⎛=4K = 4*50*2 *Zo *2 = R of resistor series a with resonator the replace can we CAD a on analysing For4 = 110005000 = 1Q Q +1Q Q5000of Q unloaded a with Resonator a use we If L U U L βββTrans-Tech have a CAD package [15] to calculate various design parameters using their dielectric resona-tors. We can use the CAD package to calculate a plot of the coupling coefficient β vs distance from the cen-tre of the micro-strip line to the centre of the DRO puck. The plot of the analysis is shown below in Figure 11.Figure 11 Plot of coupling coefficient (β) with dis-tance from the centre of the puck to the centre of the microstrip line in mmTherefore, in our example, the puck would be placed at a distance of 7.15mm from the puck centre to the mi-cro-strip line centre.3.13 TRANSMISSION LINE RESONATOR [9] Over a narrow bandwidth L-C lumped components can be realised using short-circuit and open-circuit trans-mission lines. If we analyse a transmission line termi-nated in a load Z L we can define the transformed im-pedance in terms of the characteristic line impedance and the electrical length of the transmission line. The diagram below (Figure 12) shows a transmission line loaded with Z L .Z (in) →l=0Figure 12 Transmission line loaded with load ZL[][]()()()()lj e e l e e e e Zo ee Zl e e Zo e e Zl Zo in Z e Zo Zl e Zo Zl e Zo Zl e Zo Zl Zo I Vin Z ZoZ Zo Z V V e v e v Zoe v e v IV in Z l j l j l j l j l j l j lj l j l j l j l j l j l j l j l j l j L L l j l j lj l j .sin 2)(.cos 2)()()()()(.)(.....)(1221121)(....................ββββββββββββββββββββββ=−=+⎥⎦⎤⎢⎣⎡++−−++=⎥⎥⎦⎤⎢⎢⎣⎡−−+−++==∴+−==−+==−−−−−−−+−+−+−+⎥⎦⎤⎢⎣⎡+−+−++=∴⎥⎦⎤⎢⎣⎡++=∴−−−−l j lj l j l j l j l j l j l j e Zo eZl e Zo e Zl e Zo e Zl e Zo e Zl Zo in Z l Zo l j Zl l j Zo l Zl Zo in Z .................)(.cos 2..sin 2..sin 2..cos 2..)(ββββββββββββ⎥⎦⎤⎢⎣⎡++=⎥⎥⎥⎥⎦⎤⎢⎢⎢⎢⎣⎡++⎥⎥⎥⎦⎤⎢⎢⎢⎢⎣⎡++l Zl Zo l Zo Zl Zo in Z l l j Zl l l j Zo Zo Zl Zo l l Zo l l j Zo ll j Zl l lZl Zo .tan ..tan ..)(.cos .sin ..cos .sin ...cos 2.cos 2..cos 2.sin 2..cos 2.sin 2..cos 2.cos 2...l 2cos by through divide βββββββββββββββThis equation is the general expression for the imped-ance looking into a load Z L via a length of transmis-sion line. If we now have the case where the transmis-sion line is terminated with a short circuit we find the general expression simplifies ie let Z L = 0 thenZ in Zo Zl Zo l Zo Zl l ()..tan ..tan .tan .=++⎡⎣⎢⎤⎦βββ = jZ ( Short circuit)o lWe can now plot the impedance (Figure 13) of the shorted length of transmission line vs electrical length and we get the following graph, which shows how the transmission line equates to lumped capacitance and inductance with resonance’s in between.In general Z (in) = R (in) + jX (in) For S/CCT R (in) = 0 ; X (in) = Z o tan β.LZ o tan β.L is purely reactive varies between - ∞ & + ∞ as L variesl=0θ = β.L←← λgX = Z =2 =. = .v= v o gtan .βπλβωωll l l ⎛⎝⎜⎞⎟Figure 13 Plot of impedance against length of a short circuited transmission line. The plot shows how the reactance of the transmission line varies between inductive and capacitive reactances with resonant frequency regions in between.Each region of figure 40 is now described:(1) If θ between 0 & π/2 tan β.L is positive ∴X is +ve ⇒ j(ω.L) - INDUCTIVE.(2) If π/2 < θ < π tan β.L is -ve ∴ X is -ve ⇒ j(-1/ω.C) - CAPACITIVE.(3) If θ ≈ 0, π , 2π | X | goes to a minimum ie:-| X |θ≅L.C(4) If θ ≅ π/2 , 3π/2 | X | goes to a maximum:-| X |θ≅L //CSimilarly, for a transmission line terminated by an open circuit we can repeat the analysis, but we divid-ing through by ZL. Note Zo/Z L tends to zero ie:-Z L = ∞V=Max at O/cctZ (in)[]circuit)Open ( .tan 1jZ = .tan ..ie Z by bottom & top divide .tan ..tan ..)(o L ⎟⎟⎠⎞⎜⎜⎝⎛⎥⎥⎥⎦⎤⎢⎢⎢⎣⎡++⎦⎤⎢⎣⎡++=l ββββZl ZlZl l Zo Zl Zl Zo l Zl Zo l Zo Zl Zo in ZAgain we can plot the impedance against electrical length of the transmission line (Figure 14) to see the equivalent lumped reactance and resonance points.In general Z (in) = R (in) + jX (in) For O/CCT R (in) = ∞ ; X (in) = Z o cot β.LZ o cot β.L is purely reactive varies between - ∞ & + ∞ as Ll=0θ = β.L3λ/4← λ X = Z =2 = . = .v= o gcot .βπλβωωll l l ⎛⎝⎜⎞Figure 14 Plot of impedance against length of a open circuited transmission line. The plot shows how theSheet 11 of 20reactance of the transmission line varies between inductive and capacitive reactance’s with resonant frequency regions in between.The previous graphs show that we can realise lumped components from transmission lines eg3.14 DESIGN EXAMPLE OF INDUCTOR USINGA TRANSMISSION LINEThe following section describes the process of design-ing a transmission line to have a specific inductance of 0.7nH at a frequency of 8.8GHz. The transmission line is to be etched on RT duroid substrate material, which has a relative permittivity of 2.94 and a sub-strate thickness of 0.25mm.ll for Solve 2 = where .tan 1j.Zo - = Zin 0.466pF= C C =8.8GHzat 0.7nH of inductance of 38.8 = Reactance gf 212λπββπ⎟⎟⎠⎞⎜⎜⎝⎛∴Ω⎟⎠⎞⎜⎝⎛LUsing the transmission line equation for an open-circuit stub we can calculate the electrical length re-quired for an inductance of 0.7nH.Therefore a open-circuit stub of length 3.1mm will have an inductance of 0.7nH at 8.8GHz.As the equations show the resulting impedance is a function of the characteristic of the line and generally we use a narrow high impedance line ~ 100Ω for an inductive impedance and a wide length of line ~ 20Ω, for a capacitive impedance. For completeness the em-pirical equations for calculating line widths are given in the next section:-3.1mm = 29338.950arctan = X Z arctan = 293= 0214.02= 21.4mm or 0.0214m = 53.23E8/8.8E9=therefore 2.94 is used be to material the of ty permittivi Relative o e g ⎟⎠⎞⎜⎝⎛⎟⎠⎞⎜⎝⎛=βπβελλl ff air3.15 CALCULATION OF EFFECTIVE RELATIVE PERMITTIVITY [10]The following section describes the empirical equa-tions that are used to calculate the dimensions of the micro-strip lines and characteristic impedance [8]. The first equation describes the effective relative permittiv-ity which, differs from the specified value due the width of the micro-strip track.()())1.18/(17.181432.0/)52/(/491+1 = a and 39.00.564 = b where .1012121= 3424053.0r r .r r h W Ln h W h W h W Ln w h ba eff +⎟⎠⎞⎜⎝⎛+⎥⎦⎤⎢⎣⎡++⎟⎠⎞⎜⎝⎛⎟⎟⎠⎞⎜⎜⎝⎛+−⎟⎠⎞⎜⎝⎛+−++−εεεεε Calculation of W/h (width of micro-strip/substrate thickness) for a given characteristic impedance and effective relative permitivity:ro r r r ro 2Z 377= B where 0.517-0.293+1)-Ln(B 21+1)-Ln(2B -1-B 2h W 2 - 44 Z For επεεεπε⎭⎬⎫⎩⎨⎧⎥⎦⎤⎢⎣⎡−=≤⎟⎟⎠⎞⎜⎜⎝⎛++−++−=≥r r r r 2ro 12.0226.0.1160.21= n where 28h W 2 - 44 Z For εεεεεZoe e nn3.16 INTER-DIGITAL MICRO-STRIPCAPACITORS [11]Normally resonators need to be lightly coupled in or-der to maintain a high Q, this can be done by using a filter arrangement or by using very small value capaci-tors. Normal chip capacitors can go as low as 0.1pF, but for smaller capacitance it is convenient to use transmission line inter-digital capacitors.Literature on the subject is very scarce so a basic de-sign formula was used to get the initial dimensions and the final dimensions were optimised during RF simula-tions.The basic formula for the inter-digital capacitor is given by:-fingerslong 600um =cm 06.01)-0.83(20.05= L )1(N *0.83C-:be will fingers the of length the then fingers 2 are there that assume we if and capacitor 0.05pF a want we if example For 10umof width finger a and 5um of spacing finger a assumes formula This pFin e Capacitanc C cm in fingers of Length = L fingers of Number = N Where L).1(N 0.83 = C F F F ==−=−To further aid in the evaluation of a inter-digital ca-pacitor the model was analysed in Libra RF CAD with a finger width and gaps of 0.1mm and number of fin-gers 2,3 & 4.The graph (Figure 15) shows the relationship between capacitance and finger length.Figure 15 Graph of a micro-strip inter-digital ca-pacitor vs capacitance. The plots were calculatedby analysis on HP/Eesof libra.Transmission lines may be used as single resonators capacitively coupled to the active device, but also they may be configured as a micro-strip band-pass filter. The basic principle involves using open circuit trans-mission lines of electrical length 180 degrees, which is equivalent to a ‘tuned circuit’ parallel resonator. What tends to differ in the topographies are the ways in which the resonators are coupled together. The resona-tors can be end coupled or parallel coupled using the gaps between them as the low value coupling capaci-tors. It is also possible to use inter-digital capacitors to generate coupling capacitors less than 1pF3.17 VARACTORS [12]Voltage variable capacitors or tuning diodes are best described as diode capacitors employing the junction capacitance of a reverse biased PN junction. The ca-pacitance of these devices varies inversely with the applied reverse bias voltage.The general equation for calculating the capacitance of the varactor is :-exponente Capacitanc = and 0.7V)(~ potential contact junction = voltage,applied =V e;capacitanc diode C where)(D γφφγ=+=V C C DJ3.18 DESIGN EXAMPLE OF A VARACTORDIODEThe following section describes how information from a data sheet can be used to predict the capacitance of the varactor diode for a given reverse bias. For this example the varactor diode selected is a Macom Tun-ing diode type MA46H071.The data sheet gives the following parameters for the diode:-C = 0.9-1.1pF @ 4V;cap ratio Cto/Ct20 = 5.5;Gamma=0.75;Q @ 50MHz=450075.0120.7512-J D )7.0(E 19.3 =bias given a for e capacitanc a calculate to therefore,3.19pF 0.7)+(41E = ).(C = C give to rearrange )(+==++=−V C V V C C J DJ γγφφThis is obviously the ideal case as it does not take into account the case parasitics3.19 TUNING RATIOSThe tuning or capacitance ratio, TR, denotes the ratio of capacitance obtained with two values of applied bias voltage. This ratio is given by the following:-γφφ⎥⎦⎤⎢⎣⎡+211J 2J V +V = )V (C )V (C = TRwhere C J (V 1) = junction capacitance at V 1;C J (V 2) = junction capacitance at V 2 (V 1>V 2).3.20 CIRCUIT QThe Q of the varactor can be very important, because the varactor usually directly forms the tuned circuit and the overall Q is dominated by the worst Q factor. The Q of tuning diode capacitors falls off at high fre-quencies because of the series bulk resistance of the silicon used in the diode. The Q also falls off at low frequencies because of the back resistance of the re-verse-biased diode.The equivalent circuit of a tuning diode is often shown in the form given below in Figure 16.RpFigure 16 Equivalent circuit of a typical varctor diode together with case and lead parasitic components.Where Rp = Parallel resistance /back resistance of the diode.Rs = Bulk resistance of the silicon in the di-ode.Ls’ = External lead inductance. Ls = Internal lead inductance. Cc = Case Capacitance.Normally the lead inductance and case capacitance can be ignored, which results in a simplified circuit shownin Figure 17.RpFigure 17 Simplified model of a typical varactor diode with parasitic reactance removed.The resulting Q for the above circuit is given by :-ΩΩ=922230x10 = Rp & 1 = Rs TypicallyRs.Rp C)2(+Rp +Rs C.Rp 2f f Q ππTherefore for a MA/COM MA46H071 we would ex-pect the following Q’s at different frequencies as shown in the table below:f(GHz) Q0.05 3500 2 88 6 30The degradation of Q at microwave frequencies means that the varactor, has to be lightly coupled, or Q trans-formed in order not to load the resonant circuit, lower-ing the loaded Q with the resultant degradation in phase noise performance.The following graph (Figure 18) of the varactor diode frequency response shows that at low frequencies the Q is dominated by the parallel term ie Qp = 2πf.Rp.C and at high frequencies by the series term Qs = 1/(2πfRs.C).。
Richtek RT2613A 电源控制器说明书
RT2613A®Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.©General DescriptionThe RT2613A is a high efficiency step-down converter and capable of delivering 3A output current over a wide input voltage range from 2.95V to 6V.The RT2613A provides accurate regulation for a variety of loads with ±3% accuracy. For reducing inductor size, it provides up to 2MHz switching frequency. The efficiency is maximized through the integrated 45m Ω MOSFETs and 550μA typical supply current.Under voltage lockout voltage of the RT2613A is 2.7V,and it also provides external setting by a resistor network on the enable pin.The RT2613A provides protections such as inductor current limit under voltage lockout and thermal shutdown.The over temperature threshold is 145°C.The RT2613A is available in WQFN-16L 3x3 package.2.95V to 6V Input, 3A Output, 2MHz, Synchronous Step-Down ConverterFeatures●Integrated 45m Ω MOSFETs ●Input Range : 2.95V to 6V●Adjustable PWM Frequency : 700kHz to 2MHz ●Output Current : 3A ●95% Efficiency●Adjustable Soft-Start ●Power Good Indicator ●Enable Control●Under Voltage Lockout ●Current Limit●Thermal ShutdownApplications●Low-Voltage, High-Density Power Systems ●Distributed Power Systems ●Point-of-Load ConversionsSimplified Application CircuitOrdering InformationNote :Richtek products are :❝ RoHS compliant and compatible with the current require-ments of IPC/JEDEC J-STD-020.❝ Suitable for use in SnPb or Pb-free soldering processes.Marking Information6Z= : Product CodeYMDNN : Date CodeRT2613AG : Green (Halogen Free and Pb Free)V OUTRT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Function Block DiagramPin Configurations(TOP VIEW)WQFN-16L 3x3GNDGND VIN VIN SW SW SS/TRSW A G N D C O M P F B R T /S Y N CI N N O O TG O O DRT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.OperationThe RT2613A is a synchronous step-down DC/DC converter with two integrated power MOSFETs. It can deliver up to 3A output current from a 2.95V to 6V input supply. The RT2613A's current mode architecture allows the transient response to be optimized over a wider input voltage and load range. Cycle-by-cycle current limit provides protection against shorted outputs and soft-start eliminates input current surge during start-up. The RT2613A is synchronizable to an external clock with frequency ranging from 700kHz to 2MHz. The RT2613A is available in WQFN-16L 3x3 package.High side MOSFET peak current is measured by internal sensing resistor. The Current Signal is where Slope Compensator works together with sensing voltage sensing resistor. The error amplifier adjusts COMP voltage by comparing the feedback signal (V FB ) from the output voltage with the internal 0.827V reference. When the load current increases, it causes a drop in the feedback voltage relative to the reference, the COMP voltage then rises to allow higher inductor current to match the load current.EN ComparatorThe RT2613A is enable when EN pin is higher than 1.25V.It is disable when EN pin lower than 1.18V. There is an internal pull-high current source to charge the EN pin to high when the EN pin is floating.Oscillator (OSC)The internal oscillator that provides switching frequency from 700kHz to 2MHz. It is adjusted using an external timing resistor. It also can be synchronized by an external clock in the range between 700kHz and 2MHz from RT/SYNC pin.PGOOD ComparatorWhen the feedback voltage (V FB ) rises above 93% or falls below 107% of reference voltage the PGOOD open drain output will be high impedance. The PGOOD open drain output will be internally pulled low when the feedback voltage (V FB ) falls below 88% or rises above 113% of reference voltage.Soft-Start (SS)An internal current source (2.2μA) charges an external capacitor to build the soft-start ramp voltage (V SS ). The V FB voltage will track the V SS during soft-start interval.The Soft-Start setting capacitor (C SS ) for the Soft-Start time (T SS ) can be easily calculated by the following equation :Over Temperature Protection (OTP)The RT2613A implement an internal over temperature protection. When junction temperature is higher than 145°C, it will stop switching. Until the junction temperature decreases below 125°C, the RT2613A will re-soft-start from initial condition.⨯μSS SS T (ms) 2.2 (A)C (nF) =0.827 (V)RT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Electrical Characteristics(V IN = 5V, C IN = 10μF, T A = −40°C to 85°C, unless otherwise specified)Absolute Maximum Ratings (Note 1)●Supply Input Voltage, VIN ----------------------------------------------------------------------------------------- −0.3V to 7V●Switch Node Voltage, SW ----------------------------------------------------------------------------------------- −0.3V to (V IN + 0.3V)●BOOT to SW --------------------------------------------------------------------------------------------------------- −0.3V to 7V●Other Pins ------------------------------------------------------------------------------------------------------------- −0.3V to (V IN + 0.3V)●Power Dissipation, P D @ T A = 25°CWQFN-16L 3x3-------------------------------------------------------------------------------------------------------2.128W ●Package Thermal Resistance (Note 2)WQFN-16L 3x3, θJA -------------------------------------------------------------------------------------------------47°C/W WQFN-16L 3x3, θJC -------------------------------------------------------------------------------------------------7.5°C/W ●Junction Temperature ----------------------------------------------------------------------------------------------150°C ●Lead Temperature (Soldering, 10 sec.)------------------------------------------------------------------------260°C●Storage T emperature Range -------------------------------------------------------------------------------------- −65°C to 150°C ●ESD Susceptibility (Note 3)HBM (Human Body Model)----------------------------------------------------------------------------------------2kVRecommended Operating Conditions (Note 4)●Supply Input Voltage ------------------------------------------------------------------------------------------------2.95V to 6V ●Junction T emperature Range -------------------------------------------------------------------------------------- −40°C to 125°C ●Ambient T emperature Range -------------------------------------------------------------------------------------- −40°C t o 85°CRT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Note 1. Stresses beyond those listed “Absolute Maximum Ratings ” may cause permanent damage to the device. These arestress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability.Note 2. θJA is measured at T A = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC ismeasured at the exposed pad of the package.Note 3. Devices are ESD sensitive. Handling precaution is recommended.Note 4. The device is not guaranteed to function outside its operating conditions.RT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Typical Application CircuitTable 1. Recommended Component SelectionV OUTRT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Typical Operating CharacteristicsReference Voltage vs. Temperature0.800.810.820.830.840.85-50-25255075100125Temperature (°C)R e f e r e n c e V o l t a g e (V )Frequency vs. Temperature9009209409609801000-50-25255075100125Temperature (°C)F r e q u e n c y (k H z )Frequency vs. Input Voltage0.920.930.940.950.960.970.980.991.001.011.0233.544.555.56Input Voltage (V)F r e q u e n c y (M H z )Efficiency vs. Output Current1020304050607080901000.0010.010.1110Output Current (A)E f f i c i e n c y (%)Efficiency vs. Output Current1020304050607080901000.0010.010.1110Output Current (A)E f f i c i e n c y (%)Output Voltage vs. Output Current1.801.811.821.831.841.850.511.522.53Output Current (A)O u t p u t V o l t a g e (V)RT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Output Ripple Time (500ns/Div)I L (1A/Div)V OUT (10mV/Div)V SW(5V/Div)V IN = 3.3V, V OUT = 1.8V, I OUT = 1ACurrent Limit vs. Temperature6.06.67.27.88.49.0-50-25255075100125Temperature (°C)C u r r e n t L i m i t (A )Current Limit vs. Input Voltage6.06.57.07.58.08.59.033.544.555.56Input Voltage (V)C u r r e n t L i m i t(A )Output RippleTime (500ns/Div)I L (2A/Div)V OUT (10mV/Div)V SW (5V/Div)V IN = 3.3V, V OUT = 1.8V, I OUT = 3AUVLO vs. Temperature2.32.42.52.62.72.82.9-50-25255075100125Temperature (°C)U V L O (V)Enable Threshold Voltage vs. Temperature1.001.081.161.241.321.40-50-25255075100125Temperature (°C)E n a b l e T h r e s h o l d V o l t a g e (V )RT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.6080100120140160180200R RT (k )S w i t c h i n g F r e q u e n c y (k H z )ΩSwitching Frequency vs. R RT3004005006007008009001000R RT (k )S w i t c h i n g F r e q u e n c y (k H z )ΩLoad Transient ResponseI OUT (1A/Div)V OUT(200mV/Div)V IN = 3.3V, V OUT = 1.8V, I OUT = 0A to 3ATime (100μs/Div)Load Transient ResponseTime (100μs/Div)I OUT (1A/Div)V OUT(200mV/Div)V IN = 5V, V OUT = 3.3V, I OUT = 0A to 3ART2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Application InformationThe basic RT2613A application circuit is shown in Typical Application Circuit. External component selection is determined by the maximum load current and begins with the selection of the inductor value and operating frequency followed by C IN and C OUT . The switching frequency range from 700kHz to 2MHz. It is adjusted by using a resistor to ground on the RT/SYNC pin.Output Voltage SettingThe resistive divider allows the FB pin to sense the output voltage as shown in Figure 1.Figure 1. Setting the Output VoltageThe output voltage setting range is 0.827V to 3.6V and the set by an external resistive divider is according to the following equation :⎛⎫+ ⎪⎝⎭OUT FB R1V = V 1R2where V FB is the feedback reference voltage 0.827V (typ.).Inductor SelectionFor a given input and output voltage, the inductor value and operating frequency determine the ripple current. The ripple current ΔI L increases with higher V IN and decreases with higher inductance :⎡⎤⎡⎤∆⨯-⎢⎥⎢⎥⨯⎣⎦⎣⎦OUT OUT L OSC IN V V I =1f L V Having a lower ripple current reduces the ESR losses in the output capacitors and the output voltage ripple. Highest efficiency operation is achieved at low frequency with small ripple current. This, however, requires a large inductor. A reasonable starting point for selecting the ripple current is ΔI L = 0.4 (IMAX). The largest ripple current occurs at the highest V IN . To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation :⎡⎤⎡⎤⨯-⎢⎢⎥⨯∆⎣⎦⎣⎦OUT OUT OSC L(MAX)IN(MAX)V V L =1f I V Input and Output Capacitors SelectionThe input capacitance, C IN , is needed to filter the trapezoidal current at the source of the top MOSFET. A low ESR input capacitor with larger ripple current rating should be used for the maximum RMS current. RMScurrent is given by :RMS OUT(MAX)I = I This formula has a maximum at V IN = 2V OUT , where I RMS =I OUT / 2. This simple worst case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000hours of life, which makes it advisable to either further derate the capacitor or choose a capacitor rated at a higher temperature than required. Several capacitors may also be placed in parallel to meet size or height requirements in the design. The selection of C OUT is determined by the effective series resistance (ESR) that is required to minimize voltage ripple, load step transients, and the amount of bulk capacitance that is necessary to ensure that the control loop is stable. Loop stability can be examined by viewing the load transient response as described in a later section. The output ripple, ΔV OUT , is determined by :⎡⎤∆≤∆+⎢⎥⎣⎦OUT L OSC OUT 1V I ESR 8f C Using Ceramic Input and Output CapacitorsHigher values, lower cost ceramic capacitors are nowbecoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, V IN . At best, this ringing can couple with the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at V IN large enough to damage the part.V OUT11DS2613A-01 July 2015RT2613A©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.Thermal ConsiderationsFor continuous operation, do not exceed absolute maximum junction temperature. The maximum power dissipation depends on the thermal resistance of the IC package, PCB layout, rate of surrounding airflow, and difference between junction and ambient temperature. The maximum power dissipation can be calculated by the following formula :P D(MAX) = (T J(MAX) − T A ) / θJAwhere T J(MAX) is the maximum junction temperature, T A is the ambient temperature, and θJA is the junction to ambient thermal resistance.For recommended operating condition specifications, the maximum junction temperature is 125°C. The junction to ambient thermal resistance, θJA , is layout dependent. For WQFN-16L 3x3 package, the thermal resistance, θJA , is 47°C/W on a standard JEDEC 51-7 four-layer thermal test board. The maximum power dissipation at T A = 25°C can be calculated by the following formula :P D(MAX) = (125°C − 25°C) / (47°C/W) = 2.128W for WQFN-16L 3x3 packageThe maximum power dissipation depends on the operating ambient temperature for fixed T J(MAX) and thermal resistance, θJA . The derating curve in Figure 2 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation.Figure 2. Derating Curve of Maximum Power Dissipation0.00.40.81.21.62.02.40255075100125Ambient Temperature (°C)M a x i m u m P o w e r D i s s i p a t i o n (W )Layout ConsiderationsFor the best performance of the RT2613A, the following guidelines must be strictly followed.❝The input capacitor should be placed as close as possible to the device pins (VIN and GND).❝The RT/SYNC pin is sensitive. The RT resistor should be located as close as possible to the IC and minimal lengths of trace.❝The SW node is with high frequency voltage swing. It should be kept at a small area.❝Place the feedback components as close as possible to the IC and keep away from the noisy devices.❝The GND and AGND should be connected to a strong ground plane for heat sinking and noise protection.©Copyright 2015 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation.12DS2613A-01 July 2015 RT2613AFigure 3. PCB Layout GuideW-Type 16L QFN 3x3 PackageRichtek Technology Corporation14F, No. 8, Tai Yuen 1st Street, Chupei CityHsinchu, Taiwan, R.O.C.Tel: (8863)5526789Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.DS2613A-01 July 13。
Micrel PL663-xx系列分析型频率乘法器说明书
PL663-xx XO FamiliesDESCRIPTIONAnalog Frequency Multipliers TM (AFMs) are the industry’s first “Balanced Oscillator” utilizing analog multiplication of the fundamental frequency (at double or quadruple frequency), combined with an attenuation of the fundamental of the reference crystal, without using a phase-locked loop (PLL), in CMOS technology.Patent pending PL663-xx family of AFM products can achieve up to 800 MHz differential LVPECL, LVDS, or single-ended LVCMOS output with little jitter or phase noise deterioration.PL663-xx family of products utilizes a low-power CMOS technology and is housed in GREEN/ RoHS compliant 16-pin TSSOP and 3x3 QFN packages. FEATURES∙Non-PLL frequency multiplication∙Input frequency from 30-200 MHz∙Output frequency from 60-800 MHz∙Low phase noise and jitter (equivalent to fundamental at the output frequency)∙Ultra-low jittero RMS phase jitter < 0.25 ps (12 kHz to 20 MHz)o RMS period jitter < 2.5 ps typ.∙Low phase noiseo-145dBc/**************************** o-150dBc/***************************∙Low input frequency eliminates the need for expensive crystals∙Differential LVPECL/LVDS, or single-ended LVCMOS output∙Single 2.5V or 3.3V +/- 10% power supply∙Optional industrial temperature range (-40︒C to+85︒C)∙Available in 16-pin GREEN/RoHS compliant TSSOP, and 16-pin 3x3 QFN packages.Figure 1: 2X AFM Phase Noise at 212.5 MHz (106.25 MHz 3rd overtone crystal)PL663-xx XO FamiliesOscillator AmplifierOEQQBAR FrequencyX2XIN XOUTL2X FrequencyX4L4XOnly required in x4 designsRFigure 2: Block Diagram of AFM XOFigure 3 shows the period jitter histogram of the 2x Analog Frequency Multiplier at 212.5 MHz, while Figure 4 shows the very low levels of sub-harmonics that correspond to the exceptional performance (i.e. low jitter).Figure 3: Period Jitter Histogram at 212.5MHz Figure 4: Spectrum Analysis at 212.5MHz Analog Frequency Multiplier (2x), Analog Frequency Multiplier (2x), with 106.25 MHz crystal with sub-harmonics below –69dBcOE LOGIC SELECTIONOUTPUT OESEL OE Output StateLVPECL 0 (Default)0 (Default) Enabled1 Tri-state10 Tri-state1 (Default) EnabledLVDS or LVCMOS 0 (Default)0 Tri-state1 (Default) Enabled10 (Default) Enabled1 Tri-stateOESEL and OE: Connect to V DD to set to “1”, connect to GND to set to “0”. [The ‘Default’ state is set by internal pull up/down resistor.]PL663-xx XO Families PRODUCT SELECTOR GUIDENote: Wavecrest data 10,000 hits. No Filtering was used in Jitter Calculations.Agilent E5500 was used for phase jitter measurements.Spectral specifications were obtained using Agilent E7401A.PL663-xx XO Families BOARD LAYOUT CONSIDERATIONS AND CRYSTAL SPECIFICATIONSBOARD LAYOUT CONSIDERATIONSTo minimize parasitic effects and improve performance, do the following:∙Place the crystal as close as possible to the IC.∙Make the board traces that are connected to the crystal pins symmetrical. The board trace symmetry is very important, as it reduces the negative parasitic effects to produce clean frequency multiplication with low jitter.CRYSTAL SPECIFICATIONSNote: Non-specified parameters can be chosen as standard values from crystal suppliers.CL ratings larger than 5pF require a crystal frequency adjustment.Request detailed crystal specifications from Micrel.PL663-xx XO Families EXTERNAL COMPONENT VALUESINDUCTOR VALUE OPTIMIZATIONThe required inductor value(s) for the best performance depends on the operating frequency, and the board layout specifications. The listed values in this datasheet are based on the calculated parasitic values from Micrel’s evalu ation board design. These inductor values provide the user with a starting point to determine the optimum inductor values. Additional fine-tuning may be required to determine the optimal solution.To assist with the inductor value optimization, Micrel has developed the “AFM Tuning Assistant” software. You can download this software from Micrel’s web site (). The software consists of two worksheets. The first worksheet (named L2) is used to fine-tune the ‘L2’ inductor value, and the second worksheet (named L4) is used for fine tuning of the ‘L4’ (u sed in 4x AFMs only) inductor value.For those designs using Micrel’s recommended board layout, you can use the “AFM Tuning Assistant” to determine the optimum values for the required inductors. This software is developed based on the parasitic information from Micrel’s board layout and can be used to determine the required inductor and parallel capacitor (see LWB1 and Cstray parameters) values. For those employing a different board layout in their design, we recommend to use the parasitic information of their board layout to calculate the optimized inductor values. Please use the following fine tuning procedure:Figure 5: Diagram Representation of the Related System Inductance and Capacitance DIE SIDE PCB side- Cinternal = Based on AFM Device - LWB1 = 2 nH, (2 places), Stray inductance- Cpad = 2.0 pF, Bond pad and its ESD circuitry- Cstray = 1.0 pF, Stray capacitance- C11 = 0.4 pF, The following amplifier stage - L2X = 2x inductor- C2X = range (0.1 to 2.7), Fine tune inductor if usedPL663-xx XO Families ∙There are two default variables that normally will not need to be modified. These are Cpad, and C11 and are found in cells B22 and B27 of ‘AFM Tuning Assistant’, respectively.∙LWB1 is the combined stray inductance in the layout. The DIE wire bond is ~ 0.6 nH and in the case of a leaded part an additional 1.0 nH is added. Your layout inductance must be added to these. There are 2 of these and they are assumed to be approximately symmetrical so you only need to enter this inductance once in cell B23.∙Enter the stray parasitic capacitance into cell B26. An additional 0.5 pF must be added to this value if a leaded part is used.∙Enter the appropriate value for Cinternal into B21 based on the device used (see column D). Use the ‘AFM Tuning Assistant’ soft ware to calculate L2X (and C2X if used) for your resonance frequency.PL663-xx XO Families EXTERNAL COMPONENT VALUES – 3RD OVERTONE RESISTOR SELECTIONS (R3rd)This resistor is only required when a third overtone crystal is used. The chart below indicates the calculated and the nearest “E12” resistor values versus frequency.PL663-xx XO FamiliesELECTRICAL SPECIFICATIONSABSOLUTE MAXIMUM RATINGSExposure of the device under conditions beyond the limits specified by Maximum Ratings for extended periods may cause permanent damage to the device and affect product reliability. These conditions represent a stress rating only, and functional operations of the dev ice at these or any other conditions above the operational limits noted in this specification is not implied.*Note : For performance reasons, some pins on this device do not meet Micrel ’s standard ESD protection. Therefore, the ESD protection on this device is classified as Class I HBM and Class A MM. Handling precaution is recommended.LVPECL ELECTRICAL CHARACTERISTICSOUTPECL Transistion Time Waveformt OUTOUTPECL Levels Test CircuitVDDDUTY CYCLE50%OUTOUTPECL Output Skew 2.0VPL663-xx XO Families LVDS ELECTRICAL CHARACTERISTICSOUTVOUTR = 100ΩLVDS Switching Test CircuitLVDS Levels Test CircuitLVDS Transistion Time WaveformOUTOUTOUT0V (Differential)0V20%80%20%80%t tVR = 1LVDS Transistion Time WaveformOUTOUTVOUTOUTV VLVDS Switching Test CircuitLVDS Levels Test CircuitLVDS Transistion Time WaveformOUTOUT0V (Differential)0V20%80%20%80%t tVPL663-xx XO Families LVCMOS ELECTRICAL CHARACTERISTICSPL663-xx XO Families BOARD DESIGN AND LAYOUT CONSIDERATIONSL2X: Reduce the PCB trace inductance to a minimum by placing L2X as physically close to their respective pins as possible. Also be sure to bypass each V DD connection especially taking care to place a 0.01 uF bypass at the V DD side of L2X (see recommended layout).Crystal Connections: Be sure to keep the ground plane under the crystal connections continuous so that the stray capacitace is consistent on both crystal connections. Also be sure to keep the crystal connections symmetrical with respect to one another and the crystal connection pins of the IC. If you chose to use a series capacitance and/or inductor to fine tune the crystal frequency, be sure to put symmetrical pads for this cap on both crystal pins (see Cadj in recommended layout), even if one of the capacitors will be a 0.01 uF and the other is used to tune the frequency. To further maintain a symmetrical balance on a crystal that may have more internal Cstray on one pin or the other, place capacitor pads (Cbal) on each crystal lead to ground (see recommended layout). R3rd is only required if a 3rd overtone crystal is used.V DD and GND: Bypass VDDANA and VDDBUF with separate bypass capacitors and if a V DD plane is used, feed each bypass cap with its own via. Be sure to connect any ground pin including the bypass caps with short via connection to the ground plane.OESEL: J1 is recommended so the same PCB layout can be used for both OESEL settings.PL663 (2x AFM) TSSOP LayoutPL663-xx XO FamiliesPACKAGE PIN DESCRIPTION AND ASSIGNMENTGNDOSCXIN DNC GNDANAL2X VDDOSC VDDANA OESEL VDDBUF QBAR QGNDBUFDNC DNC OEXOUTG N D O S CQX I NGNDANA DNC OESEL G N D B U FV D D B U FQ B A RL2XVDDOSCVDDANA D N CD N C2x AFM Package Pin OutPIN ASSIGNMENTSPL663-xx XO Families PACKAGE INFORMATION16 PIN TSSOP16 PIN 3x3 QFNPL663-xx XO Families ORDERING INFORMATION (GREEN PACKAGE COMPLIANT)Micrel Inc., reserves the right to make changes in its products or specifications, or both at any time without notice. The information furnished by Micrel is believed to be accurate and reliable. However, Micrel makes no guarantee or warranty concerning the accuracy of said information and shall not be responsible for any loss or damage of whatever nature resulting from the use of, or reliance upon this product.LIFE SUPPORT POLICY: Micrel’s products are not authorized for use as critical components in life support devices or systems without the express written approval of the President of Micrel Inc.。
射频与通信集成电路-振荡器
概述 振荡器基本原理 环形振荡器 LC振荡器 振荡器的干扰和相位噪声 相位噪声带来的问题 正交信号的产生 振荡器优化设计
概述
» 振荡器(oscillator)是将直流电源能量转换成交流能量的电路。 » 振荡器必须有正反馈和足够的增益以克服反馈路径上的损耗,同时还需要
Psig
10lg f
dBm
87 2.64 10 lg 91103
1MHz
133.9 dBc / Hz
振荡器的干扰和相位噪声
‒ 由相位噪声测试曲线可以直接读出不同频率偏移下的相位噪声。
f
L() (dBc/Hz)
10 kHz -89.51
100kHz -112.41
1MHz -133.97
或
L
Pn
dBm
Psig
10lgf
dBm
‒ 已知:分辨率带宽为Res BW =
91kHz , 振 荡 频 率 为 1.14136GHz ,
振荡信号功率为-2.64dBm,在偏
移振荡频率1MHz处的噪声功率
约为-87dBm ,计算在偏移振荡
频率1MHz处的相位噪声为
-87dBm
L
Pn
dBm
I0 gmc
f
x
I05 x
3rs02C1C2
gm0 g mc
2
9 2
1
2x
2
2
x
2
cos 1 x 3x
1
x2
1 x2 3x cos 1 x2
其中
f x gmcVm
I0
x Vth VB , 1 x 1 Vm
振荡幅度正比于偏臵电流,反比于临界跨导。
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a r X i v :q -a l g /9609028v 1 23 S e p 1996MRC-PH-TH-11-96q-alg/9609028A q–oscillator Green FunctionH.Ahmedov ∗and I.H.Duru ∗†∗-TUBITAK -Marmara Research Centre,Research Institute for Basic Siences,Department of Physics,P.O.Box 21,41470Gebze,Turkey†-Trakya University,Mathematics Department,P.O.Box 126,Edirne,Turkey.AbstractBy using the generating function formula for the product of two q-Hermite polynomials q-deformation of the Feynman Green function for the harmonic oscillator is obtained.PACS numbers:03.65.Fd and 02.20.-aSeptember 199611Introductionq-oscillators are the most extensively studied deformed dynamical systems. They have been presented in severent different types[1].The e-functions of these q-oscillators are expressible in terms of either the discrete or the continuous q-Hermite polynomials[2].Although the literature on q-oscillators is very rich,the corresponding q-Green functions have not been investigated.This looks suprising when one considers the fact that the exact closed form of the Green function of the undeformed oscillator has been known for many decades[3].It is the purpose of this note to obtain the Green function for one of the q-oscillators.The q-oscillator we deal with is the one which is solved in terms of the continuous q−1-Hermite polynomials[4].In Section II we briefly review the q-oscillator realization of Ref.4.Section III is devoted to the derivation of the q-oscillator Green function which is the deformation of the well known Feynman formula.The method of the calculation of the non-trivial q→1limit,which is essential for arriving at the usual Feynman Green function is outlined in the Appendix.2A q-oscillator realizationRecently Atakishiev,Frank and Wolf introduced a simple difference real-ization of the Heysenberg q-algebra[4].They also studied the correspond-ing q-oscillator Hamiltonian and its e-functions in terms of the q−1-Hermite polynomials.The q-annihilation and creation operators acting on the smooth functions f(ξ)withξ∈(−∞,∞)are given byb q=12β∂ξ)−qβξexp(−121/2q1/4βv(ξ)(q−βξexp(−12β∂ξ))v(ξ)(2)wherek=−log q,β=1(cosh(kβξ))1/2(3)andξis the dimensionless variable(with¯h=1)2ξ=√21/2(ξ+d21/2(ξ−df(ξ)g(ξ)(8)satisfies the e-value equationH qΨq n(ξ)=[n]Ψq n(ξ).(9) Here[n]is defined as usual as[n]=1−q nπ(1−q))1/4q(n+1/2)2/42)1/2ξ):3lim q→1−(22])Φq n(ξ,t)=ζD qζΦq n(ξ,t).(14)Hereζis the exponantial time parameter given byζ=exp(−iωt)(15) andΦq n(ξ,t)is the time dependent wave function:Φq n(ξ,t)=exp(−iω(n+1/2))Ψq n(ξ)=ζn+1/2Ψq n(ξ).(16) The action of the q-derivative on the time dependent factor of the above wave functionζD qζζn+1/2=[n+1/2]ζn+1/2=(q1/2[n]+[1ζζ′=exp(−iω(t′−t)).(19)4To execute the summation over n in(18)we recall the following generating function formula for the product of two continuous q-Hermite polynomials [5]:(z2;q)∞(z exp(−i(θ−φ));q)∞=∞n=0z nH n(cosθ|q)H n(cosφ|q)(1−z2)1/2exp[−12n nH n(ξ)H n(ξ′)(22)withξ=(1−q2)1/2cosφ.(23)Note that by the help of(22)one can derive the well known Feynman formula for the undeformed oscillator[3](with T=t′−t)K(ξ,ξ′;t′−t)=(mω2sin(ωT)((ξ2+ξ′2)cos(ωT)−2ξξ′)](24) from the Green function written in the wave function decomposition form [7].5To derive the q-oscillator Green function we first insert q −1in place of q by recalling the relation [5]1(q ;q )n.(25)After making the required analytic continuations in θand φwe arrive atE −1q (qz 21−q exp −(θ+φ))E q (−qz 1−qexp −(θ−φ))E q (qz(q ;q )nq (n +1/2)2/2z n(26)which is the formula suitable to our q −1-Hermite polynomials.The q-ex-ponantials employed in the above equation are qiven in terms of the n →∞limit of the q-factorials as [5]E 1/q (−x )=E −1q (x )=((1−q )x ;q )∞.(27)When we introduce the formula of (26)into (18)we obtain the final formof the q-oscillator Green function:K q (ξ,ξ′;z )=q −1/8(2k 1−q )E q (−qz 1−q exp(kβ(ξ+ξ′))E q (qz1−qexp(kβ(ξ−ξ′))(28)By the process sketched in the Appendix the above equation is reduced to the Feynman formula of (24)in q →1−limit.In T →0(z →1)limit we distinguish two cases :(i)For ξ=ξ′by the virtue of the first exponantialE −1q (qz 2we havelimT→0K q(ξ,ξ′;z)=0.(30) (ii)Forξ=ξ′on the other hand by the virtue of the1st and3th1−q)E q(−qz1−q)=limz→1∞n=0(1−q n z2)1−z(32)type.It is easy to conclude then that the Green function(28)behaves as the δ-functionδ(exp(kβξ)−exp(kβξ‘))in T→0(z→1)limit.Acknowledgement.We thank¨O.F.Dayi for reading the manuscript.APPENDIXUsing the definitions in(21)and(23)we can rewrite(20)as∞j=0(1−z2q j)(1−z2q2j)−2(1−(1−q)A j)−1=∞ n=0z n2)1/2ξ|q)H n((1−q(1−z2q2j)2(A.2)q→1−limit of the right hand side(r.h.s)of(A.1)(with(q;q)n=(1−q)n[n]) is7lim q→1−(r.h.s)=∞n=0z nk(1+q k)=−log(1−z2)1/2q2(A.5)with[5](1−z)a q=ψ1,0(q−a,q;z)(A.6) Then(l.h.s.)can be rewritten as(l.h.s.)=1where q i is the sequence:lim i→∞q i=1−.By the virtue of(A.9)the functions (1−q i)|A j|satisfy the condition(A.10)too.Thus the logarithm function (A.9)can be expanded in the Taylor series in q→1−limit aslim q→1−F(ξ,ξ′,z|q)=limq→1−∞k=1(1−q)k1−z2(A.13)¿From(A.8)and(A.14)we getlim q→1−(l.h.s.)=11−z2](A.14)which together with(A.3)establishes the desired limit of(22).9References[1]A.J.Macfarlane,J.Phys.A22,983(1989);E.V.Daskinsky and P.P.Kulish,Zap.Nauchn.Sem.LOMI189,13(1993).,[2]R.Askey and M.Ismail in“Studies in Pure Mathematics”P.Erd¨o sed.,Birkh¨a user,Basel,1983;and R.Askey“q-Series and Partitions”D.Stanton ed.,Springer,New York,1989.[3]R.P.Feynman and A.R.Hibbs,Quantum Mechanics and Path Integrals,Mc Graw-Hill,New-york,1965.[4]M.M.Atakishiev,A.Frank and K.B.Wolf J.Math.Phys.25,401(1982).10。