开关电源设计白皮书
《开关电源设计（第三版）》
《开关电源设计（第三版）》
佚名
【期刊名称】《电源技术》
【年(卷),期】2016(40)4
【摘要】本书为二十几年来世界公认最权威的电源的设计指导著作《开关电源设计》的再版（第三版）。
书中系统地论述了开关电源最常用拓扑的基本原理、磁性元件的设计原则及闭环反馈稳定性和驱动保护等。
本书在讲述的过程中应用教学式、How＆Why方法，讨论时结合了大量设计实例、设计方程和图表。
本书同时涵盖了开关电源技术、材料和器件的最新发展等内容。
【总页数】1页(P894-894)
【关键词】开关电源设计;第三版;开关电源技术;驱动保护;闭环反馈;磁性元件;应用
教学;设计实例
【正文语种】中文
【中图分类】TN86
【相关文献】
1.《开关电源设计(第三版)》 [J],
2.《开关电源设计(第三版)》 [J],
3.《开关电源设计(第三版)》 [J],
4.《开关电源设计(第三版)》 [J],
5.《开关电源设计(第三版)》 [J],
因版权原因，仅展示原文概要，查看原文内容请购买。
开关电源的设计资料
3.1.2
ＴＬ４３１反馈电路
Ａ．由上图可知,在ＴＬ４３１的反馈电路中, 电路的输出公差一般为±１％，负载调整 率为±０．２％，线性调整率为±０．２ ％． Ｂ．调整ＴＬ４３１参考端的电阻值，得到 正确的输出电压值，由Ｖref=2.5V得出， Ｒ＝（ＶＯ－２．５）＊１０Ｋ／２．５， 其中ＶＯ为输出电压值，也可以调整 １０Ｋ的值，使Ｒ也为常用档位的电阻 Ｃ．串联在光藕上方的电阻也需要根据光藕 和开关芯片控制脚电流的大小调到合适的 值
• 3.9 开关变压器的设计 • 3.9.1 根据下表选择开关变压器磁芯,框架.
• 3.9.2 开关变压器磁芯的选择也可以用以下公式计算 Ap=Ae*Aq=Pt*10 /2ŋ*fs*Bm*δ*Km*Kc • 3.9.3 磁芯的磁滞回线如图,
6
• • • • •
磁芯的选择尽量遵循以下原则 1) 磁导率尽量高 2)矫顽力Hc尽量小,磁滞回线尽量窄 3)电阻率要高,涡流损耗小 4)饱和磁感应强度要高
• 3.9.12 次级绕组和辅助绕组峰值电压的计 算
• 3.10 输出整流二极管的选择 • 1) 原则: • 常用整流二极管型号如下表
• 3.11 输出电容的选择 • 原则 • 1) 在105℃时,必须大于或等于Iripple • 2)必须有小的ESR的电解电容.一般采用两个并联 的方式减小ESR. • • 3.12 输出滤波电感量的选择 • 原则 2.2μH≥L≤6.8μH,当输出电流<1A时,采用 铁氧体磁珠.>1A时,采用绕线电感,且大小随电流的 增大而增大.
• • • •
3.2.2钳位二极管选取 钳位二极管电压Vclo一般取1.5Vor. 钳位二极管过冲电压Vclm一般取1.4Vclo. 具体参数选择如下图
数据中心供配电系统白皮书[1]
![数据中心供配电系统白皮书[1]](https://img.taocdn.com/s3/m/66d25f0828ea81c759f57831.png)
数据中心供配电系统应用白皮书一引言任何现代化的IT设备都离不开电源系统，数据中心供配电系统是为机房内所有需要动力电源的设备提供稳定、可靠的动力电源支持的系统。
供配电系统于整个数据中心系统来说有如人体的心脏-血液系统。
1.1 编制范围考虑到数据中心供配电系统内容的复杂性和多样性以及叙述的方便，本白皮书所阐述的“数据中心供配电系统”是从电源线路进用户起经过高/低压供配电设备到负载止的整个电路系统，将主要包括：高压变配电系统、柴油发电机系统、自动转换开关系统（ATSE，Automatic Transfer Switching Equipment）、输入低压配电系统、不间断电源系统（UPS，Uninterruptible Power System）系统、UPS列头配电系统和机架配电系统、电气照明、防雷及接地系统。
如下图：图1 数据中心供配电系统示意方框图高压变配电系统：主要是将市电（6kV/10kV/35kV，3相）市电通过该变压器转换成（380V/400V，3相），供后级低压设备用电。
柴油发电机系统：主要是作为后备电源，一旦市电失电，迅速启动为后级低压设备提供备用电源。
自动转换开关系统：主要是自动完成市电与市电或者市电与柴油发电机之间的备用切换。
输入低压配电系统：主要作用是电能分配，将前级的电能按照要求、标准与规范分配给各种类型的用电设备，如UPS、空调、照明设备等。
UPS系统：主要作用是电能净化、电能后备，为IT负载提供纯净、可靠的用电保护。
UPS输出列头配电系统：主要作用是UPS输出电能分配，将电能按照要求与标准分配给各种类型的IT设备。
机架配电系统：主要作用是机架内的电能分配。
此外，数据中心的供配电系统负责为空调系统、照明系统及其他系统提供电能的分配与输入，从而保证数据中心正常运营。
电气照明：包括一般要求，照明方案、光源及灯具选择。
防雷及接地系统：包括数据中心防雷与接地的一般要求与具体措施。
1.2 编制依据《电子信息系统机房设计规范》GB 50174—2008《电子信息机房施工及检验规范》GB50462—20081.3 编制原则1．具有适应性、覆盖性、全面性的特征。
关于开关电源的书
关于开关电源的书
以下是一些关于开关电源的推荐书籍：
1. 《开关电源设计工程师宝典》 - 张洪瑾, 丁寿楠, 耿聪, 胡思远, 郑建等著。
这本书是国内开关电源设计领域的经典著作，详细介绍了开关电源的基础知识、设计流程、设计方法和工程实例。
2. 《开关电源设计指南》 - 狄可和, 陈若风, 李航等著。
该书是一本系统介绍开关电源设计理论和实践的综合参考书，内容涵盖了开关电源的基本原理、拓扑结构、器件选择和设计技巧等方面。
3. 《Switchmode Power Supply Handbook》 - Keith Billings, Taylor Morey等著。
这本书是国际上广受欢迎的开关电源参考书之一，适用于初学者和专业人士。
内容包括开关电源拓扑、控制方法、滤波和EMI抑制等关键主题。
4. 《Switching Power Supply Design and Optimization》 - Sanjaya Maniktala著。
该书重点介绍了开关电源的设计和优化技巧，强调了高效率和高性能的关键技术。
书中还提供了大量的设计示例和案例分析。
这些书籍都可以帮助你深入了解开关电源的原理、设计方法和实践经验，选择一本符合自己需求的进行阅读将有助于你的学习和工作。
第六章内容
6Converter CircuitsWe have already analyzed the operation of a number of different types of converters: buck, boost,buck-boost, Cuk, voltage-source inverter, etc. With these converters, a number of different functionscan be performed: step-down of voltage, step-up, inversion of polarity, and conversion of dc to ac orvice versa.It is natural to ask, Where do these converters come from? What other converters occur, and whatother functions can be obtained? What are the basic relations between converters? In this chapter,several different circuit manipulations are explored, which explain the origins of the basic converters.Inversion of source and load transforms the buck converter into the boost converter. Cascadeconnection of converters, and simplification of the resulting circuit, shows how the buck-boost andCuk converters are based on the buck and the boost converters. Differential connection of the loadbetween the outputs of two or more converters leads to a single-phase or polyphase inverter. A shortlist of some of the better known converter circuits follows this discussion.Transformer-isolated dc-dc converters are also covered in this chapter. Use of a transformer allowsisolation and multiple outputs to be obtained in a dc-dc converter, and can lead to better converter optimization when a very large or very small conversion ratio is required. The transformer is modeledas a magnetizing inductance in parallel with an ideal transformer; this allows the analysis techniquesof the previous chapters to be extended to cover converters containing transformers. A number ofwell-known isolated converters, based on the buck, boost, buck-boost, single-ended primaryinductance converter (SEPIC), and Cuk, are listed and discussed.Finally, the evaluation, selection, and design of converters to meet given requirements areconsidered. Important performance-related attributes of transformer-isolated converters include:whether the transformer reset process imposes excessive voltage stress on the transistors, w hether theconverter can supply a high-current output without imposing excessive current stresses on thesecondary-side components, and whether the converter can be well-optimized to operate with a widerange of operating points, that is, with large tolerances in V g and P load. Switch utilization is asimplified figure-of-merit that measures the ratio of the converter output power to the total transistorvoltage and current stress. As the switch utilization increases, the converter efficiency increases whileits cost decreases. Isolated converters with large variations in operating point tend to utilize theirpower devices more poorly than nonisolated converters which function at a single operating point.Computer spreadsheets are a good tool for optimization of power stage designs and for trade studiesto select a converter topology for a given application.6.1 CIRCUIT MANIPULATIONSThe buck converter (Fig. 6.1) was developed in Chapter 1 using basic principles. The switch reducesthe voltage dc component, and the low-pass filter removes the switching harmonics. In the continuousconduction mode, the buck converter has a conversion ratio of M =D. The buck converter is thesimplest and most basic circuit, from which we will derive other converters.Fig. 6.1. The basic buck converter.6.1.1 Inversion of Source and LoadLet us consider first what happens when we interchange the power input and power output ports of a converter. In the buck converter of Fig. 6.2(a), voltage V 1 is applied at port 1, and voltage V 2 appears at port 2. We know thatV 2 = DV 1 (6.1)This equation can be derived using the principle of inductor volt-second balance, with the assumption that the converter operates in the continuous conduction mode. Provided that the switch is realized such that this assumption holds, then Eq. (6.1) is true regardless of the direction of power flow.Fig. 6.2. Inversion of source and load transforms a buck converter into a boost converter: (a) buck converter, (b) inversion of source and load, (c) realization of switch.So let us interchange the power source and load, as in Fig. 6.2(b). The load, bypassed by the capacitor, is connected to converter port 1, while the power source is connected to converter port 2. Power now flows in the opposite direction through the converter. Equation (6.1) must still hold; by solving for the load voltage V 1, one obtains211V DV =(6.2) So the load voltage is greater than the source voltage. Figure 6.2(b) is a boost converter, drawn backwards. Equation (6.2) nearly coincides with the familiar boost converter result, M (D )= 1/D ', except that D ' is replaced by D .Since power flows in the opposite direction, the standard buck converter unidirectional switch realization cannot be used with the circuit of Fig. 6.2(b). By following the discussion of Chapter 4, one finds that the switch can be realized by connecting a transistor between the inductor and ground, and a diode from the inductor to the load, as shown in Fig. 6.2(c). In consequence, the transistor duty cycle D becomes the fraction of time which the single-pole double-throw (SPDT) switch of Fig. 6.2(b)spends in position 2, rather than in position 1. So we should interchange D with its complement D ' in Eq. (6.2), and the conversion ratio of the converter of Fig. 6.2(c) is21'1V D V =(6.3) Thus, the boost converter can be viewed as a buck converter having the source and load connections exchanged, and in which the switch is realized in a manner that allows reversal of the direction of power flow.Fig. 6.3. Cascade connection of converters.6.1.2 Cascade Connection of Converters Converters can also be connected in cascade, as illustrated in Fig. 6.3 [1,2]. Converter 1 has conversion ratio M 1(D ), such that its output voltage V 1 isg V D M V )(11= (6.4)This voltage is applied to the input of the second converter. Let us assume that converter 2 is driven with the same duty cycle D applied to converter 1. If converter 2 has conversion ratio M 2(D ), then the output voltage V is12)(V D M V = (6.5) Substitution of Eq. (6.4) into Eq. (6.5) yields)()()(21D M D M D M V V g== (6.6) Hence, the conversion ratio M (D ) of the composite converter is the product of the individual conversion ratios M 1(D ) and M 2(D ).Fig. 6.4. Cascade connection of buck converter and boost converter.Let us consider the case where converter 1 is a buck converter, and converter 2 is a boost converter. The resulting circuit is illustrated in Fig. 6.4. The buck converter has conversion ratioD V V g=1 (6.7) The boost converter has conversion ratioDV −11So the composite conversion ratio isDD V V g −=1 (6.9) The composite converter has a noninverting buck-boost conversion ratio. The voltage is reduced whenD < 0.5, and increased when D > 0.5.Fig. 6.5. Simplification of the cascaded buck and boost converter circuit of Fig. 6.4: (a) removal of capacitor C 1, (b) combining of inductors L 1 and L 2.The circuit of Fig. 6.4 can be simplified considerably. Note that inductors L 1 and L 2, along with capacitor C 1, form a three-pole low-pass filter. The conversion ratio does not depend on the number of poles present in the low-pass filter, and so the same steady-state output voltage should be obtained when a simpler low-pass filter is used. In Fig. 6.5(a), capacitor C 1 is removed. Inductors L 1 and L 2 are now in series, and can be combined into a single inductor as shown in Fig. 6.5(b). This converter, the noninverting buck-boost converter, continues to exhibit the conversion ratio given in Eq. (6.9).Fig. 6.6. Connections of the circuit of Fig. 6.5(b): (a) while the switches are in position 1, (b) while the switches are in position 2.The switches of the converter of Fig. 6.5(b) can also be simplified, leading to a negative output voltage. When the switches are in position 1, the converter reduces to Fig. 6.6(a). The inductor is connected to the input source V g and energy is transferred from the source to the inductor. When the switches are in position 2, the converter reduces to Fig. 6.6(b). The inductor is then connected to the load, and energy is transferred from the inductor to the load. To obtain a negative output, we can simply reverse the polarity of the inductor during one of the subintervals (say, while the switches are in position 2). The individual circuits of Fig. 6.7 are then obtained, and the conversion ratio becomesDV g −1Note that one side of the inductor is now always connected to ground, while the other side is switched between the input source and the load. Hence only one SPDT switch is needed, and the converter circuit of Fig. 6.8 is obtained. Figure 6.8 is recognized as the conventional buck-boost converter.Fig. 6.7. Reversal of the output voltage polarity, by reversing the inductor connections while the switches are in position 2: (a) connections with the switches in position 1, (b) connections with the switches in position 2.Fig. 6.8. Converter circuit obtained from the subcircuits of Fig. 6.7.Thus, the buck-boost converter can be viewed as a cascade connection of buck and boost converters. The properties of the buck-boost converter are consistent with this viewpoint. Indeed, the equivalent circuit model of the buck-boost converter contains a 1:D (buck) dc transformer, followed by a D ':1 (boost) dc transformer. The buck-boost converter inherits the pulsating input current of the buck converter, and the pulsating output current of the boost converter.Other converters can be derived by cascade connections. The Cuk converter (Fig. 2.20) was originally derived [1,2] by cascading a boost converter (converter 1), followed by a buck (converter 2).A negative output voltage is obtained by reversing the polarity of the internal capacitor connection during one of the subintervals; as in the buck-boost converter, this operation has the additional benefit of reducing the number of switches. The equivalent circuit model of the Cuk converter contains a D ':1 (boost) ideal dc transformer, followed by a 1:D (buck) ideal dc transformer. The Cuk converter inherits the nonpulsating input current property of the boost converter, and the nonpulsating output current property of the buck converter.6.1.3 Rotation of Three-Terminal CellThe buck, boost, and buck-boost converters each contain an inductor that is connected to a SPDT switch. As illustrated in Fig. 6.9(a), the inductor-switch network can be viewed as a basic cell having the three terminals labeled a , b , and c . It was first pointed out in [1,2], and later in [3], that there are three distinct ways to connect this cell between the source and load. The connections a-A b-B c-C lead to the buck converter. The connections a-C b-A c-B amount to inversion of the source and load, and lead to the boost converter. The connections a-A b-C c-B lead to the buck-boost converter. So the buck, boost, and buck-boost converters could be viewed as being based on the same inductor-switch cell, with different source and load connections.Fig. 6.9. Rotation of three-terminal switch cells: (a) switch/inductor cell, (b) switch/capacitor cell.A dual three-terminal network, consisting of a capacitor-switch cell, is illustrated in Fig. 6.9(b). Filter inductors are connected in series with the source and load, such that the converter input and output currents are nonpulsating. There are again three possible ways to connect this cell between the source and load. The connections a-A b-B c-C lead to a buck converter with L-C input low-pass filter. The connections a-B b-A c-C coincide with inversion of source and load, and lead to a boost converter with an added output L-C filter section. The connections a-A b-C c-B lead to the Cuk converter. Rotation of more complicated three-terminal cells is explored in [4].6.1.4 Differential Connection of the LoadIn inverter applications, where an ac output is required, a converter is needed that is capable of producing an output voltage of either polarity. By variation of the duty cycle in the correct manner, a sinusoidal output voltage having no dc bias can then be obtained. Of the converters studied so far in this chapter, the buck and the boost can produce only a positive unipolar output voltage, while the buck-boost and Cuk converter produce only a negative unipolar output voltage. How can we derive converters that can produce bipolar output voltages?Fig. 6.10. Obtaining a bipolar output by differential connection of load.A well-known technique for obtaining a bipolar output is the differential connection of the load across the outputs of two known converters, as illustrated in Fig. 6.10. If converter 1 produces voltagedc source V 1, and converter 2 produces voltage V 2, then the load voltage V is given by21V V V −= (6.11)Although V 1 and V 2 may both individually be positive, the load voltage V can be either positive or negative. Typically, if converter 1 is driven with duty cycle D , then converter 2 is driven with its complement, D ', so that when V 1 increases, V 2 decreases, and vice versa.Several well-known inverter circuits can be derived using the differential connection. Let’s realize converters 1 and 2 of Fig. 6.10 using buck converters. Figure 6.11(a) is obtained. Converter 1 is driven with duty cycle D , while converter 2 is driven with duty cycle D '. So when the SPDT switch of converter 1 is in the upper position, then the SPDT switch of converter 2 is in the lower position, and vice versa. Converter 1 then produces output voltage V 1 = DV g , while converter 2 produces output voltage V 2 = D'V g . The differential load voltage isg g V D DV V '−= (6.12)Simplification leads tog V D V )12(−= (6.13)This equation is plotted in Fig. 6.12. It can be seen the output voltage is positive for D > 0.5, and negative for D < 0.5. If the duty cycle is varied sinusoidally about a quiescent operating point of 0.5, then the output voltage will be sinusoidal, with no dc bias.The circuit of Fig. 6.11 (a) can be simplified. It is usually desired to bypass the load directly with a capacitor, as in Fig. 6.11(b). The two inductors are now effectively in series, and can be combined into a single inductor as in Fig. 6.11(c). Figure 6.11(d) is identical to Fig. 6.11(c), but is redrawn for clarity. This circuit is commonly called the H-bridge , or bridge inverter circuit. Its use is widespread in servo amplifiers and single-phase inverters. Its properties are similar to those of the buck converter, from which it is derived.Fig. 6.11. Derivation of bridge inverter (H-bridge): (a) differential connection of load across outputs of buck converters, (b) bypassing load by capacitor, (c) combining series inductors, (d) circuit (c) redrawn in its usual form.Fig. 6.12. Conversion ratio of the H-bridge inverter circuit.Fig. 6.13. Generation of dc-3Фac inverter by differential connection of 3Ф load.Polyphase inverter circuits can be derived in a similar manner. A three-phase load can be connected differentially across the outputs of three dc-dc converters, as illustrated in Fig. 6.13. If the three-phase load is balanced, then the neutral voltage V n will be equal to the average of the three converter output voltages:)(21321V V V V n ++= (6.14) If the converter output voltages V 1, V 2, and V 3 contain the same dc bias, then this dc bias will also appear at the neutral point V n . The phase voltages V an , V bn and V cn are given by ncn n bn nan V V V V V V V V V −=−=−=321 (6.15)It can be seen that the dc biases cancel out, and do not appear in V an , V bn , and V cn .Let us realize converters 1, 2, and 3 of Fig. 6.13 using buck converters. Figure 6.14(a) is then obtained. The circuit is redrawn in Fig. 6.l4(b) for clarity. This converter is known by several names, including the voltage-source inverter and the buck-derived three-phase bridge .Inverter circuits based on dc-dc converters other than the buck converter can be derived in a similar manner. Figure 6.14(c) contains a three-phase current-fed bridge converter having a boost-type voltage conversion ratio. Since most inverter applications require the capability to reduce the voltage magnitude, a dc-dc buck converter is usually casacded at the current-fed bridge dc input port. Several other examples of three-phase inverters are given in [5-7], in which the converters are capable of both increasing and decreasing the voltage magnitude.Fig. 6.14. Derivation of dc-3Фac inverters: (a) differential connection of 3Ф load across outputs of buck converters; (b) simplification of low-pass filters to obtain the dc-3Фac voltage-source inverter; (c) the dc-3Фac current-source inverter.Fig. 6.14. Continued.6.2 A SHORT LIST OF CONVERTERSAn infinite number of converters are possible, and hence it is not feasible to list them all here. A short list is given here.Let’s consider first the class of single-input single-output converters, containing a single inductor. There are a limited number of ways in which the inductor can be connected between the source and load. If we assume that the switching period is divided into two subintervals, then the inductor should be connected to the source and load in one manner during the first subinterval, and in a different manner during the second subinterval. One can examine all of the possible combinations, to derive the complete set of converters in this class [8-10]. By elimination of redundant and degenerate circuits, one finds that there are eight converters, listed in Fig. 6.15. How the converters are counted can actually be a matter of semantics and personal preference; for example, many people in the field would not consider the noninverting buck-boost converter as distinct from the inverting buck-boost. Nonetheless, it can be said that a converter is defined by the connections between its reactive elements, switches, source, and load; by how the switches are realized; and by the numerical range of reactive element values.Fig. 6.15. Eight members of the basic class of single-input single-output converters containing a single inductor.Fig. 6.15. Continued.The first four converters of Fig. 6.15, the buck, boost, buck-boost, and the noninverting buck-boost, have been previously discussed. These converters produce a unipolar dc output voltage.Converters 5 and 6 are capable of producing a bipolar output voltage. Converter 5, the H-bridge, has previously been discussed. Converter 6 is a nonisolated version of a push-pull current-fed converter [11-15]. This converter can also produce a bipolar output voltage; however, its conversion ratio M(D) is a nonlinear function of duty cycle. The number of switch elements can be reduced by using a two-winding inductor as shown. The function of the inductor is similar to that of the flyback converter, discussed in the next section. When switch 1 is closed the upper winding is used, while when switch 2 is closed, current flows through the lower winding. The current flows through only one winding at any given instant, and the total ampere-turns of the two windings are a continuous function of time. Advantages of this converter are its ground-referenced load and its ability to produce a bipolar output voltage using only two SPST current-bidirectional switches. The isolated version and its variants have found application in high-voltage dc power supplies.Converters 7 and 8 can be derived as the inverses of converters 5 and 6. These converters are capable of interfacing an ac input to a dc output. The ac input current waveform can have arbitrary waveshape and power factor.The class of single-input single-output converters containing two inductors is much larger. Several of its members are listed in Fig. 6.16. The Cuk converter has been previously discussed and analyzed. It has an inverting buck-boost characteristic. The SEPIC (single-ended primary inductance converter) [16], and its inverse, have noninverting buck-boost characteristics. Two-inductor converters having conversion ratios M(D) that are biquadratic functions of the duty cycle D are also numerous. An example is converter 4 of Fig. 6.16 [17]. This converter can be realized using a single transistor and three diodes. Its conversion ratio is M(D) =D2. This converter may find use in nonisolated applications that require a large step-down of the dc voltage, or in applications having wide variationsin operating point.Fig. 6.16. Several members of the basic class of single-input single-output converters containing two inductors.6.3 TRANSFORMER ISOLATIONIn a large number of applications, it is desired to incorporate a transformer into a switching converter, to obtain dc isolation between the converter input and output. For example, in off-line applications (where the converter input is connected to the ac utility system ), isolation is usually required by regulatory agencies. Isolation could be obtained in these cases by simply connecting a 50 Hz or 60 Hz transformer at the converter ac input. However, since transformer size and weight vary inversely with frequency, significant improvements can be made by incorporating the transformer into the converter, so that the transformer operates at the converter switching frequency of tens or hundreds of kilohertz. When a large step-up or step-down conversion ratio is required, the use of a transformer can allow better converter optimization. By proper choice of the transformer turns ratio, the voltage or current stresses imposed on the transistors and diodes can be minimized, leading to improved efficiency and lower cost.Multiple dc outputs can also be obtained in an inexpensive manner, by adding multiple secondary windings and converter secondary-side circuits. The secondary turns ratios are chosen to obtain the desired output voltages. Usually only one output voltage can be regulated via control of the converter duty cycle, so wider tolerances must be allowed for the auxiliary output voltages. Cross-regularion is a measure of the variation in an auxiliary output voltage, given that the main output voltage is perfectly regulated [18-20].A physical multiple-winding transformer having turns ratio n 1:n 2:n 3:... is illustrated in Fig. 6.17(a).A simple equivalent circuit is illustrated in Fig. 6.17(b), which is sufficient for understanding the operation of most transformer-isolated converters. The model assumes perfect coupling between windings and neglects losses; more accurate models are discussed in a later chapter. The ideal transformer obeys the relations...)()()(0...)()()(332211332211+++====t i n t i n t i n n t v n t v n t v(6.16)In parallel with the ideal transformer is an inductance L M, called the magnetizing inductance, referred to the transformer primary in the figure.Fig. 6.17. Simplified model of a multiple-winding transformer (a) schematic symbol, (b) equivalent circuit containing a magnetizing inductance and ideal transformer.Fig. 6.18. B-H characteristics of transformer core.Physical transformers must contain a magnetizing inductance. For example, suppose we disconnect all windings except for the primary winding. We are then left with a single winding on a magnetic core - an inductor. Indeed, the equivalent circuit of Fig. 6.17(b) predicts this behavior, via the magnetizing inductance.The magnetizing current i M(t) is proportional to the magnetic field H(t) inside the transformer core. The physical B-H characteristics of the transformer core material, illustrated in Fig. 6.18, govern the magnetizing current behavior. For example, if the magnetizing current i M(t) becomes too large, then the magnitude of the magnetic field H(t) causes the core to saturate. The magnetizing inductance then becomes very small in value, effectively shorting out the transformer.The presence of the magnetizing inductance explains why transformers do not work in dc circuits: at dc, the magnetizing inductance has zero impedance, and shorts out the windings. In a well-designed transformer, the impedance of the magnetizing inductance is large in magnitude over the intended range of operating frequencies, such that the magnetizing current i M(t) has much smaller magnitudethan i l (t ). Then i l ’(t ) ≈ i l (t ), and the transformer behaves nearly as an ideal transformer. It should be emphasized that the magnetizing current i M (t ) and the primary winding current i l (t ) are independent quantities.The magnetizing inductance must obey all of the usual rules for inductors. In the model of Fig.6.17(b), the primary winding voltage v l (t ) is applied across L M , and hencedtt di L t v M Ml )()(= (6.17) Integration leads to ∫=−tl M M M d v L i t i 0)(1)0()(ττ (6.18) So the magnetizing current is determined by the integral of the applied winding voltage. The principle of inductor volt-second balance also applies: when the converter operates in steady-state, the dc component of voltage applied to the magnetizing inductance must be zero:∫=s T l s dt t v T 0)(10 (6.19)Since the magnetizing current is proportional to the integral of the applied winding voltage, it is important that the dc component of this voltage be zero. Otherwise, during each switching period there will be a net increase in magnetizing current, eventually leading to excessively large currents and transformer saturation.The operation of converters containing transformers may be understood by inserting the model of Fig. 6.17(b) in place of the transformer in the converter circuit. Analysis then proceeds as described in the previous chapters, treating the magnetizing inductance as any other inductor of the converter.Practical transformers must also contain leakage inductance . A small part of the flux linking a winding may not link the other windings. In the two-winding transformer, this phenomenon may be modeled with small inductors in series with the windings. In most isolated converters, leakage inductance is a nonideality that leads to switching loss, increased peak transistor voltage, and that degrades cross-regulation, but otherwise has no influence on basic converter operation.There are several ways of incorporating transformer isolation into a dc-dc converter. The full-bridge , half-bridge , forward , and push-pull converters are commonly used isolated versions of the buck converter. Similar isolated variants of the boost converter are known. The flyback converter is an isolated version of the buck-boost converter. These isolated converters, as well as isolated versions of the SEPIC and the Cuk converter, are discussed in this section.Fig. 6.19. Full-bridge transformer-isolated buck converter: (a) schematic diagram, (b) replacement of transformer with equivalent circuit model.6.3.1 Full Bridge and Half-Bridge Isolated Buck ConvertersThe full-bridge transformer-isolated buck converter is sketched in Fig. 6.19(a). A version containing a center-tapped secondary winding is shown; this circuit is commonly used in converters producing low output voltages. The two halves of the center-tapped secondary winding may be viewed as separate windings, and hence we can treat this circuit element as a three-winding transformer having turns ratio 1:n :n . When the transformer is replaced by the equivalent circuit model of Fig. 6.17(b), the circuit of Fig. 6.19(b) is obtained. Typical waveforms are illustrated in Fig. 6.20. The output portion of the converter is similar to the nonisolated buck converter - compare the v s (t ) and i (t ) waveforms of Fig. 6.20 with Figs. 2.1(b) and 2.10.Fig. 6.20. Waveforms of the full-bridge transformer-isolated buck converter.During the first subinterval 0 < t < DT s , transistors Q 1 and Q 4 conduct, and the transformer primary voltage is v T = V g . This positive voltage causes the magnetizing current i M (t ) to increase with a slope of V g /L M . The voltage appearing across each half of the center-tapped secondary winding is nV g , with the polarity mark at positive potential. Diode D 5 is therefore forward-biased, and D 6 is reverse-biased. The voltage v s (t ) is then equal to nV g , and the output filter inductor current i (t ) flows through diode D 5.Several transistor control schemes are possible for the second subinterval DT s < t < T s . In the most common scheme, all four transistors are switched off, and hence the transformer voltage is v T = 0. Alternatively, transistors Q 2 and Q 4 could conduct, or transistors Q 1 and Q 3 could conduct. In any event, diodes D 5 and D 6 are both forward-biased during this subinterval; each diode conducts approximately one-half of the output filter inductor current.Actually, the diode currents i D 5 and i D 6 during the second subinterval are functions of both the output inductor current and the transformer magnetizing current. In the ideal case (no magnetizing current), the transformer causes i D 5(t ) and i D 6(t ) to be equal in magnitude since, if i l ’(t ) = 0, then ni D 5(t )=ni D 6(t ). But the sum of the two diode currents is equal to the output inductor current:)()()(65t i t i t i D D =+ (6.20)Therefore, it must be true that i D 5 = i D 6 = 0.5i during the second subinterval. In practice, the diode currents differ slightly from this result, because of the nonzero magnetizing current.The ideal transformer currents in Fig. 6.19(b) obey0)()()('65=+−t i t ni t i D D l (6.21)The node equation at the primary of the ideal transformer is0)(')()(=+=t i t i t i l M l (6.22)。
开关电源电路设计指引
电控设计规范NCP1001PG 开关电源电路设计指引（发布日期：2005-10-7）1范围本设计指引对美的变频空调室外机电控板应用的NCP1001PG 开关电源的基本原理，硬件电路的参数计算选择，相关技术要求和应用的有关问题进行了阐述。
2引用标准下列标准所包含的条文，通过在本标准中引用而构成为本标准的条文。
本标准出版时，所示版本均为有效。
所有标准都会被修订，使用本标准的各方应探讨使用下列标准最新版本的可能性。
GB4706.1-2005 家用和类似用途电器的安全第一部分：通用要求GB4706.32-2004 家用和类似用途电器的安全热泵、空调器和除湿机的特殊要求QJ/MK02.008-2003 空调器电子控制器3定义3.1脉冲宽度调制方式 Pulse Width Modulation脉冲宽度调制方式，简称脉宽调制（缩写为PWM）方式。
其特点是固定开关频率，通过改变脉冲宽度来调节占空比。
目前，集成开关电源大多采用PWM方式。
4安森美NCP1001PG 开关电源方案简介4.1安森美主芯片NCP1001P基本简况主芯片NCP1001P内置集成700V高耐压开关管，采用PWM控制方式，固定100KHZ开关频率,外围器件少，设计简单的单芯片开关电源方案。
整流二极管建议使用超快恢复型MUR220 (2A/200V)、MUR120(1A/200V)，可降低省耗，提高转换效率，减少噪声。
（附表为主要参数）4.2主要特征a.宽电源输入电压范围85Vac-265Vac。
b.电源转换效率高，满载时可到达75％。
c.多路电压输出，电压稳定性好。
d.有输出过功率保护max30w和短路保护自恢复功能。
e.有输出过压保护自恢复功能（如反馈回路器件失效）。
f.有内置过热迟滞（30°自恢复保护（IC结温超过140°）g.低功待机小于0·5w4.3安森美开关电源方案优、缺点优点：a、外围器件少，设计调试简单。
开关电源设计手册（看2遍就懂）.pdf
开关电源设计⼿册（看2遍就懂）.pdf 反激式开关电源变压器设计计学习培训教材反激式开关电源变压器设计(2)⼀、变压器的设计步骤和计算公式：1.1 变压器的技术要求:输⼊电压范围；输出电压和电流值；输出电压精度；效率η；磁芯型号；⼯作频率f；最⼤导通占空⽐Dmax；最⼤⼯作磁通密度Bmax;其它要求。
1.2 估算输⼊功率，输出电压，输⼊电流和峰值电流:1）估算总的输出功率：P o=V01x I01+V02x I02……2）估算输⼊功率：P in= P o/η3）计算最⼩和最⼤输⼊电流电压V in(MIN)=AC MIN x1.414(DCV)V in(MAX)=AC MAX x1.414(DCV)技术部培训教材反激式开关电源变压器设计(2)4）计算最⼩和最⼤输⼊电流电流I in(MIN)=P INx VIN (MAX)Iin(MAX)=PINx VIN (MIN)5）估算峰值电流：K POUTI PK = VIN (MIN)其中：K=1.4(Buck 、推挽和全桥电路）K=2.8(半桥和正激电路） K=5.5(Boost ，Buck- Boost 和反激电路）技术部培训教材反激式开关电源变压器设计(2)1.3 确定磁芯尺⼨确定磁芯尺⼨有两种形式，第⼀种按制造⼚提供的图图表表，，按按各各种种磁磁芯芯可传递的能量来选择磁芯，例如下表：表⼀输出功率与⼤致的磁芯尺⼨的关系输出功率/W MPP环形E-E、E E--L L等等磁磁芯芯磁芯直径/（in/mm) (每边)/()/(in/mm)in/mm)<5 0.65(16) 0.5(11)5(11)<25 0.80(20) 1.1(30)1(30)<50 1.1(30) 1.4(35)4(35)<100 1.5(38) 1.8(47)8(47)<250 2.0(51) 2.4(60)4(60)技术部培训教材反激式开关电源变压器设计(2)第⼆种是计算⽅式，⾸先假定变压器是单绕组，每增加加⼀⼀个个绕绕组组并并考考虑安规要求，就需增加绕组⾯积和磁芯尺⼨，⽤“窗⼝利⽤⽤因因数数””来来修修整整。
开关电源制作设计(电路原理图+PCB)
一、工作原理我们先熟悉一款开关电源的工作原理，该电源可输出5V电压，如图1所示。
1. 抗干扰电路在电网输入端首先设置一个NTC5D-9负温度系数热敏电阻，作用是保护后面的整流桥，刚开机时热敏电阻处于冷态，阻值比较大，可以限制输入电流，正常工作时，电阻比较小。
这样对开机时的浪涌电流起到有效的缓冲作用。
电容CY1、CY2、CY3、CY4用以滤除从工频电网上进入开关稳压电源和从开关稳压电源进入工频电网的不对称杂散信号，电容CX1、CX2用以滤除从工频电网上进入开关稳压电源和从开关稳压电源进入工频电网的对称杂散信号，用电感L1抑制从工频电网上进入开关稳压电源和从开关稳压电源进入工频电网的频率相同、相位相反的杂散干扰电流信号。
采用高频特性好的瓷片电容和铁芯电感，实现开关稳压电源电路中的高频辐射不污染工频电网和工频电网上的杂散电磁波不会窜入开关稳压电源电路中而干扰和影响其工作，对高频分量或工频的谐波分量具有急剧阻止通过功能，而对于几百赫兹以下的低频分量近似一条短路线。
图1 开关电源的工作原理图2. 整流滤波电路在电路中D1、D2、D3、D4组成全桥整流电路，把输入的交流电压进行全波整流，然后用C1进行滤波，最后变成直流输出供电电压，为后级的功率变换器供电，整流滤波后的电压约为300V。
3. UC3842供电与振荡300V的脉动直流电压，此电压经R12降压后给C4充电，供电UC3842的7脚，当C4的电压达到UC3842的启动电压门槛值时，UC3842开始工作并提供驱动脉冲，由6脚输出推动开关管工作。
一旦开关管工作，反馈绕组的能量经过D6整流，C4滤波，又供电到UC3842的7脚，这时可以不需要R12的启动了。
C9、R11接UC3842的定时端，和内部电路构成振荡电路，振荡的工作频率计算为：f=1.8/（Rt*Ct）代入数据可计算工作频率：f=68.18K4. 稳压电路该电路主要由精密稳压源T L 4 3 1 和线性光耦P C 8 1 7 组成，假设输出电压↑→经过R 1 6 、R 1 9 、R20、RES3的取样电压↑→TL431的1脚电压↑，当该脚电压大于TL431的基准电压2.5V时，TL431的2、3脚导通，→通过光电耦合到UC3842的2脚，于是UC3842的6脚驱动脉冲的占空比↓→开关变压器T1绕组上的能量↓→输出电压↓，达到稳压作用；反之，假设输出电压下降，则稳压过程与上相反。
ti 开关电源的原理和设计手册
开关电源指的是利用开关管进行开关控制的电源，相较于传统的线性电源，开关电源具有体积小、效率高、可靠性强等优点，因此得到了广泛的应用。
开关电源的原理和设计手册是开发和应用工程师们必备的基础知识，本文将围绕开关电源的原理和设计手册展开详细的介绍。
一、开关电源的工作原理1. 开关电源的基本结构开关电源一般由整流器、滤波器、开关管、变压器、控制电路、稳压电路等部分组成。
其中开关管作为关键部件，通过不断地打开和关闭来控制电压的变化，从而实现电源的输出。
2. 开关电源的工作原理开关电源的工作原理是通过开关管控制输入电压的断断续续，将高压直流电转换成低压直流电，再通过稳压电路保证输出电压的稳定性。
在开关管导通时，电压源充电，并将能量储存在电感中；在开关管关断时，电感释放能量，输出电压使负载得到供电。
二、开关电源的设计手册1. 开关电源设计的基本流程（1）确定设计需求和规格要求在设计开关电源之前，需要明确所需的电压、电流、功率等参数，以及工作环境、安全标准等规格要求。
（2）选择合适的开关元件和辅助元件根据设计需求，选择合适的开关管、变压器、电感、电容等元件，保证电源的性能和可靠性。
（3）设计控制电路和稳压电路通过合理的控制电路和稳压电路设计，实现对输入电压的精确控制和输出电压的稳定性。
（4）进行系统仿真和调试利用仿真软件对设计的开关电源进行系统仿真，验证电源的性能和稳定性，并在实际电路中进行调试和优化。
2. 开关电源的设计要点（1）电源的高效率高效率是开关电源设计的重要目标，可通过合理选择元件和优化电路结构来提高电源的效率。
（2）电源的稳定性稳定的输出电压是电源设计的关键，需要通过稳压电路和反馈控制来保证电源输出的稳定性。
（3）电源的过流、过压、过温保护为了保护电源和负载安全，需要在设计中考虑过流、过压、过温保护功能，避免出现意外故障和损坏。
（4）电源的EMI设计开关电源在工作时会产生电磁干扰，需要在设计中考虑电源的EMI设计，减小对周围电路的干扰。
开关电源设计从入门到精通
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4.地连续性-以 25A,1 分钟检查,确认安全接地的阻抗小于 0.1Ω. 1.7 电磁兼容 符合邮电部通信电源标准. 2 机械规范 尺寸：略 重量： 略 安装方向： 模块设计安装方向是面板垂直放置,使空气垂直通过模块. 通风和冷却： 模块的顶部和底部都有通风槽,使空气流通过模块,经过散热器.因此在系统中应当没有阻碍地对流冷却 模块,并应强迫冷却装置使冷却空气经过模块自由流通. 3 环境条件 环境温度： 在 0～55℃温度范围内满功率工作.在模块下 50mm 处模块的入口测量温度. 存储温度：－40～+85℃ 湿度：5％～80％,不结冰. 高度： -60m～2000m 工作;-60m～10000m 不工作. 4 可靠性 MTBF 大于 100000 小时.
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为后备能源。但是电池涉及到电化学和冶金学知识，已超出一般电气工程师的知识范畴。这里介绍一 些使用电池基础知识，使你知道设计充电电源和使用电池供电时应注意的一些问题。 利用电化学可逆原理做成的最基本的单元电池叫单体电池。典型的单体电池是由两个金属极板和 构成它们之间导电通路工作介质组成，这种通路材料可能是液体或固体，与特定化学机理有关。这种 结构关键在于是否能够更有效进行电－化学反应（可再充电，即二次电池，也称为蓄电池。不能再充 电叫一次电池）。根据不同通路材料的安排，一个金属极板为电池的阳极－正极，另一个则为阴极－ 负极。如将两个金属极板（阴极和阳极）接到电源上，电的作用改变了工作介质的化学状态，这就是 储能。如将已储能的电池极板接到负载，材料化学作用放出电荷返回到原始状态，释放出电能。 单体电池一般很低，例如铅酸蓄电池单体电池额定电压为 2V。因此较高电压的电池一般由许多单 应当注意：不要自己将电池连接成你需要的电压和容量，电池不能直接并联！你只 体电池串联组成。应 能按制造厂系列产品选择你需要的电池容量和电压。如果在每个电池端串联一个二极管就可以并联。 在电池工作范围内，电池看起来像一个理想电压源，但实际电源并非如此。首先，当充电时，端电压 会升高；放电时，端电压会降低。这就说明蓄电池存在内电阻，图 1.1 是标称电流压 12V 的 NiH 电池 的伏安特性，随着输出电流的增加，输出电压下降（类似正弦双曲线）。标称电压为 12V，电池放出 电流为负，充电电流为正。电池放出小电流时，电池端有一个类似电阻的压降，电流加倍压降也几乎 加倍；在大电流时，电压降增加减慢；在端电压下降到零以前，电流可以达到非常大的数值，但绝对 不能将电池短路，如果将 NiH 电池输出短路将引起电池爆炸！其次，电池不是与频率无关的电压源， 在充电和放电时，产生电化学作用需要一定的时间，等效为电容与内阻并联。此外，在典型开关频率 20kHz 或更高时，电池有很大内阻抗。这是因为电池端子间，内部极板间存在小电感；例如，一个 NiH（镍－氢）电池可能具有 200nH 的感抗，五个这样的电池串联（获得 6V 电压）有大约 1μH 电 感。如果开关频率为 200kHz，阻抗大约 1Ω。所以这时电池不是理想电压源，不可能吸收你的变换器 产生的开关纹波，为此，通常在电池的两端并联一个电容，减少内电感的影响。 电池输出电流和输出电压的关系还与温度以及电池剩余电荷量有关。如果放电电流太大，会损伤 电池。几乎所有电池，如果在远低于它的工作温度下放电，也会损坏电池。例如密封铅酸电池在低于 －10℃不能工作,这就是为什么在很冷的天气发动不了你的汽车。 制造厂标定电池的容量一般以电池具有的电 I(A) 荷量－安时（电流×时间＝电荷 AH）来表示。 这使得电源设计者感到为难，你不能够简单得到 2.00 电池输出参数与多大能量的关系，因为它不等于 电池容量乘以输出电压；何况输出电压又与输出 0.00 电流有关。这些参数关系由制造厂以曲线形式提 供的，而曲线似乎不能直接找到你设计需要的工 -2.00 作点，需要从这些曲线来回参照得到你需要的数 据。你自己测试电池是不切实际的，因为每个制 造厂制造的电池总有些小的差别，所以你不能假 10.0 11.0 12.0 13.0 14.0 定每个电池具有相同的化学特性和安时定额，以 电池电压(V) 及它们在同一场合具有相同的运行时间。 图 1.1 典型 12V 电池 V-I 特性 另一个现象是自放电。如果你充好电的电池 放置在那里，不接任何负载，它自己会逐渐失去存储的能量。失去能量所需要的时间与化学工作介质 有关：如 NiH 电池 24 小时；密封铅酸蓄电池在温度 25℃下约 16 月容量损失 50%，温度升高 10℃,时 间缩短一半。而某些锂电池可达几年不等。所以放置不用的铅酸电池一般每 3 个月得进行充放电维护 一次。 电池不可能无限期充放电使用，电池也有寿命。在一定时间范围内，电池经过多次充电/放电周期 以后，不再能存储额定容量，这个时间就是电池寿命的终止。它取决于电池如何工作，它经历了多少 个充电/放电周期，放电的深度如何等等。例如，铅酸密封电池放电深度 50%额定容量，充放电可达 500～600 次；放电深度 100%，寿命仅 200～300 充放电周期。即使电池用于备份，所谓浮充状态（总 是保持充满状态），在 5～10 年内也需要更换。 电池是一个不愉快的能源，它也是一个不舒服的负载。当你对电池补充充电－均衡充电时，你不 能用一个电压源对其充电，因为电池充电电流与电压成指数关系，会造成充电电流热失控，将导致电 池损坏。因此所有类型电池充电必 必须采取限流措施。如果电池充满，即达到额定电压时，应当转换到 浮充电状态，补充自放电失去的能量，以保证电池保持满容量状态。
开关电源设计手册 SMPS design
开关电源设计手册目录1 隔离式电源设计1.1 有源功率因数校正1.2 反激式电源设计1.3 正激式电源设计2 非隔离式电源设计2.1 非隔离式降压型电源设计1.1 有源功率因数校正APFC: Active Power Factor Correction一, 功率因数校正的基本原理理论上: P.F.= P/S=(REAL POWER)/(TOTAL APPARENT POWER)=Watts/V.A.=有功功率/视在功率对于输入电压和电流都是理想的正弦波的情况, 如果把输入电压和输入电流的相位差定义为φ, 那么, P.F.=P/S=Cosφ. 相应的功率相量图如下:对于非理想的正弦波, 假设输入电压为正弦波, 输入电流为周期性的非正弦波, 比如在实际的AC-DC线路中广泛应用的全波整流, 只有当输入电压大于电容的电压时, 才有市电电流给电容充电.在这种情况下, 电压有效值Vrms=Vpeak/√2周期性的非正弦波电流经过傅里叶变换为:(Io: 电流直流分量; I1RMS: 电流基波分量, 頻率与V相同; I2RMS….I nRMS: 电流谐波分量, 频率为基波的2….n 倍. )对于纯净的交流信号, Io=0; I1RMS基波分量有一个同向成份I1RMSP和一个求积成份I1RMSQ.于是电流有效值可以表达为:有功功率P=V RMS*I1RMSP=V RMS*I1RMS*Cosφ1(φ1: 输入电压和输入电流基波分量I1RMS的相位差)S=V RMS*IRMS total于使功率因数Power Factor 可以表达为:P.F.=P/S= (I1RMS/I RMS total)* Cos φ1;定义电流失真系数K= I1RMS/I RMS total = Cosθ; θ为失真角(Distortion angle); K 为与电流谐波(Harmonic) 分量有关的系数. 如果总的谐波分量为零, K 就为1.最后, 可以表达为: P.F.=Cos φ1*Cos θ ; 功率向量图如下:φ1 是电压V与电流基波I1RMS之间的相量差;θ是电流失真角;可见功率因数 (PF) 由电流失真系数 ( K ) 和基波电压、基波电流相移因数( Cos φ1) 决定。
开关电源设计很全的资料
1目录1 开关电源的特点与分类 (1)1.1 线性、开关电源的特点 (1)1.2 开关电源的电路类型 (1)1.3 开关电源的工作模式 (4)1.4 零电压开关(ZVS)和零电流开关(ZCS)方式 (5)2 开关电源的拓扑结构 (8)2.1 BUCK变换器的基本原理 (8)2.2 BOOST变换器的基本原理 (9)2.3 BUCK/BOOST变换器的基本原理 (10)2.4 反激变换器的基本原理 (12)2.5 正激变换器的基本原理 (16)2.6 推挽式变换器的基本原理 (18)2.7 电流型半桥变换器的基本原理 (20)2.7.1 基本原理 (20)2.7.2 控制要求 (23)2.8 电压式半桥式变换器 (24)2.9 全桥式变换器的基本原理 (25)2.10 半桥LLC谐振变换器的基本原理 (27)3 变压器的设计 (33)3.1 变压器的工作原理 (33)3.2 变压器的模型 (35)3.3 高频变压器对磁芯材料的要求 (37)3.4 高频变压器设计考虑的几个问题 (38)3.5 寄生参数和影响 (39)3.6 高频变压器设计步骤 (41)4 高频变压器的绕组 (48)4.1 Ansys 有限元分析软件 (48)4.2 通电导线的集肤和邻近效应 (53)4.3 不同绕组结构对高频变压器电磁参数的影响 (54)4.4 不同绕组结构高频变压器的设计示例 (58)5 AC/DC开关电源实例 (63)5.1 65W 反激开关开关电源 (63)5.1.1 产品特色 (64)5.1.2 典型应用及引脚功能描述 (65)5.1.3 TOP264-271 功能描述 (66)5.1.4 65 W通用输入适配器电源 (70)5.2 24W 反激开关电源的设计 (74)5.2.1 电路原理图 (74)5.2.2 电路描述 (76)5.2.3 变压器规格 (78)5.3 带PFC的半桥谐振LLC开关电源 (80)5.3.1 PFC电路 (80)5.3.2 LLC部分 (86)5.4 120W/19V双开关反激式开关电源 (116)5.4.1 FAN6920介绍 (116)5.4.2 功能说明 (119)5.4.3 电路图 (140)6 DC/DC开关电源实例 (141)6.1 隔离式正激DC-DC变换器 (141)6.1.1 基本性能和典型应用 (142)6.1.2 应用信息 (144)6.1.3 控制信息 (160)6.2 30W正激DC/DC开关电源 (169)6.2.1 产品特色 (170)6.2.2 功能描述 (171)6.2.3 引脚功能描述 (172)6.2.4 DPA-Switch产品系列功能描述 (173)6.2.5 正激30 W开关电源 (176)6.3 6-42V输入、5V输出的DC/DC变换器 (178)6.3.1 LM25574芯片介绍 (178)6.3.2 工作描述 (182)6.3.3 应用信息 (191)6.4 500W DC/DC变换器 (203)6.4.1 L6599简介 (203)6.4.2 全桥LLC 变换器的工作原理分析 (204)6.4.3 LLC 全桥谐振变换器主电路参数设计 (208)6.4.4 LLC 全桥谐振变换器控制电路参数设计 (209)6.5 120W/24V LLC谐振变换器 (211)6.5.1 引言 (211)6.5.2 工作原理和基波近似 (213)6.5.3 设计流程 (224)7 LED电源实例 (233)7.1 50W 直流小功率恒流源 (233)7.1.1 功能介绍 (233)7.1.2 典型电路和实际电路 (235)7.2 交流大功率恒流源 (245)7.2.1 芯片特性和引脚功能 (245)7.2.2 充电电流控制的工作原理 (247)7.2.3 混合控制(PWM+PFM) (252)7.2.4 电流检测 (253)7.2.5 软启动和输出电压调节 (258)7.2.6 功能设置 (260)7.3 交流18W LED恒流源驱动 (276)7.3.1 电源管理芯片DU8633 (278)7.3.2 电路参数设计 (279)7.4 70W LED 照明灯电源 (285)7.4.1 BCM 升压PFC 转换器的基本工作原理 (288)7.4.2 准谐振反激式转换器的工作原理 (290)7.4.3 设计思路 (292)7.4.4 直流-直流部分 (298)7.5 16.8 W/24V LED反激式驱动电源 (310)7.5.1 芯片描述 (310)7.5.2 电源设计 (324)8 数字电源实例 (327)8.1 UCD3138的数字电源 (327)8.1.1 器件概述 (327)8.1.2 描述 (336)8.1.3 总体概览、系统模块与IDE计算 (348)8.1.4 DPWM工作模式 (352)8.1.5 自动模式开关 (359)8.1.6 滤波器 (365)8.1.7 典型应用 (368)8.2 小功率数字充电电源 (379)8.2.1 国内外数字电源发展现状 (380)8.2.2 设计指标 (382)8.2.3 系统总体设计 (383)8.2.4 基于L6562的PFC电路设计 (385)8.2.5 控制软件 (391)9 参考文献 (397)1开关电源的特点与分类1.1线性、开关电源的特点线性电源（SwichingMode Power Supply）首先通过工频变压器降压，再用整流桥整流，之后利用功率半导体器件工作在线性放大状态，通过调节调整管的线性阻抗来达到调节输出电压的目的。
llc开关电源设计书籍
llc开关电源设计书籍LLC开关电源设计是电力电子领域的一个重要课题，本文将介绍一些相关的书籍，帮助读者更好地了解LLC开关电源的设计原理和方法。
1.《LLC Resonant Converter: Analysis, Control, and Design》这本书是由Ming Xu和Gerhard W. Semmelhack合著的，是关于LLC谐振变换器分析、控制和设计的权威指南。
书中详细介绍了LLC拓扑的工作原理、分析方法和设计步骤，并提供了实际的设计示例和实验结果。
此书适合电力电子工程师和研究人员阅读，对于深入理解LLC开关电源的原理和设计方法非常有帮助。
2.《Design of LLC Resonant Converter with Adaptive Control》这本书是由Li Yang编写的，主要介绍了带自适应控制的LLC谐振变换器的设计方法。
书中首先介绍了LLC拓扑的基本原理和特点，然后详细讲解了自适应控制的原理和设计步骤。
此书还包含了大量的仿真结果和实验验证，可以帮助读者更好地理解和应用自适应控制技术。
3.《Power Electronics: Converters, Applications, and Design》这本书是由Ned Mohan、Tore M. Undeland和William P. Robbins合著的，是电力电子领域的经典教材之一。
书中系统地介绍了各种电力电子变换器的原理、应用和设计方法，包括LLC谐振变换器。
此书内容丰富，结构清晰，适合作为电力电子专业的教材或参考书使用。
4.《Switching Power Supply Design》这本书是由Abraham I. Pressman、Keith Billings和Taylor Morey合著的，是关于开关电源设计的经典教材之一。
书中系统地介绍了开关电源的设计原理、电路拓扑和控制方法，包括LLC谐振变换器的设计。
高功率HE系列电子开关白皮书说明书
HIGH POWER RELAYSHE SERIESSwitching high loads on the PCB made easyWHITEPAPERCONTENTSIntroductionSwitching High Loads on the Circuit Board Overview of the HE series Application ScenariosInvertersCharging stationBattery storageConclusion 3 4 5 5 6 7 8 9INTRODUCTIONAs global awareness about ecological and energy concerns broadens, newly developed relays that meet the changing needs of the energy and automotive sectors have been in-creasingly in demand. Energy generation from wind and es-pecially solar panels is becoming more widespread than ever, as private households install their own photovoltaic systems and connect them with battery storage systems. At the same time, electromobility is promising to usher in a new age of transportation.”Higher switching capacities and smaller dimensions – these are the demands relays are facing today.“These are exciting new markets for relay technologies, but they come with a need for new switching solutions. Higher switching capacities and smaller dimensions – these are the demands relays are facing today. Switch solutions that can handle the high currents involved in energy management sys-tems are becoming paramount. At the same time, components such as Printed Circuit Boards (PCBs) have been getting more intricate, both in size and complexity. New multilayer circuit boards, high power connectors and advanced materials are just some of the innovations that have been introduced in the last few years.As a result, downsizing and energy saving have become more and more pertinent for electromechanical relays as well. As a leading force in switching technology, Panasonic Industry has made massive strides in this direction. The overall volume in cm³ has more than halved in the last twenty years, whereas output density rose from around 0.8 to 1.1 KVA/cm³ since 2010. Today more than ever, it is easier to control high loads with increasingly smaller relays.Combining the new sophisticated PCBs with a cutting-edge high-power relay portfolio, Panasonic Industry offers a solu-tion that switches high loads directly on the PCB: the HE series for high power applications.SWITCHING HIGH LOADS ON THE CIRCUIT BOARDThe HE series can withstand loads up to 120A for the HE-N and up to 1000VDC for the HE-V in ambient temperatures of up to 85°C. This makes the PCB relays suited for high power appli-cations, such as solar inverters, automotive charging stations, or battery storage systems. The relay is no longer responsible for controlling a secondary contactor, switching about 500mA up to a few amperes – it becomes the central load-carrying el-ement, making other contactors unnecessary in a lot of cases. Switching high currents on the PCB becomes easier than ever. This comes with a number of advantages: compared to contac-tors, the high-power PCB relays are much more compact. The HE-S, for example, manages to integrate two NO contacts into extremely small dimensions of 30x36x40mm. Furthermore, as the relays are mounted directly on the PCB, they require no control cabinet. A developer can thus switch from a screwed or wired solution to a switching element that is directly con-nected to the PCB. This not only saves space, but also makes manual assembly unnecessary and installation a lot easier – two factors that contribute to cost savings.…Extremely low power dissipation atthe contacts is achieved by reducing the contact resistance to between 1 and 3mΩ.“Apart from component size, minimal energy consumption is a central feature of the HE series relays. Extremely low power dissipation at the contacts is achieved by reducing the contact resistance to between 1 and 3mΩ. Thus, no significant heating occurs at the contact and thermal losses are reduced – with a current of 35A and a contact resistance of 2mΩ, power dis-sipation is only 2.54W. Combined with the use of Pulse Width Modulation technology (PWM), which brings down the relay’s operating power (170mW for the HE-S), energy consumption is decreased significantly. This is further helped by the fact that, due to reduced waste heat, ventilation is no longer necessary in most cases. All relays in the series offer a contact distance of at least 2.5mm up to 3.8mm and high creepage clearance. This makes them well-insulated and guarantees high dielectric strength and protection against surge voltages. The unique relay struc-ture of the HE-S allows a normally closed (1FormB) moni-toring contact to be implemented, which recognizes when welding of the main contacts occurs. This feedback contact is compliant with EN60947-4-1 for safety circuits and con-forms to EN61851-1, which makes it suitable for automotive charging solutions.With a mechanical life of at least one million operations, the HE series guarantees a problem-free and long service life. Overall, the PCB relays help to cut back on space, energy con-sumption, and costs.The HE power relayscan switch high loads.APPLICATION SCENARIOS”Specifically, the use of HE relays in (solar) inverters, automotive charging solutions and battery storage systems has proven beneficial.“The PCB high power relays in the HE series make them perfect for the field of energy manage-ment, where high currents need to be switched safely, reliably, and cost-efficiently. Specifically, the use of HE relays in (solar) inverters, automotive charging solutions and battery storagesystems has proven beneficial.INVERTERSIn the three years between 2012 and 2015, global sales of solar inverters have risen from approx. 2,800,000 to 4,000,000 units. Since then, this trend has only accelerated. Private house-holds are routinely installing photovoltaic systems in their homes, connecting them to storage batteries and the power grid. These wall-mounted inverters are becoming smaller and lighter, while at the same time, higher performance classes between 60kW and 100kW are becoming more common. This also has an effect on the relays used, which are increasingly required to deliver higher switching capacities.”While the inverter usually switches currents by a semiconductor, a relay is used to bypass it during the pre-charging process.“Inside the converter, very high currents occur during the charging of the capacitor. This component is used to prevent voltage fluctuations on the input side when a large load is con-nected to the output side. To limit the inrush current, a pre-re-sistor is used to pre-charge the capacitor with approximately 10A for 0.5 seconds. While the inverter usually switches cur-rents by a semiconductor, a relay is used to bypass it duringthe pre-charging process. As soon as the current stabilizes, the relay closes and the current flows through the closed con-tacts.The reasons for using a relay in this instance are simple: elec-tromechanical relays like HE-Y5 have a much lower and more stable contact resistance than high power semiconductors.Thus, unwanted consequences like power dissipation and heat generation remain minimal.”Compared to the HE-Y5 relay, industrial contactors take up a lot of space, have a high power dissipation and are more expensive.“For switching currents above 60 A, industrial contactors have been mainly used so far. As the IEC62109-1 solar industry standard requires single fault protection as soon as inverters are connected to the public grid, two separately controlled contacts per phase must therefore be connected in series. This ensures that, if one contact is welded, the second contact can still open safely. With industrial contactors, two three-pole contactors are therefore required for three phases. Even though six relays are necessary for the same purpose, the to-tal balance in terms of energy and cost efficiency is still bet-ter: compared to the HE-Y5 relay, industrial contactors take up a lot of space, have a high power dissipation and are more expensive. While energy savings for each individual relay are rather small, the surplus adds up when considering the sheer number of inverters currently on the market. Therefore, se-lecting a higher quality relay instead of a secondary contactor pays off.With its low contact resistance, the HE-Y5 is perfectly suited for the use in solar inverters.CHARGING STATIONSWith national and global commitments to reduce CO2 emis-sions, e-mobility concepts have experienced a veritable boost in the span of just a few years. Consequently, the necessary charging infrastructure, rather neglected until recently, has also returned to the attention of politics and engineers. In ad-dition to a public charging infrastructure, household charging systems are increasingly emerging as a viable option. Current-ly, drivers of e-cars can charge their vehicles either at a AC/ DC charging station, a AC/DC wallbox installed in their garage, or a special charging cable that gets connected to a regular household outlet or CEE-socket.A whole arsenal of mains isolating relays is currently required for a safe recharging, depending on the charging system and national standards. Thus, standard IEC 61851-1 distinguishes between charging modes on the basis of their installation and system components. However, all require a switching device in the power range from 16A/250VAC to 63 A/380VAC. This is used to connect and disconnect the electricity supply system to and from the vehicle. The latter is usually the case when a fault mode occurs, e.g. when a creepage current is detected.The HE-S power relay has already been approved for many charging stations in the market and is especially suited for wallbox systems with charging capacities of up to 22kW and 43kW. It is able to carry high charging currents at ambient temperatures of 85°C for several hours, even under direct sunlight. Furthermore, the relay’s lowest holding power is 170mW, and features a clearance and creepage distance of more than 8mm between the coil and the contacts.”The HE-S relay accommodates two NO bridge contacts with a contact gap of 3.2mm and is also available with an auxil-iary contact. With welded main contacts, this contact retains an opening of at least 0.5mm and can switch 1A at 230VAC.“Similar to the solar industry, automotive charging systems require a contact distance of at least 1.8mm; for many applica-tions, an electrically protective separation of 3mm is mandato-ry. The HE-S relay accommodates two NO bridge contacts with a contact gap of 3.2mm and is also available with an auxiliary contact. With welded main contacts, this contact retains an opening of at least 0.5mm and can switch 1A at 230VAC. This configuration is certified for mirror contacts by the VDE in ac-cordance with IEC 60947-4-1. As the auxiliary contact is switched via a separate actuator, it is electrically isolated from the con-tact set – its connections are routed outwards parallel to the coil terminals. This, combined with a large contact gap, makesthe HE-S perfectly suited for automotive charging scenarios. mandatory regulations.BATTERY STORAGE”In addition to a public charging infra-structure, household charging systems are increasingly emerging as a viable option.“Connected to the application scenarios mentioned above is a growing need for safe and effective battery storage solutions. It is therefore hardly surprising that fixed storage battery sales are growing – in J apan alone, the sales have almost tripled in the years from 2012 to 2015, especially in the infrastruc-ture and household sectors (approx. 10.000 to approx. 30.000 units).Relays that are used for charging and discharging batteries must naturally be able to carry high currents over a longer pe-riod of time. Even more important, however, is that the relay is able to cut off even a high current safely and reliably. An emer-gence cut-off occurs when equipment fails or a malfunction occurs. Thus, when lightning strikes a solar panel that is con-nected to a battery storage system, the resulting overcharge can cause damage and poses a safety hazard. DC Cut-off is therefore required for all major solar and battery storage sys-tems on the market.Power relays like the HE-V can be used for cutting off both positve and negative lines on the DC side, with a maximum of 1,000VDC. It uses a blow-out magnet mechanism and serial contact connection to maintain the required arc and gap length for high DC voltage cut-off. If a problem occurs in a photovol-taic system, the relay can short the connection between the panel and junction box. The same principle can also be applied to increase energy efficiency when a panel is in the shade. Fur-thermore, the HE-V power relay prevents inrush currents to the storage battery and is suitable for a rapid shutdown sys-tem, where the cut-off is effected by the push of an emergency button. The HE-S relay, on the other hand, is designed for AC safety cut-offs. Thus, the power relays in the HE series im-prove safety both on the input and the output side of energy storage solutions.Solar Panels Solar InverterBattery Inverter/Controller Batteries Smart MeterOff-Grid Cut Off Switch National GridCONCLUSION”It seems highly unlikely that the trends towards renewable energy and alternative driving dynamics should grind to a halt anytime soon.“Energy management, be it in electric vehicles, solar or wind power stations or photovoltaic systems in private households, is a market that is still far from reaching its peak potential. It seems highly unlikely that the trends towards renewable en-ergy and alternative driving dynamics should grind to a halt anytime soon. For electrical companies, this is a great oppor-tunity, but it also calls for new solutions that can handle high loads in a way that is both efficient and safe.The power relays of the HE series have the potential to rev-olutionize the role relays play in this field. With an improved internal architecture and compact design, they are able to control high currents directly in the circuit board. For man-ufacturers, this opens up new possibilities in terms of design and efficiency. The potential benefits are wide-reaching, as the application scenarios in this whitepaper show, ranging from significant improvements in energy efficiency to safety con-siderations. In a nutshell, trading in secondary connectors for power relays mounted on the PCB means saving space, ener-gy, and ultimately costs – without compromising on quality or performance.PANASONIC INDUSTRYEUROPE GMBHRobert-Koch-Straße 10085521 OttobrunnTelephone:+49 (0) 89 45 354-1000**********************.com。
开关电源计划书
开关电源计划书引言开关电源是一种将电源输入 voltage 持续转换为所需输出 voltage 的电子设备。
在电子设备中，开关电源广泛应用于各种领域，如计算机、通信设备、消费电子产品等。
本文档旨在介绍开关电源的基本原理和设计计划，并提供了一份详细的指南，以帮助开发人员正确设计和实施开关电源。
1. 开关电源原理开关电源是基于电力开关器件的电源系统，通过对电力开关器件的开关控制，将输入电源的电压进行变换、调整、稳定，并输出给电子设备使用。
常见的开关电源工作原理包括以下几个方面：1.1 变压器开关电源中的变压器被用于将输入的交流电转换为需要的直流电压。
变压器的构造包括一个主线圈和一个副线圈，通过施加不同的电压比例，实现输入电源电压的变换。
1.2 整流器整流器用于将交流电转换为直流电。
整流器一般包括二极管桥等元器件，将变压器输出的交流电转换为这直流电。
1.3 滤波器滤波器用于平滑变压器和整流器输出的电压波动，使输出电压更加稳定。
常见的滤波器包括电容滤波器和电感滤波器。
1.4 控制电路控制电路是开关电源的核心部分，它通过对电力开关器件的控制，确保输出电压的稳定性。
控制电路中通常包括开关电源的控制芯片、反馈电路和保护电路等。
2. 开关电源设计计划在设计开关电源时，需要考虑以下几个方面：2.1 输出功率需求根据电子设备的功率需求，确定需要设计的开关电源的输出功率。
输出功率需求将直接影响到开关电源系统的设计规格，包括变压器和电容器等元器件的选择。
2.2 输入电压范围确定开关电源能够接受的输入电压范围。
根据输入电压范围的不同，可以选择不同的变压器和整流器等元器件。
2.3 输出电压稳定性确定开关电源输出电压的稳定性要求。
输出电压稳定性是衡量开关电源质量的一个重要指标，对于不同的应用场景可能有不同的要求。
2.4 效率要求确定开关电源的转换效率要求。
高效率的开关电源可以减少能量损耗，提高整个系统的效能。
2.5 控制电路的选择根据实际需求，选择合适的开关电源控制芯片和保护电路。
ACDC开关电源的方案设计书
作者：Pan Hon glia ng仅供个人学习大连理工大学网络教育学院《电源技术》课程设计题目：PWM控制芯片SG3524的特殊应用研究学习中心：扬州奥鹏远程教育中心层次： ____________ 高起专________专业：电气自动化及其运用年级：2011年秋季学号： __________________________学生： ___________ 韩盛华________辅导教师： ________ 张春卫________完成日期：2011 年08月12 日1 绪论开关电源（Switching Mode Power Supply，英文缩写为SMPS）又称为开关稳压电源，问世后在很多领域逐步取代了线性稳压电源和晶闸管相控电源。
随着全球对能源问题的越来越重视，电子产品的耗能问题将愈来愈突出，如何降低其待机功耗，提高供电效率成为一个急待解决的问题。
传统的线性稳压电源虽然电力结构简单、工作可靠，但它存在着效率低（只有40%〜50%）、体积大、铜铁消耗量大，工作温度高及调整范围小等缺点。
为了提高效率，人们研究出了开关式稳压电源，它的效率可达85%以上，稳压范围宽；除此之外，还具有稳压精度高的特点，是一种较理想的稳压电源。
开关电源具有效率高、体积小、重量轻、应用广泛等优点，现已成为稳压电源的主流产品。
正因为如此，开关电源被誉为高效、节能型电源，代表着稳压电源的发展方向，并已广泛应用于各种电子设备中⑴o 1.1 开关电源的特点1.1.1 开关电源的优点（1）功耗小，效率高。
晶体管V在激励信号的激励下，它交替地工作在导通一截止和截止一导通的开关状态，转换速度很快，频率一般为50kHz左右，在一些技术先进的国家，可以做到几百或者近1000kHz。
这使得开关晶体管V的功耗很小，电源的效率可以大幅度地提高，其效率可达到80%o(2) 体积小，重量轻。
采用高频技术，省掉了体积笨重的工频变压器。
由于调整管V上的耗散功率大幅度降低后，又省去了较大的散热片。
开关电源培训资料共46页文档48页PPT
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29、勇猛、大胆和坚定的决心能够抵得上武器的精良。——达·芬奇
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30、意志是一个强壮的盲人，倚靠在明眼的跛子肩上。——叔本华
谢谢！
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Hale Waihona Puke 开关电源培训资料共46页文档
1、合法而稳定的权力在使用得当时很 少遇到 抵抗。 ——塞 ·约翰 逊 2、权力会使人渐渐失去温厚善良的美 德。— —伯克
3、最大限度地行使权力总是令人反感 ；权力 不易确 定之处 始终存 在着危 险。— —塞·约翰逊 4、权力会奴化一切。——塔西佗
5、虽然权力是一头固执的熊，可是金 子可以 拉着它 的鼻子 走。— —莎士 比
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26、要使整个人生都过得舒适、愉快，这是不可能的，因为人类必须具备一种能应付逆境的态度。——卢梭
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27、只有把抱怨环境的心情，化为上进的力量，才是成功的保证。——罗曼·罗兰
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28、知之者不如好之者，好之者不如乐之者。——孔子
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分立电源设计中，正确选择元件非常重要，但将其正确 布置在IC附近同样重要，这就要求高水平的技巧和经验。 设计者需要时刻注意大电流通路的长度和尺寸，关注高 频节点，谨慎提防IC及输入电源的地回路。如果电感和 电容离IC太远，会增大电流环路的寄生电容和电感，从 而引发问题。(市场上的大多数模块采用屏蔽电感，这 有助于减小与开关稳压器相关的EMI。)如果设计不正 确，补偿和反馈电路也受地噪声的影响。将模块密封在 密闭封装内有助于保护IC不受PCB布局的影响，而这类 问题在分立电源设计中普遍存在。由于模块的焊接与标 准IC类似，并且补偿电路、FET及电感全部位于内部， 接地设计也有利于控制敏感器件附近的地电流。有利于 保护电源电路不受接地反弹和其它系统噪声的影响(系 统级噪声会注入到补偿电路)，最终获得效率和可靠性 更高的电源。
由此形成可靠的新一代系统级封装(SiP)电源模块，避免 分立设计问题，同时也解决了上述问题，允许工程师将 时间投入到其它关键领域(图1)。
经过验证的同步稳压器是设计保障
IC工艺和设计的改进推动了开关电源中MOSFET晶体管 的集成，这种集成又进而推动了同步整流电源的开发， 彻底改变了DC-DC电源市场，尤其是高压应用领域。最 新的同步降压转换器具有出色的高效率、低温工作以及 较小尺寸。
电源模块如何简化设计过程
即使采用这些先进的同步降压IC，可靠的电源设计仍然 面临诸多要求，需要克服许多困难。设计者必须评估输 入电压、输出电压、负载电流、温度、抗噪性和/或辐射 等。与开关电源设计相关的难题是外部元件选择、元件 布局、PCB布局，以及控制问题，如电磁干扰(EMI)、射 频干扰(RFI)和射频抗扰性(RFS)。如果有其中任何问题 未解决，就可能引入噪声，进而耦合到供电电路或向外 耦合。
简化设计
IC工艺和封装技术领域的创新推动着集成电源模块的发展，无论是IC级还是封装级。这些新一代电源模块已经避 免了与分立元件相关的复杂问题，同时提供完备、可靠的电源方案。新一代电源模块具有高效率，几乎能够克服 大多数PCB噪声；与传统的非同步设计相比，更容易控制EMI；工作温度也低许多。此类电源模块允许系统设计 人员利用有限的时间和资源快速获取所需要的电源，将更多时间投入到其它重要领域。 Maxim Integrated将电源模块推动到了一个新高度，其模块采用经过客户验证的喜马拉雅同步降压稳压器IC，具 有工作温度低、尺寸小等优势。电源模块的设计以工程化为核心，包括需要最少的外部BOM元件、小封装、方便 的QFN类引脚排列、引脚兼容的不同电压/电流版本，并允许在大批量投产时很容易地从模块移植至IC。 电源模块的发展趋势是更高集成度、工作温度更低、尺寸更小，并重视成本削减。有了这些优势，电源设计就会 前所未有地简单、轻松。
开关电源控制器
MOSFET功率开关
电感
电源模块
补偿器件
其它无源器件
图1. 电源模块整合构建完备电源所需的全部主要器件
3
特色技术 同步电源IC相对于非同步电源IC的优势
图2所示为同步与非同步电源设计之间的差异。传统 的非同步转换器使用外部肖特基二极管进行整流，并 在高边晶体管关断期间续流。理论上，该技术比较简 单。不幸的是，实际应用中难以设计——控制更加困 难，即使该方法已经普遍采用了数十年。其最大的缺 点是二极管由于正向偏压的原因发热量巨大，所以造 成系统效率极低。 同步转换器集成了低边功率MOSFET，代替外部整流 二极管。与非同步转换器的二极管相比，MOSFET的 低电阻压降小很多；MOSFET也可在不需要时关断。 所以，大幅减小转换期间的功率损耗。这意味着电路 发热更低——效率更高。低边整流MOSFET和传统的 外部元件成为IC本身的一部分。 为了更好地理解该技术的益处，我们简单计算一下功 率损耗，将同步与非同步方案进行比较。 根据计算结果可知，同步整流方案将整流二极管的功 耗降低了60%！很伟大——毫不夸张！ 对应的热图像清晰表明，与非同步方案相比，同步 DC-DC转换器工作时的发热更少。由于温度会缩短电 子元件的使用寿命，这一点非常重要。引用Svante Arrhenius的一句话： “温度每降低10度，电路寿命将 延长一倍。 ”假设温差相差30°C，那么同步方案的寿 命将是非同步方案的8倍。
相隔离，提供附加保护。低至2.8mm的封装高度是新一 代电源模块的另一关键特征，支持其卡类应用(高度非常 重要)，也使其更容易集成散热器——对于需要耗散更多 热量的大功率应用尤其重要(图5)。
MAXM17504 5V至60V
9mm
MAXM17516 2.5V至5V
6.5mm
15mm 2.8mm 2.8mm
此外，分立电源设计要求许多外部元件，需要花费时间 和精力采购、管理库存以及安装，很难保证整体可靠性。 分立电源设计也往往意味着PC板布局面积较大，占用宝 贵的基板面积，而空间在任何时候都非常珍贵。
电源模块是解决途径
更小尺寸的工艺、IC设计以及封装优势允许模块制造商 将电源所需的无源元件及基础功能IC集成到单一芯片， 构成小尺寸电源。同步开关稳压器内置FET，比老式开 关电源尺寸更小、效率更高、准确度更高。最新的电源 模块将新型同步开关与电阻、电容、MOSFET、电感等 元件整合在一起，组成简单易用的电源模块，减小尺寸、 降低成本和布局复杂度。
MAXM17504 9mm x 15mm x 2.8mm
EP2 LX
MODE V SGND CC 大面积裸焊盘
OUT OUT OUT OUT
PCB布线难度大
图4. Maxim的QFN封装与其他球栅阵列布局的引脚排列比较
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轻松移植，满足不同的电压和电流需求
在项目设计的不同阶段，电源要求可能频繁变化。那么，如果电压或电流要求变化时，客户为什么要被迫重新设 计和重新修改电路板呢？不仅成本高，而且非常耗时！ 使用引脚兼容、具有不同电流范围和电压范围的电源模块系列产品，允许相同的布局支持不同的模块需求，不影 响PCB，进而加快上市时间。
电源设计：并非易事
从零开始设计一款可靠的电源并非易事，尤其是涉及到 开关稳压集成电路(IC)时。典型 设计是分立元件的复杂组合，要 求具备较高的专业知识和经验， 以保证电路无故障供电。电源 在系统中举足轻重，可能会延 长产品上市时间，如果处理不 当，甚至会造成系统现场失效。
随着技术的进步，模块制造 商能够在单一芯片上将无源 元件和基础功能IC集成在一 起，实现更小的电源设计。
新一代电源模块 有效简化电源设计
作者：Robert Nicoletti 战略应用工程师经理 Maxim Integrated
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摘要
分立电源设计通常要求较高的专业水准，并耗费很长时间，而电源模块提供了一条非 常简捷的途径，采用完整设计、即插即用的可靠方案，可快速实现原型或最终产品的 开发。 目前市场上的新型电源模块具有较高集成度，可实现较小尺寸系统级封装。新工艺和 新封装技术使得这些器件结构更紧凑、工作效率更高，同时避免了设计隐患，使得电 源任务更加轻松，系统设计人员能够将更多的时间用于核心电路设计，保证最快的产 品上市时间。
削减成本的途径：移植到分立IC
部分设计者对采用电源模块犹豫不决，因为电源模块不像分立式电源方案那样可随意定制，并且一般价格较高。 缺少的环节是可移植能力，不过现在已经实现。现在的设计者可从模块开始，实现快速开发；然后可以选择无缝 移植到相同IC的方案，采用分立方案。这种灵活性优化了大批量生产的性能和成本，对于追求完美的设计者非常 宝贵。
限流 VIN 控制器
频率设置 VIN 转换器 VIN VOUT VOUT 补偿 升压电容 模块
VOUT
偏置
PGOOD
升压电容
偏置
PGOOD
3A. MAX15046同步降压控制器
3B. MAX17503同步降压转换器
3C. MAXM17503模块
图3. 电源方案的演变过程
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便利性和灵活性是关键
正像您看到的，最新电源模块的外形尺寸明显减小。但 这仅仅是采用模块的优势之一。另一项优势是简单。 新形布局配置通过QFN引脚输出将引脚布置在封装周边， 使设计者更容易进行PCB布局，费用更低。将关键信号 引脚布置在封装的周边，不再需要多层电路板通过过孔 连接到模块内部的中心引脚，而采用球栅阵列封装模块 时就是遇到这种情况(图4)。周边引脚位置也提高了模块 底部可用于裸焊盘的空间，有助于模块散热，实现低温 工作。多个独立的裸焊盘将敏感的模块区域与其它区域
方案整体功耗 = 2.3W
温度低30°C
图2. 同步与非同步整流功耗比较
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通过集成补偿电路，同步整流提高了反馈调节精度。更 重要的是，整个输出电压范围的内部补偿省去了外部元 件，显著减少元件数量，缩小外形尺寸。附加利益是高 精度内部电压基准，实现更高精度的稳压——在扩展工 作温度范围内接近±1%。 使用这些带同步整流的新型集成FET开关稳压器作为电 源模块的基础，电源能够提供高效、低温升、小尺寸等 优势，并具有更高的稳压精度。例如，Maxim将喜马拉 雅IC与其它元件集成在一起，构建喜马拉雅家族电源 模块。
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电源模块也有差别
现在市场上的许多电源模块仅仅是比IC更容易使用，但 并未完全解决所有难题。理想的模块可加速产品上市时 间，并兼具低成本等关键优势，例如： • 高效率与低功耗，基于经过客户验证的可靠IC • 小尺寸，集成更多元件 • 容易使用，引脚兼容方案支持不同的电压、电流要求， 提高设计灵活性 • 灵活性，可选择低成本移植，从模块至IC，实现批量 生产
10mm
图5. Maxim的喜马拉雅模块的尺寸和高度非常小
竞争器件 11.25mm x 15mm x 4.32mm
8 7 6 5 4 3 2 1 A B BANK 1 VOUT BANK 2 GND GND SYNC ADJ PGOOD SHARE RT RUN/ SS BANK 3 FIN N.C. SYNC SS BIAS AUX BANK 4 V IN CF FB RT N.C. C D E F G H J K L 1 2 3 4 5 6 7 8 9 10 11 PGND 12 EP1 SGND PGND EP3 OUT 13 14 15 18 17 16 OUT OUT OUT RESET EN 29 28 IN 27 PGND 26 BST LX 25 24 LX 23 LX 22 LX 21 20 19 LX LX LX