A New Sensorless Vector Control Method of PMSM

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基于高频注入法的pmsm无位置传感器控制

摘要永磁同步电机(PMSM)因其体积小、效率高、能量密度高等特点，已经在工业生产、日常生活、新能源汽车等领域中得到了广泛的应用。

常用的永磁同步电机控制策略都需要实时获知转子的位置，目前一般是通过角度传感器来获得转子位置，但与此同时，带有角度传感器的控制系统往往需要控制系统提供额外的接口电路，而且需要考虑传感器的稳定性和成本等问题，一些工作情况比较恶劣的情况下甚至不允许系统加装传感器。

鉴于这些原因，无位置传感器的PMSM控制成为当前需要解决的一个问题。

本文针对这一问题，研究了基于高频信号注入法的PMSM无位置传感器的控制策略。

本文首先分析了PMSM的基本结构以及数学模型，然后介绍了空间矢量脉冲宽度调制(SVPWM)的理论。

在SVPWM的基础上，介绍了PMSM的矢量控制，即通过坐标变换解耦，把控制系统的励磁分量和转矩分量单独控制。

在矢量控制系统的大框架下，介绍了高频信号注入法的基本工作原理，即在电机的基波电压中注入幅值远低于直流总线电压、频率远高于转子电角度频率的正弦信号，然后对高频信号激励下的定子电流进行采样，通过滤波器获得含有转子位置的高频信号，再通过一系列数学运算解算出转子位置。

在这些理论基础上，建立了旋转高频注入法和脉振高频注入法的MATLAB/Simulink模型，仿真结果表明两种高频注入法都能较好的跟踪转子位置。

设计了以MKV46F256VLH16为核心的PMSM无位置传感器控制系统，并在图形化上位机FreeMASTER平台运行了基于脉振高频注入法的实验，得到了详细的实验波形和数据。

论文最后通过仿真和实验结果，得出结论。

关键词：永磁同步电机无位置传感器矢量控制高频注入法AbstractPermanent Magnet Synchronous Motor(PMSM) has been widely used in the field of industrial production, daily life, new energy vehicles and so on due to its small volume, high efficiency, high energy density, etc. In general, common control strategy for PMSM needs real-time rotor position, which is usually obtained by rotor position sensor. Meanwhile, control system with position sensor should offer additional interface electric circuit, and the stability and cost of position sensor should be taken into consideration. In addition, position sensor could not be installed in harsh situation. In consideration of these reasons, sensorless control system for PMSM need to be proposed. This paper aims at this issue and studies strategy of sensorless control on PMSM based on high frequency signal injection.This paper analyzes the basic structure and mathematic model of PMSM, and introduces the theory of Space Vector Pulse Width Modulation(SVPWM). B ased on SVPWM, vector control system of PMSM is introduced, which decouples excitation and torque variable using coordinates transform, so two variables could be controlled alone. Basic principle of high frequency signal injection is introduced based on the frame of vector control. Sinusoidal signal is injected into motor basic voltage, whose amplitude is far below dc bus voltage and frequency is far higher than rotor electrical frequency. After sampling stator current which is generated by high frequency injection, high frequency signal with rotor position information could be obtained by filter. Rotor position could be solved with mathematic operation by high frequency signal. Based on these theoretical analysis, MATLAB/Simulink model of rotating high frequency signal injection and fluctuating high signal frequency injection are built, which have superior performance on rotor position trace. At last, a sensorless PMSM control system experiment platform is designed, which uses the MKV46F256VLH16 chip as the core component, and experiment of high frequency signal injection is operated on graphic upper-computer FreeMASTER, and detailed experimental waveforms and data are obtained.Finally, this paper draw a conclusion based on simulation and experiment.Keywords：PMSM; Sensorless; Vector Control; High Frequency Signal Injection目录摘要 (I)Abstract ................................................................................................................................................... I I 目录. (III)第一章绪论 (1)1.1研究背景 (1)1.2国内外发展现状及分析 (3)1.3本文主要研究内容 (5)第二章PMSM的数学模型与控制 (7)2.1永磁同步电机的基本结构 (7)2.2 PMSM的数学模型 (8)2.3 SVPWM算法的原理与实现 (12)2.4 PMSM的矢量控制 (15)2.5本章小结 (17)第三章高频信号注入法的PMSM无位置传感器控制 (18)3.1 高频激励下的PMSM数学模型 (18)3.2 旋转高频电压注入法的PMSM无传感器控制 (20)3.3 脉振高频电压注入法的PMSM无传感器控制 (23)3.3.1 脉振高频电压注入法的基本原理 (23)3.3.2 基于跟踪观测器的转子位置估计方法 (25)3.3.3 基于PLL转子位置估计方法 (26)3.4 转子极性判断 (28)3.5 本章小结 (30)第四章高频注入法的Simulink仿真 (32)4.1 基于SVPWM的FOC控制算法仿真 (32)4.1.1 SVPWM算法仿真模块 (32)4.1.2 基于SVPWM的FOC控制算法仿真 (35)4.2旋转高频电压注入法系统仿真 (37)4.3脉振高频电压注入法系统仿真 (41)4.4 两种高频注入法的比较 (43)4.5 本章小结 (43)第五章PMSM无传感器矢量控制系统设计 (45)5.1 系统硬件结构 (45)5.1.1 主控制芯片 (46)5.1.2 电源电路 (46)5.1.3 IPM功率电路 (48)5.1.4 信号采集电路 (49)5.1.5 通信电路 (51)5.2 系统软件结构 (51)5.2.1 主程序设计 (52)5.2.2 中断子程序设计 (52)5.2.3 SVPWM程序设计 (53)5.2.4 PID程序设计 (54)5.2.5 脉振高频注入法检测转子位置程序设计 (55)5.3 基于高频注入法的无位置传感器永磁同步电机矢量控制系统试验 (56)5.4本章小结 (60)结论与展望 (61)参考文献 (63)攻读硕士学位期间取得的研究成果 (67)致谢 (68)第一章绪论第一章绪论1.1研究背景能源一向是人类生活、工业生产必不可缺的物质根本。

Sensorless Maximum Power Point Tracking Control in Wind Energy Generation using PMSG

Indian Institute of Technology Kanpur, U.P. - 208016, India Email: suresh snv@yahoo.co.in
Partha Sarathi Sensarma, Member, IEEE Department of Electrical Engineering Indian Institute of Technology Kanpur, U.P. - 208016, India Email: sensarma@iitk.ac.in

power ﬂow control. In this paper the control of the active rectiﬁer is presented. The load is assumed to be perfect sink for energy generated by the WECS. The application of ‘T’ type ﬁlter topology for wind energy applications is also proposed which is shown in Fig.2. Here, Vabc,t, Vabc,i represent three phase voltages at terminals of the generator and inverter respectively. The voltage control mode of operation is possible with this topology as the capacitor voltage manifests as a state in the dynamical equations (5-7) of the system. With this conﬁguration the current switching ripple can be effectively attenuated, and hence the electromagnetic torque oscillations. Further, it attenuates the voltage switching harmonics at the terminals of generator and hence the voltage and dielectric stresses due to switching harmonics. The disadvantages with these stresses are discussed in [10].

变频器的控制模式 VF 矢量控制

PUBLICMotor Control ModesVariable Frequency DrivesBasic Control TypesVolts/Hertz Control(V/Hz)Sensorless Vector Control (SVC )Flux Vector Control (FVC)Field Oriented Control (FVC/FOC)PWM AC Drive Block DiagramPower Conversion Unit (PWM)This figure shows a block diagram of the power conversion unitin a PWM drive. In this type of drive, a diode bridge rectifierprovides the intermediate DC circuit voltage. In the intermediateDC circuit, the DC voltage is filtered in a LC low-pass filter.Output frequency and voltage is controlled electronically bycontrolling the width of the pulses of voltage to the motor..Motor Current▪Flux Producing Component -Id ▪Torque Producing Component -Iq ▪Total Current (FLA)-ItSo what is the relationship?22IqId It +=IdLoad 1Load 2IqItItVolts/Hertz Control▪Volts/Hertz control is a basic control method▪Requires the least setup▪Varies the voltage and frequency at the outputof the drive▪Mainly used for fan and pump applications▪Most commonly used motor control modeDrive maintains a linear relationship between Voltage & Frequency460 Vac / 60 Hz =7.667 V/Hz230 Vac / 60 Hz =3.833 V/HzCustom V/Hz or Pump/Fan curveV/HZ Controlpoor torque at low speedsPer Unit TorqueSensorless Vector▪Sensorless Vector does …▪uses the V/Hz core▪provides better breakaway torque than V/Hz▪provides better torque throughout the speed range than V/Hz▪requires an autotune procedure to be performed on the motor▪Sensorless Vector does not…▪require a feedback device▪regulate torque▪regulate speedSensorless VectorSensorless Vector Speed vs. Torque▪Flux Vector control provides more precise speed and torque control with dynamic response using a voltage angle and a voltage magnitude.▪Flux current and torque are independently controlled and speed is indirectly controlled by a torque reference signal.▪Used when high performance speed regulation or torque regulation is required.Sensorless Flux VectorFlux Vector Drive OperationField Oriented Control•Field Oriented Control drives provide the best speed and torqueregulation available for AC motors by controlling both the flux andtorque components of the motor.•It provides DC like performance for AC motors, and is well suited for typical DC retrofit applications.•All current Rockwell Automation / Allen-Bradley products (FOC and FVC)Vector Vs Field Oriented Control▪Vector Control▪Acknowledges that motor current is the vector sum of thetorque and flux currents and uses this information toprovide better control of motor speed/torque.▪Field Oriented Control▪The ability to independently control the flux and torque in amotor for the purpose of accurate torque and powercontrol.▪Force Technology uses patented, high bandwidth currentregulators in combination with an adaptive controller, toseparate and control the motor flux and torque.Force Technology w/FeedbackForce Technology -SensorlessFOC Drive OperationAbove is a plot of a drive using the Sensorless version of Force Technology. Notice that thetorque output is consistent from no load to full load over a very wide speed range.Performance ComparisonTorque and Speed ControlTorque per Ampere ComparisonThe result is that a motor run at low loads will dissipate higher losses when controlled by aVolts/Hertz drive. At slower speeds, this could cause unnecessary motor overheating.PUBLIC Questions。

Realization of sensorless vector control system based on MRAS with DSP

Proceedings of the 27th Chinese Control ConferenceJuly 16-18, 2008, Kunming,Yunnan, ChinaRealization of Sensorless Vector Control System Based onMRAS with DSPHuang Yushui, Liu DanDepartment of Electrical and Automatic Engineering,Nanchang University,Nanchang 330031, P. R. ChinaE-mail: huangyushui@Abstract: Sensorless vector control of an induction motor drive essentially means vector control without any speed sensor. It is possible to estimate the speed signal from machine terminal voltages and currents with the help of a DSP. Sensorless vector controlled drives are commercially available at this time. A sensorless vector control system based on model referencing adap-tive system (MRAS) is introduced. The calculation model of rotor flux was analyzed. The sensorless vector control system was realized based on the chip of TMS320LF2407A. The simulation and test result verified that the method of sensorless vector control is feasible and valid.Key Words: MRAS, Sensorless vector control, Speed estimation1INTRODUCTIONIn the field of AC machines and their system, the speed sensor is usually used to the closed-loop control system for the realization of precise control. Not only the speed sensor increased the system cost, but reduced the system stability and the reliability. Moreover the speed sensor’s work precision is influenced by outside factor and other working conditions. These questions has been limited the application in the field of AC machines and their system, therefore the sensorless vector control sys-tem research and the application receive more and more attention.According to the three-phase asynchronous machines model and the vector control principle, the model refer-encing adaptive system(MRAS) is used to carry on the calculation to the rotor speed, the SVPWM control is carried by the control circuit based on TMS320 LF2407A to the asynchronous machines.According to the system software simulation as well as the system hardware debugging, the result has confirmed that the method of sensorless based on MRAS is feasible and the valid.2SENSORLESS VECTOR CONTROL STRATEGYThe invention of vector control and the demonstration that an induction motor can be controlled like a sepa-rately excited dc motor, brought a renaissance in the high-performance control of ac drives. Undoubtedly, vector control and the corresponding feedback signal processing, particularly for modern sensorless vector control, are complex and the use of powerful micro-computer or DSP is mandatory.2.1Rotor flux estimationThe rotor flux estimation may obtain through the rotor flux current model and the rotor flux voltage model.Rotor flux current model: i sĮand i sȕ are obtained in stationary reference frame according to the actual survey three-phase stator current through 3/2 transfor-mation. The asynchronous machines rotor flux current model equations are obtained according to the mathe-matical model [1]in stationary reference frame. The equations are:)(11ψψβααωirrsmrirTiLT p−+=(1))(11ψψαββωirrsmrirTiLT p++=(2) Where:ȌȖĮi isĮ-axis rotor flux linkage in rotor flux current model;ȌȖȕiis ȕ-axis rotor flux linkage in rotor flux current model;L m is mutual inductance;Ȧ is the rotor speed;R r is rotor resistance;T r =L r/R r is the rotor current time constant.The rotor flux current model in stationary reference frame is built based on the MATLAB software as shown in Fig. 1.Fig. 1 Rotor flux current modelWhere:is_abc is the surveyed three-phase stator current;psir_Į is Į-axis rotor flux linkage;psir_ȕ is ȕ-axis rotor flux linkage.Rotor flux voltage model: the stator flux and the rotor flux are calculated according to the actual voltage and current signal. Then the asynchronous machines rotor flux voltage model equations are obtained according to the mathematical model [1] in stationary reference frame. The equations are:])([iLiRuLLsssssmrvrdtαααασψ−−=³(3)691692])([i L i R u L L s s s s s mr v r dt ββββσψ−−=³ (4)where:ȌȖĮȞis Į-axis rotor flux linkage in rotor flux voltage model;ȌȖȕȞis ȕ-axis rotor flux linkage in rotor flux voltage model;u s Įis Į-axis static voltage; u s ȕis ȕ-axis static voltage; L r is rotor inductance; L s is stator inductance R s is stator resistance; ı=1-L m ²/(L s ×L r ) .The rotor flux voltage model in stationary reference frame is built based on the MATLAB software as shownin Fig. 2.Fig. 2 Rotor flux voltage modelWhere: u A , u B , u C is the surveyed three-phase stator volt-age.2.2Model referencing adaptive system (MRAS) [2-6] The speed can be calculated by the model referencing adaptive system (MRAS), where the output of a refer-ence model is compared with the output of an adjustable or adaptive model until the errors between the two mod-els vanish to zero. Consider the voltage model’s sta-tor-side equation by (3) and (4), which are defined as a reference model. The model calculates the flux vector signals, as indicate. The current model flux equations, (1) and (2), are defined as an adaptive model. This model can calculate fluxes from the input stator currents only if the speed signal Ȧis known.In designing the adaptation algorithm for MRAS, it is important to take account of the overall stability of the system and to ensure that the estimated speed will con-verge to the desired value with satisfactory dynamic characteristics. Using Popov’s criteria for hyperstability [7]for a globally asymptotically stable system, we can derive the following relation for speed estimation:))((ψψψψωαββαv r i r v r i r i p r s kk −+= (5)3SIMULATION ANALYSISThe system simulation model is built based on the soft-ware of MATLAB according to preceding text analysis.The IGBT inverter module, vector control module, speed estimation module are included in the systemsimulation model as shown in Fig.3.Fig. 3 System simulation modelThe system parameter establishment is: Three-phase asynchronous machines U N =380V , f =50Hz, P =2,R s =0.085ȍ, L ls =0.53mH, R r =0.225ȍ,L lr =0.53mH, L m =3.5mH, J =1.562kg.m ²; In the simulation system, the surveyed rotor speed value is replaced by calculated rotor speed Ȧ', the control effect based on the MRAS sensorless vector control system can be observed, simu-lation result as shown in Fig.4.a. Surveyed rotor speedb. Calculated rotor speed Fig. 4 Simulation r esultAccording to Fig.4, it can be see that the difference be-tween surveyed rotor speed and calculated rotor speed are very small, surveyed rotor speed relative smoother, as a whole calculated rotor speed can track surveyed rotor speed. Moreover, through the comparisons of fig.4, may discover: The machine from starts to stably, in the entire process calculated rotor speed and surveyed rotor speed both continuously maintained accurately and fast synchronized following; This had proven that it is feasi-ble for the three-phase asynchronous machines sensor-less vector control based on MRAS.4CONTROL CIRCUIT4.1System structureSystem structure as shown in Fig.5:693Fig. 5 System str uctur eThe overall system is a AC-DC-AC circuit, mainlycomposed by the main circuit, the protection circuit, the control circuit. The main circuit is composed by the un-controlled rectification bridge, the capacity filter circuit, the IPM inversion circuit and the drive circuit. The IPM inversion circuit carries on the inversion after rectifica-tion DC voltage, obtains three-phase AC which has ad-justable voltage and frequency to supplies machine. The DSP chip TMS320LF2407A is taken as a core in the control circuit, and constituted formidable numeral vec-tor control system. The PWM signal is controlled by the DSP chip, which carries on the vector operation to sur-vey the stator current and voltage. This system has owes the voltage examination circuit, the overvoltage exami-nation circuit as well as failure detection circuits and so on. This system movement is safer reliably.4.2System software designsThe system software is composed by the master pro-gram and the PWM interrupt service subroutine. The software and the hardware initialization work is com-pleted by the master program (Fig.6). The master pro-gram is also responsible for the parameter, owes the voltage protection, overvoltage protection, overflowcurrent protection.Fig. 6 The master programThe system software used DSP to interrupt INT1.INT1 is the timer T 1 cycle interrupt. The PWM interrupt ser-vice subroutine (Fig.7) mainly completes some timely processing work, including electric current, voltage sig-nal sampling, coordinate transformation, speed estimate and adjustment algorithm, SVPWM wave productionand so on.Fig. 7 PWM interrupt service subroutine4.3Experiments analyzesThe system hardware debugging is carried based on TMS320LF2407A in IMCD2407 platform. The IMCD2407 platform is offered by Intelligent Motion Digital Signal Processing Co,.LTd. It may obtain the reference voltage U ref of modulation wave in the View/Graph window of IMCD2407 platform. When modulation percentage M =0.5 and 1, the reference volt-age U ref of modulation wave in sampling period Ts asshown in Fig. 8.a.M=0.5b.M =1Fig. 8 Modulation waveThe above test result conforms to the SVPWM voltagemodulation wave of the SVPWM theoretical analysis, proved this article uses the three-phase asynchronous machines sensorless vector control method in the prac-tical application based on MARS is feasible and valid.5CONCLUSIONA method based on the MRAS sensorless vector control is proposed in the paper. The method establishes the self-adaptive model according to the rotor flux current model and the rotor flux voltage model, carries on the calculation to the rotational speed and realizes the reac-tion control. The control system based on the MRAS sensorless vector control has many merits such as the structure is simple, the load computation is small, reali-zation is easy and the machine parameter is little af-fected.The TMS320LF2407A is taken as the core structure of system to realize the sensorless vector control based on MRAS, the simulation and test result verified that the method of sensorless vector control is feasible and valid. REFERENCES[1]GAO J D, WANG X H, LI F H. Analysis of AC machinesand their system (second edition) [M]. Beijing: Tsinghua University Press, 2005. [2]BIMAL K B.Modern Power Electronics and AC Drives[M].Prentice Hall PTR Prentice-Hall, 2002.[3]RAJASHEKARA K, KAWAMURA A, MATSUSE K .et al.Sensorless Control of AC Drives, IEEE Press, NY, 1996. [4]LIN F J , WAI R J, KUO R H et al. A comparative study ofsliding model and model reference adaptive speed observers for induction motor drive. Electric Power Systems Re-search[J] ,1998,(44): 163-174.[5]YANG G, CHEN B S. Review the Methods for the SpeedSensor-less Control of Induction Motor [J]. Electrical drive 2001(3):3-8.[6]LIAO Y, ZHANG F R. Research of Sensorless Vector Con-trol System and Speed Estimation [J]. Transactions of China electrotechnical society 2004(2):36-40.[7]LANDAU Y D, Adaptive Control-The Model ReferencingApproach, Marcel Dekker, 1979.694。

感应电机二阶滑模次优算法定子磁链观测器设计

感应电机二阶滑模次优算法定子磁链观测器设计潘月斗;陈泽平;郭映维【摘要】提出了基于二阶滑模次优算法的感应电机定子磁链观测方法,设计了定子磁链观测器,并应用到感应电机直接转矩控制中.本文设计的磁链观测器,通过准确的跟踪电流及其变化率,从而实现对转子磁链的准确估算,然后利用转子磁链与定子磁链的关系,估算出定子磁链.由于本文设计的定子磁链观测器是一个多输入多输出(MIMO)系统,稳定性分析非常复杂,为此将磁链估算误差的微分看作扰动处理,从而将MIMO的观测器模型分解成两个独立的单输入单输出(SISO)系统,简化了稳定性分析.将该观测器用于感应电机直接转矩控制中,达到了很好的控制效果.仿真和实验验证了该方法的有效性.【期刊名称】《控制理论与应用》【年(卷),期】2015(032)005【总页数】5页(P641-645)【关键词】感应电机;二阶滑模;次优算法;电流观测;磁链观测;直接转矩控制【作者】潘月斗;陈泽平;郭映维【作者单位】北京科技大学自动化学院,北京100083;北京科技大学钢铁流程先进控制教育部重点实验室,北京100083【正文语种】中文【中图分类】TM343感应电机被广泛应用于工农业生产、国防、科技及社会生活等各个方面,随着直接转矩控制和矢量控制技术的出现,使其逐渐进入了伺服控制领域[1].相对于矢量控制,直接转矩控制方法直接把转矩作为被控量,并由电流和定子磁链估算,无需进行磁场定向和矢量变换,更为简单和实用,具有快速的动态响应能力[2].直接转矩控制中,定子磁链观测值的精确度直接影响控制效果[3].定子磁链观测的基本方法有电压模型法和电流模型法.电压模型法结构简单,观测时仅需确定定子电阻.但是电压模型法在运算过程中需开环积分(纯积分),微小的直流偏移误差和初始值误差都将导致积分饱和[4].电流模型法可解决电压模型积分漂移和无法建立初始磁链的问题,但观测精度与转速相关,易受电动机转速变化的影响[5].为了更好的观测磁链,已提出了很多方法,如滑模变结构方法[6–7]、自适应方法[8]、卡尔曼滤波器方法[9–10]、神经网络方法[11]等.相比其他方法,滑模变结构方法对系统的不确定性因素具有较强的鲁棒性和抗干扰性,同时控制设计简单,物理上易于实现,因此得到广泛应用.但是在实际应用中,滑模变结构控制也存在一些问题,其中最主要的是抖振现象[12].近些年提出的高阶滑模控制理论[13],是对传统滑模控制理论的进一步推广.相比传统滑模,高阶滑模不仅保持了传统滑模的优点,同时抑制了系统的抖振,除去了相对阶的限制,并且提高了控制精度.二阶滑模控制是目前应用最广泛的高阶滑模控制方法,因为它的控制器结构简单且所需要的信息不多.二阶滑模控制中常见的4种算法有:twisting(螺旋)算法、sub-optimal(次优)算法、prescribed convergence law(给定收敛律)算法和Super-Twisting(超螺旋)算法.本文设计了一种基于二阶滑模次优算法的感应电机定子磁链观测器.将磁链估算误差的微分看作扰动处理,从而将MIMO的观测器模型分解成两个独立的SISO系统,简化了稳定性分析.将该观测器用于感应电机直接转矩控制中,达到了很好的控制效果.仿真及实验结果验证了该方法的有效性.设感应电机的磁路是线性的,忽略铁损的影响,在静止坐标系(α–β)下,感应电机的数学模型的状态方程为[14]δ=ηRs+Lmλθ;isα,isβ,usα,usβ,ψrα,ψrβ分别为α轴和β轴的定子电流、定子电压和转子磁链;ωr为转子电角速度;Ls,Lr,Lm分别为定子电感、转子电感和定转子间互感;Rs,Rr分别为定子电阻和转子电阻.定子磁链和转子磁链存在如下关系[15]:设计如下感应电机转子磁链观测器:其中:分别为定子电流和转子磁链的状态估计变量,vα和vβ为控制信号,分别为α轴和β轴的定子电流观测误差.定子电压和定子电流usα,usβ,isα,isβ都是可以检测到的,定子电压是原实际系统(感应电机)的输入量,定子电流可作为原实际系统的输出量;针对此观测器而言,定子电流检测量isα,isβ作为给定输入量(也作为干扰输入的一部分),定子电压检测量usα,usβ以及转子电角速度看作干扰输入的一部分;,作为观测器的反馈量.式(1)减式(2),可以得到定子电流和转子磁链观测误差方程电流观测误差方程写成如下形式:由式(5)可知,电流误差方程系统相对于控制信号v是1阶系统,因此可以采用二阶滑模控制,设计控制信号v,使得滑模变量s趋于零,并保持二阶滑动模态,即s==0.如果选取s=,采用二阶滑模控制,即可使得=0.二阶滑模次优算法(sub-optimal)形式如下:其中:s∗是最近的时间内,=0时s的值;k1,k2为控制参数,令s(t,x)=0为所定义的滑模面,控制目标是使系统的状态在有限时间内收敛到滑模流形s== 0.选取滑模面s=设计如下控制律:其中:对于式(5),将看作扰动处理,可将其分成α轴和β轴方向两个独立的SISO(单入单出)系统,如下:文献[16]给出了次优算法有限时间收敛的充分条件:其中Km,KM,C满足如下条件:对于本文设计的观测器系统,α轴方向分析如下:上式对时间求导,可得系统有限时间收敛的充分条件[16]如下:如果参数kα1,kα2满足式(9),则系统必能在有限时间内到达滑模面满足如下条件: β轴方向的稳定性分析同上.利用转子磁链观测器估算得到的转子磁链和定子电流,可估算定子磁链基于二阶滑模次优算法的感应电机定子磁链观测器系统框图如图1所示.为了检验所设计的基于二阶滑模次优算法的感应电机定子磁链观测器的有效性,进行了MATLAB仿真与实验.电机参数为:额定电压UN=220V,定子电阻Rs=94Ω,转子电阻Rr=83.9Ω,定子自感Ls= 5.387H,转子自感Lr=5.387H,互感Lm=5.082H,转动惯量J=0.105kg·m2.观测器控制参数为:kα1=kβ1=10,kα2=kβ2=5.电机施加220V,15Hz的三相交流电,在开环下空载运转,4s时,施加3N·m负载转矩.仿真时间7s,仿真结果如图2–5所示.从图3和图4可以看出,观测电流误差及其微分(由于实际对磁链观测误差有影响的是,所以图4实际是δ的值),在一定时间内渐近趋于0,从而说明了给二阶滑模次优算法控制的有效性.从图5可以看出,观测磁链在一定时间内达到稳定.为了验证基于二阶滑模次优算法的感应电机定子磁链观测器的有效性,将其应用到感应电机直接转矩控制中.电机参数与开环时一样,定子磁链给定值ψ=1Wb,给定转速600r/min.转速调节器采用PID控制,其中比例系数KP=10,积分系数KI= 0.001,微分系数KD=0.5.仿真时间20s,仿真结果如图6所示.为了验证二阶滑模次优算法定子磁链观测器的实际可行性,利用“电力电子与电气传动综合实验台”进行实验.实验台组成包括:功率挂箱、主控挂箱、加载控制箱、电动机、上位机,如图7所示.实验电机为鼠笼式三相异步电动机,参数与仿真时所用电机参数相同.转速给定值600r/min,实验结果如图8所示.从仿真和实验结果可以看出,二阶滑模次算法定子磁链观测器能够很好的观测定子磁链,电机转速也最终稳定在了给定值600r/min,从而证明了本文所提出的基于二阶滑模次算法的感应电机定子磁链观测器的实际可行性.本文提出的二阶滑模次优算法定子磁链观测器,首次将二阶滑模次优算法应用到感应电机定子磁链观测器设计中,并将此观测器应用到直接转矩控制中.从仿真和实验结果可以看出,该观测器能够准确的估算定子磁链,将其用于感应电机直接转矩控制中,也达到了很好的控制效果.仿真实验验证了该方法的有效性.潘月斗(1966–),男,博士,副教授,目前研究方向为交流电动机智能控制理论研究及高速高精交流电动机驱动系统的计算机数字控制系统设计,E-mail:****************;陈泽平(1989–),男,硕士研究生,目前研究方向为电气传动及自动化,E-mail:**********************;郭映维(1990–),男,硕士研究生,目前研究方向为异步电机控制理论及数字化设计,E-mail:*****************.【相关文献】[1]PELLEGRINO G,GUGLIELMI P,ARMANDO E,et al.Selfcommissioning algorithm for inverter nonlinearity compensation in sensorless induction motor drives[J].IEEE Transactions on Industry Applications,2010,46(4):1416–1424.[2]张细政,王耀南,袁小芳,等.基于滑模与自适应观测器的感应电机非线性控制新策略[J].控制理论与应用,2010,27(6):753–760.(ZHANG Xizheng,WANG Yaonan,YUAN Xiaofang,et al.New nonlinear controller forinduction motor based on sliding-mode control and adaptive observer[J].Control Theory&Applications,2010, 27(6):753–760.)[3]张猛,肖曦,李永东.基于扩展卡尔曼滤波器的永磁同步电机转速和磁链观测器[J].中国电机工程学报,2007,27(36):36–40.(ZHANG Meng,XIAO Xi,LI Yongdong.Speed and flux linkage observer for permanent magnet synchronous motor based on EKF[J]. Proceedings of the CSEE,2007,27(36):36–40.)[4]李红,罗裕,韩邦成,等.带通滤波器法电压积分型定子磁链观测器[J].电机与控制学报,2013,17(9):8–16.(LI Hong,LUO Yu,HAN Bangcheng,et al.Voltage integral model for stator flux estimator based on band-pass filter[J].Electric Machines and Control,2013,17(9):8–16.)[5]SPICHARTZ M,STEIMEL A,Stator-flux-oriented control with high torque dynamics in the whole speed range for electric vehicles[C] //Emobility-Electrical Power Train.New York:IEEE,2010:1–6.[6]LI J C,XU L Y,ZHANG Z.An adaptive sliding-mode observer for induction motor sensorless speed control[J].IEEE Transactions on Industry Applications,2005,41(4):1039–1046.[7]REHMAN H.Elimination of the stator resistance sensitivity and voltagesensorrequirementproblemsforDFOcontrolofaninductionmachine[J].IEEE Transactions on Industrial Electronic,2005,52(1): 263–269.[8]刘艳红,霍海娟,楚冰,等.感应电机转矩跟踪无源控制及自适应观测器设计[J].控制理论与应用,2013,30(8):1021–1026.(LIU Yanhong,HUO Haijuan,CHU Bing,et al.Passivity-based torque tracking control and adaptive observer design of induction motors[J].ControlTheory&Applications,2013,30(8):1021–1026.)[9]BARUT M,BOGOSYAN S,GOKASAN M.Speed-sensorless estimation for induction motors using extended Kalman filters[J].IEEE Transactions on IndustrialElectronics,2007,54(1):272–280.[10]HAQUE M E,ZHONG L,RAHMAN M F.A sensorless initial rotor position estimation scheme for a direct torque controlled interior permanent magnet synchronous motor drive[J].IEEE Transactions on Power Electronics,2003,18(6):1376–1383.[11]SIMOES M G,BOSE B K.Neural network based estimation of feedback signals for a vector controlled induction motor drive[J].IEEE Transactions on IndustryApplication,1995,31(3):620–629.[12]YOUNGK D,UTKIN V I,OZGUNER U.A control engineer’s guide to sliding mode control[J].IEEE Transactions on Control Systems Technology,1999,7(3):328–342.[13]FRIDMAN L,LEVANT A.Higher order sliding modes as a natural phenomenon in control theory[J].Robust Control via Variable Structure and LyapunovTechniques.Heidelberg,Berlin:Springer,1996: 107–133.[14]LI J,XU L,ZHANG Z.An adaptive sliding-mode observer for induction motor sensorless speed control[J].IEEE Transactions on Industry Applications,2005,41(4):1039–1046. [15]MITRONIKASED,SAFACASAN.Animprovedsensorlessvectorcontrol method for an induction motor drive[J].IEEE Transactions on Industrial Electronics,2005,52(6):1660–1668.[16]LEVANT A.Principles of 2-sliding mode design[J].Automatica, 2007,43(4):576–586.。

一种无传感器PMSM效率优化控制方法

一种无传感器PMSM效率优化控制方法吕一松;李旭春;贺骥;吴正礼【摘要】提出了一种新的永磁同步电机无位置传感器正弦波控制方法。

该方法不同于传统的转子位置估算思路,而是利用以电流为基准的永磁同步电机矢量图,得到改进的效率最优策略,进而提出“强制同步-效率优化”控制方法。

文章介绍了新方法的实现方案,并借助永磁同步电机功角特性曲线证明了方法的稳定性。

方法思路清晰,复杂度低;参数适应性强;硬件系统简单,易于实现。

实验结果证明了该方法的可行性。

【期刊名称】《电工技术学报》【年(卷),期】2010(000)006【总页数】6页(P12-17)【关键词】180°直流无刷电机;正弦波永磁同步电机;无传感器;强制同步;d轴电流【作者】吕一松;李旭春;贺骥;吴正礼【作者单位】清华大学自动化系,北京100084【正文语种】中文【中图分类】TP3411 前言永磁同步电机（Permanent Magnet Synchronous Motor，PMSM）在转子轴上往往安装有位置传感器（如霍尔位置传感器、编码器、测速发电机等），传感器的使用不仅增加了成本，降低了系统的可靠性，而且受到诸如温度、湿度和振动等条件的限制，使之不能广泛适用于各种场合。

为了克服传感器给系统带来的缺陷，学者们进行了无传感器永磁同步电机控制研究，比较典型的控制方法有：直接计算法、模型参考自适应法、观测器法、高频注入法以及基于人工智能的方法等[1]。

然而，为达到电机的最优运行效率，上述方法无一例外都需对电机进行三相-两相变换（3-2变换）、估算转子位置，进而跟随转子位置输出适当的同步电压或电流（本文称其为“跟随同步”控制方法）。

但是，这种跟随同步方法的3-2变换和估算转子位置的过程，要在一个或几个 PWM周期（时间一般小于200µs）内完成，有时还需使用迭代算法，程序编写复杂，计算量大，硬件要求高。

同时，转子位置估计算法中用到的电机参数如电阻、磁通对温度和电流等工作环境非常敏感，算法鲁棒性差。

Sensorless+Control+of+PMSM+Based+on+Adaptive+Sliding+Mode+Observer

ˆα + f (iα ) < 0 ΔA ⋅ iα i ˆβ + f (iβ ) < 0 ΔA ⋅ iβ i
2 MODELS AND OBSERVER
In stationary (α , β ) reference frame, the mode for PMSM is characterized by (1)
diα R 1 uα = − iα + eα + dt L L L diβ R uβ 1 = − iβ + eβ + dt L L L eα = −λ0ω e sin(θ e )
speed can be derived as
& ≈ λ B( S sin θ ˆ − S cosθ ˆ) ˆ ω e 0 1 2
And (10) can be rewritten as
(10)
− R L . Because the variation of L
& ≈ λ B ⋅ [(i ˆ − (i ˆ] ˆα − iα ) sinθ ˆβ − iβ ) cosθ ˆ ω e 0
s e e
ˆα di R 1 ˆα + 1 e ˆα + uα + f (iα ) = (− + ΔA)i dt L L L ˆβ di R 1 ˆβ + 1 e ˆβ + uβ + f (iβ ) = (− + ΔA)i dt L L L
（2） Where superscript “ ^ ” represents the estimated quantities, “—”represents the error quantities, ΔA is the variation of

Sensorless vector control of induction motors at very low speed using a nonlinear inverter model and

Sensorless Vector Control of Induction Motors at Very Low Speed Using a Nonlinear Inverter Model and Parameter IdentificationJoachim Holtz,Fellow,IEEE,and Juntao QuanAbstract—The performance of vector-controlled induction motor drives without speed sensor is generally poor at very low speed.The reasons are offset and drift components in the acquired feedback signals,voltage distortions caused by the non-linear behavior of the switching converter,and the increased sensitivity against model parameter mismatch.New modeling and identification techniques are proposed to overcome these problems.A pure integrator is employed for stator flux estima-tion which permits high-estimation pensation of the drift components is done by offset identification.The nonlinear voltage distortions are corrected by a self-adjusting inverter model.A further improvement is a novel method for on-line adaptation of the stator resistance.Experiments demonstrate smooth steady-state operation and high dynamic performance at extremely low speed.Index Terms—Induction motor,low-speed operation,parameter identification,sensorless control,vector control.I.I NTRODUCTIONC ONTROLLED induction motor drives without speedsensor have developed as a mature technology in the past few years.However,their performance at very low speed is poor.The main reasons are the limited accuracy of stator voltage acquisition,the presence of offset and drift compo-nents in the acquired voltage signals,their limited bandwidth, offsets and unbalances in the current signals,and the increased sensitivity against model parameter mismatch.These deficiencies degrade the accuracy of flux estimation at low speed.The dynamic performance of a sensorless drive then deteriorates.Sustained operation at very low speed becomes im-possible as ripple components appear in the machine torque and the speed starts oscillating,eventually leading to instable oper-ation of the system.Paper IPCSD02–025,presented at the2001Industry Applications Society Annual Meeting,Chicago,IL,September30–October5,and approved for publication in the IEEE T RANSACTIONS ON I NDUSTRY A PPLICATIONS by the Industrial Drives Committee of the IEEE Industry Applications Society. Manuscript submitted for review October15,2001and released for publication May10,2002.J.Holtz is with the Electrical Machines and Drives Group,University of Wup-pertal,42097Wuppertal,Germany(e-mail:j.holtz@).J.Quan is with the Danaher Motion Group,Kollmorgen-Seidel,Duesseldorf, Germany(e-mail:jquan@).Publisher Item Identifier10.1109/TIA.2002.800779.II.S OURCES OF I NACCURACY AND I NSTABILITYA.Estimation of the Flux Linkage VectorMost sensorless control schemes rely directly or indirectly on the estimation of the stator flux linkagevectoris the stator resistance.Time is normalizedasis the nominal stator frequency[3].The addedsymbol in (1)represents all disturbances such as offsets,unbalances,and other errors that are contained in the estimated inducedvoltage(2)where is the coupling factor of the rotorwindings,is the total leakage flux vector.The estimation of one of the flux vectors according to(1)or (2)requires performing an integration in real time.The use of a pure integrator has not been reported in the literature.The reason is that an integrator has an infinite gain at zero frequency.The unavoidable offsets contained in the integrator input then make its output gradually drift away beyond limits.Therefore,instead of an integrator,a low-pass filter usually serves as a substitute.A low-pass filter has a finite dc gain which eases the drift problem, although drift is not fully avoided.However,a low-pass filter in-troduces severe phase angle and amplitude errors at frequencies around its corner frequency,and even higher errors at lower fre-quencies.Its corner frequency is normally set to0.5–2Hz,de-pending on the existing amount of offset.The drive performance degrades below stator frequencies2–3times this value;the drive becomes instable at speed values that correspond to the corner frequency.Different ways of compensating the amplitude and phase-angle errors at low frequencies have been proposed[4]–[7].0093-9994/02$17.00©2002IEEEOhtani [4]reconstructs the phase-angle and amplitude error pro-duced by the low-pass filter.A load-dependent flux vector refer-ence is synthesized for this purpose.This signal is transformed to stator coordinates and then passed through a second low-pass filter having the same time constant.The resulting error vector is added to the erroneous flux estimate.Although the benefits of this method are not explicitly documented in [4],improved performance should be expected in an operating range around the corner frequency of the low-pass filter.With a view to improving the low-speed performance of flux estimation,Shin et al.[5]adjust the corner frequency of the low-pass filter in proportion to the stator frequency,while com-pensating the phase and gain errors by their respective steady-state values.It was not demonstrated,though,that dynamic op-eration at very low frequency is improved.Hu and Wu [6]try to force the stator flux vector onto a circular trajectory by propor-tional plus integral (PI)control.While this can provide a correct result in the steady state,it is erroneous at transient operation and also exhibits a large error at startup.A practical application of this method has not been reported;our investigations show loss of field orientation following transients.B.Acquisition of the Stator VoltagesThe induced voltage,which is the signal to be integrated for flux vector estimation,is obtained as the difference between the stator voltage and the resistive voltage drop across the ma-chine windings.When a voltage-source inverter (VSI)is used to feed the machine,the stator voltages are formed by pulse trains having a typical rise time of 2–10kV/,whereis the fundamental componentofcaused by the switching characteristics of the inverter.C.Acquisition of the Stator CurrentsThe stator currents are usually measured by two Hall sensors.They are acquired as analog signals,which are subsequently digitized using A/D converters.The sources of errors in this process are dc offsets and gain unbalances in the analog signal channels [9].After the transformation of the current signals to synchronous coordinates,dc offsets generate ac ripple compo-nents of fundamental frequency,while gain unbalances produce elliptic current trajectories instead of circular trajectories.The disturbance in the latter case is a signal of twice the fundamentalfrequency.Fig.1.Effect of a dc offset in one of the current signals on the performance of a vector-controlled drivesystem.Fig.2.Effect of a gain unbalance between the acquired current signals on the performance of a vector-controlled drive.The following oscillograms demonstrate the effect of such disturbances on the performance of a vector-controlled drive system.The respective disturbances are intentionally intro-duced,for better visibility at a higher signal level than would normally be expected in a practical implementation.Fig.1shows the effect of 5%dc offset in one of the current signals on the no-load waveform ofthe.The drive is operated is at astator frequency of 2Hz.The transformed current signals gen-erate oscillations in the torque-producing current .Resulting from this are torque pulsations of 0.06nominal value,and cor-responding oscillations in the speedsignal,where isthe power factor of the motor.Fig.2shows the same signals under the influence of 5%gain unbalance between the two current channels.Oscillations of twice the stator frequency are generated in the torque-producing current,and also in the speed signal.D.Estimation of the Stator ResistanceAnother severe issue,in addition to the integration problem and to the nonlinear behavior of the inverter,is the mismatch be-tween the machine parameters and the respective model param-eters.In particular,adjusting the stator resistanceHOLTZ AND QUAN:SENSORLESS VECTOR CONTROL OF INDUCTION MOTORS1089Fig.3.Forward characteristics of the power devices.flux estimation,and for stable operation at very low speed.The actual valueofand an averagedifferentialresistance [12].The variations with temperatureof the thresholdvoltageof about equal magnitude to all the threephases,and it is the directions of the respective phase currents that determine their signs.The device thresholdvoltage(4)where.Thesectorindicator is a unity vector that indicates the re-spectivein thecomplex plane.The locations are determined by the respective signs of the three phase currents in (3),or,in other words,by a maximumof.The referencesignalof the stator voltagevectoris less than its referencevalue,and of the resis-tive voltage drops of the power devicesthrough1090IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.38,NO.4,JULY/AUGUST2002(a)(b)(c)Fig.4.Effect at PWM of the forward voltagesuof the power semiconductors.(a)Switching state S.(c)Switching state S);the dotted linesindicate the transitions at which the signs of the respective phase currents change.Notethatis the resulting threshold voltage vector.We have,therefore,from (4),the unusualrelationshipis one parameter of the invertermodel.It is determined during a self-commissioning process from the distortions of the reference voltagevectorand of the reference voltage vector are acquiredwhile using the current controllers to inject sinusoidal currents of very low frequency into the stator windings.In such condi-tion,the machine impedance is dominated by the stator resis-tance.The stator voltages are then proportional to the stator cur-rents.Any deviation from a sinewave of the reference voltages that control the pulsewidth modulator are,therefore,caused by the inverter.As an example,an oscillogram of the distorted referencevoltagewaveformsand ,measured at sinusoidal currents ofmagnitude ,is shown in Fig.7.The amplitude of the fundamental voltage is very low which is owed to the low frequency of operation.The distortions of the voltage waveforms in Fig.7are,therefore,fairly high.They are predominantly caused by the dead-time effect of the ing such distorted voltages to represent the stator voltage signal in a stator flux estimator would lead to stability problems at low speed.Accurate inverter dead-time compensation [13]is,therefore,mandatory for high-performance applications.Fig.8shows the same components of the reference voltagevectoraccording to (4);the locationsofare shown in Fig.4.It follows from (4)that both the larger step change and the amplitudeofhave the magnitude4/3from thewaveformof(or )in Fig.8appears quite inaccurate.A better method is subtracting the fundamentalcomponent from,e.g.,,which then yields a square-wave-like,stepped waveform as shown in Fig.9.The fundamental component iseasily extracted from a set of synchronous samplesofby fast Fourier transform.The differential resistance of the powerdevices,in (6),es-tablishes a linear relation between the load current and its in-fluence on the inverter voltage.Functionally,it adds to the re-sistance)is estimated by an online tuning process described inSection III-D.HOLTZ AND QUAN:SENSORLESS VECTOR CONTROL OF INDUCTION MOTORS1091(a)(b)Fig.6.Effect of inverter nonlinearity.The trajectory u represents the average stator voltage (switching harmonics excluded).(a)At motoring.(b)Atregeneration.Fig.7.Effect of inverter dead time on the components of the voltage vectoruas in Fig.7;inverteroperated with dead-time compensation.C.Stator Flux EstimationThe inverter model (6)is used to compensate the nonlinear distortions introduced by the power devices of the inverter.The model estimates the stator voltagevector(8)is the estimated effective offset voltage vector,while is theestimated stator field angle.The offset voltagevectorin (7)is determined such that the estimated stator fluxvector rotates close to a circular trajectory in the steady state,which follows from (7)and from the right-hand side of (8).To enable the identificationofin (8),the stator field angle is estimatedas(9)as illustrated in the right portion of Fig.10.The magnitude of the stator flux linkage vector is then obtainedas1092IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.38,NO.4,JULY/AUGUST2002Fig.10.Signal flow graph of the inverter model and the stator flux estimator.The gain constantserve this purpose in a satisfactorymanner.The stator frequency signal is computed byis determined,for instance,with reference to[2]of the stator current,as shown in Fig.11.We haveand,consequently,.Of the superscripts,component of the vector product of the statorvoltage and current vectors.The system equation,for example given in[3],isHOLTZ AND QUAN:SENSORLESS VECTOR CONTROL OF INDUCTION MOTORS1093Fig.12.Signal flow graph of the stator resistance estimator.wherecomponent of all terms in(19)and assumingfieldorientation,,wehave10toa n d=!=wp w wp p f p p w1094IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.38,NO.4,JULY/AUGUST2002Fig.16.Identification of the stator resistance,demonstrated by a30%stepincrease of the resistance value.Fig.17.Reversal of speed between the set-point values w=60:04;torqueis constant at50%nominal value.the speed is negative.Finally,the performance of the stator re-sistance identification scheme is demonstrated in Fig.17.Thestator resistance is increased by30%in a step-change fashion.The disturbance causes a sudden deviation from the correct fieldangle,which produces a wrong value.The new value ofHOLTZ AND QUAN:SENSORLESS VECTOR CONTROL OF INDUCTION MOTORS1095 Juntao Quan was born in Jiangxi,China,in1964.Hereceived the B.Eng.degree from Jiangxi PolytechnicCollege,Nanchang University,Nanchang,China,the M.Eng.degree from Northeast-Heavy MechanicInstitute,Yanshan University,Qinhuangdao,China,and the Ph.D.degree from Wuppertal University,Wuppertal,Germany,in1983,1989,and2002,respectively,all in electrical engineering.He was an Assistant Electrical Engineer for threeyears at the Nanchang Bus Factory,Nanchang,China.From1989to1994,he was a Lecturer at YanshanUniversity.During this time,he also worked on various projects for applicationsof power electronics.In1995,he joined the Electrical Machines and Drives Lab-oratory,Wuppertal University,where he worked and studied toward the Ph.D.degree.In June2000,he joined the Danaher Motion Group,Kollmorgen-Seidel,Duesseldorf,Germany.His main interests are in the areas of adjustable-speeddrives,microprocessor-embedded real-time control,power electronics applica-tions,and advanced motion control.。

Implementation of sensorless vector control for super-high-speed PMSM of turbo-compressor

Implementation of Sensorless Vector Control for Super-High-Speed PMSM of Turbo-Compressor Bon-Ho Bae,Member,IEEE,Seung-Ki Sul,Fellow,IEEE,Jeong-Hyeck Kwon,Member,IEEE,and Ji-Seob ByeonAbstract—This paper describes the implementation of twovector control schemes for a variable-speed131-kW perma-nent-magnet synchronous motor drive in super-high-speedapplications.The vector control with a synchronous referenceframe current regulator was implemented with challengingrequirements such as an extremely low stator inductance(28Fig.3.Power circuit diagram of the proposed super-high-speed PMSM drive.and the operation of the current regulator was tested up toan excitation frequency of1200Hz.In addition,the vectorcontrol schemes with and without the discrete Hall sensors areproposed.In the case of a vector control with three discreteHall sensors,the discrete Hall sensors provide rough positioninformation with a resolution ofs.Becausethe general-purpose microprocessors cannot meet the requiredcalculation time,the TMS320VC33-150digital-signal-pro-cessor(DSP)-based digital controller was developed for theimplementation.The experimental data showed that it takesless than20BAE et al.:SENSORLESS VECTOR CONTROL FOR SUPER-HIGH-SPEED PMSM OF TURBO-COMPRESSOR813Fig.4.Block diagram of the synchronous reference frame current regulator with the inductor.(a)(b)(c)Fig.5.Current waveform of synchronous reference frame current regulator with an inductor load.(a)Current waveforms with excitation frequency of20Hz.(b)Current waveforms with excitation frequency of1200Hz.(c)Current waveforms with excitation frequency of1200Hz.(Magnitude of current reference is changed from350to200As,which is relatively large consid-ering the short PWM update period,33.33814IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.39,NO.3,MAY/JUNE2003 Fig.5shows the experimental results with the proposed cur-rent regulator.A three-phase air-core reactor is used for the test,and the inductance is set to the same value of the stator induc-tance of the PMSM,28,-phasecurrent measured by the current probe set(Tektronix CT-4,A6302and AM503),.The traces shown in Fig.5(a)present the experimental results using an excitation frequencyof20Hz.From the results shown in Fig.5(a),it can be seenthat the performance of the current regulator was degraded atthe zero crossings of the phase currents because of the effectof the dead time and the zero-current clamping even aftercareful compensation[7],[8],[10].The traces in Fig.5(b)show the experimental results with an excitation frequency of1200Hz.Because the dead-time effect and the zero-currentclamping effect are reduced at high frequency,the currents arewell controlled sinusoidally without degradation.However,themeasured currents in Fig.5(a)and(b)show large ripplesdue to the very small load inductance.The delay in the sampledcurrentto200A.The bottom trace in Fig.5(c)shows the magnitude of thecurrent vector,,30from the input of the latestangle pulse to the sampling point was measured by the pro-grammable logic device.The speed of the motor,BAE et al.:SENSORLESS VECTOR CONTROL FOR SUPER-HIGH-SPEED PMSM OF TURBO-COMPRESSOR815Fig.9.Block diagram of sensorless vector control scheme without position sensor.Based on the assumption that the rotor speed does not change between the pulses,the rotor angle at the samplingpoint s at thenormal speed and the speed does not change rapidly,reasonably accurate speed information can be calculated by the algorithm.The test results with the proposed vector control scheme using the angle information of the Hall-effect sensors are shown in Fig.7.For the experiment,three discrete Hall-effect sensors with a sensing magnet were installed in the PMSM and a special digital logic circuit was implemented to measuretheangleof Fig.8using the programmable logic device.From top to bottom,the traces showthe-axis current ,the motor phasecurrent-axis current are shown.For the test,the extrapolation of the rotor angle has not been carried out at low speed,where the speed information is not reliable.Therefore,the discontinuous angle information deteriorated the performance of the current regulator.Fig.7(b)shows the acceleration from 17000to 20000r/min.Because precise angle information is available using the extrapolation,the vector control scheme provides the effective current and speed regulation.Compared to the conventional position sensors,the discrete Hall sensors were much more reliable in a super-high-speed op-eration.However,the installation of the Hall sensors and the sensing magnet limits the mechanical design,and the sensorless control is a better solution for super-high-speed applications.In the case of the turbo-compressor application,the installation of a sensing magnet even causes difficulties in the aerodynamic design.V .S ENSORLESS V ECTOR C ONTROL S CHEME W ITHOUTANY P OSITION S ENSOR Fig.9shows a block diagram of the sensorless vector control scheme without the position sensor.In the diagram,the feedfor-wardtermsis the error between the real rotorangle-axis voltageerror,which has to be compensated for bytheis small,the output voltage ofthe(11)In the proposed estimator in Fig.9,the error signal of (11)is processed by the PI compensator to derive the rotor speed and the rotor angle is calculated by integrating the estimated speed.In the conventional method [3],a differentiation process816IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.39,NO.3,MAY/JUNE2003Fig.10.Frequency pattern for constant current control with pre-patternedfrequency.is used to calculate the speed but this makes the system vul-nerable to measurement noise.The experimental study revealsthat the proposed estimator provides a very accurate and robustspeed information for the application.However,at the zero andlow speed,the back-EMF voltage is not high enough for theproposed vector control.Hence,for the initial alignment andstarting from zero speed,the current was controlled with a con-stant magnitude using a pre-patterned angular frequency.In ad-dition,the angle for the synchronous reference frame was calcu-lated by integrating the frequency.Fig.10shows the frequencypattern for the initial alignment and starting.As shown in step Iof Fig.10,the initial value of the frequency pattern is set to asmall but constant speed for the initial rotor alignment.After the alignment,according to the speed pattern in step II,themotor is accelerated up to the threshold speed.Over thethreshold speed,the motor is then controlled by the proposedsensorless control and the speed is estimated by the proposedestimator shown in Fig.9.VI.E XPERIMENTAL R ESULTS W ITH S ENSORLESS C ONTROLThe experimental results with the proposed sensorless con-trol are shown in Figs.11–14.Fig.11shows the starting char-acteristics from zero speed to20000r/min.The PMSM wasaccelerated by the rotating current vector with the precalcu-lated frequency pattern shown in Fig.10.In order to align therotor,the current vector was rotated with the starting frequency-axis current tracks the command with a smallripple current,which is caused by the dead time and zero-currentclamping effects.Because the current control bandwidth was setlow for the sensorless algorithm,the actual current tracks thecommand with a delay.The bottom trace shows the measuredphase current,,which was measured by a current probeset(Tektronix AP504CX and AM503B).Due to the extremelysmall stator inductance,the fundamental current was accompa-nied by a significant ripple current,which was caused by the15-kHz PWM switching.Fig.13(b)shows the magnified wave-forms of the last two periods in Fig.13(a).The traces show thatthe phase current is controlled sinusoidally by the precise sam-pling of the fluctuating current.In Fig.13,the measured phase current shows a rela-tively large current ripple,which can increase the temperatureof the rotor.Because a high temperature can degrade theBAE et al.:SENSORLESS VECTOR CONTROL FOR SUPER-HIGH-SPEED PMSM OF TURBO-COMPRESSOR817(a)(b)Fig.13.Current waveforms with the proposed sensorless control (acceleration from 58000to 60000r/min).(a)Current waveforms.(b)Magnified figures of the last two periods of current waveforms in (a).mechanical stability of the rotor,it is desirable to reduce the cur-rent ripple.Fig.14shows the current waveforms with a28-,shows that the current ripple is reducedremarkably by the external inductor.As shown in Fig.14,the current ripple can be easily reduced by the external inductor.However,because the adoption of an external inductor requires a higher inverter output voltage,the value of the inductor can be determined by a tradeoff between the current waveform and the inverter output voltage capability.VII.C ONCLUSIONThe development of a super-high-speed drive for a 131-kW 70000-r/min PMSM for use in a turbo-compressor wasdis-(a)(b)Fig.14.Current waveforms with an external inductor of 28 H (at speed of 65000r/min).(a)Current waveforms.(b)Magnified waveforms.cussed.The synchronous reference frame current regulator was implemented with challenging requirements such as a low stator inductance(28818IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS,VOL.39,NO.3,MAY/JUNE2003[2]W.L.Soong,G.B.Klima,R.N.Johnson,R.A.White,and ler,“Novel high-speed induction motor for a commercial centrifugal com-pressor,”IEEE Trans.Ind.Applicat.,vol.36,pp.706–713,May/June2000.[3]L.Xu and C.Wang,“Implementation and experimental investigationof sensorless control schemes for PMSM in super-high variable speedoperation,”in Conf.Rec.IEEE-IAS Annu.Meeting,vol.1,1998,pp.483–489.[4]M.Mekhiche,J.L.Kirtley,M.Tolikas,E.Ognibene,J.Kiley,E.Hol-mansky,and F.Nimblett,“High speed motor drive development for in-dustrial applications,”in Conf.Rec.IEMD’99,1999,pp.244–248.[5]T.R.Rowan and R.L.Kerkman,“A new synchronous current regulatorand an analysis of current-regulated PWM inverter,”IEEE Trans.Ind.Applicat.,vol.22,pp.678–690,July/Aug.1986.[6] D.W.Novotony and T.A.Lipo,Vector Control and Dynamics of ACDrives.New York:Oxford Univ.Press,1996.[7]J.-W.Choi and S.-K.Sul,“Inverter output voltage synthesis using noveldead time compensation,”IEEE Trans.Power.Electron.,vol.11,pp.221–227,Mar.1996.[8]J.W.Choi and S.K.Sul,“New dead time compensation eliminating zerocurrent clamping in voltage-fed PEM inverter,”in Conf.Rec.IEEE-IASAnnu.Meeting,1994,pp.977–984.[9]V.Blasko,V.Kaura,and W.Niewiadomski,“Sampling of discontinuousvoltage and current signals in electrical drives:A system approach,”IEEE Trans.Ind.Applicat.,vol.34,pp.1123–1130,Sept./Oct.1998.[10]S.-H.Song,J.-W.Choi,and S.-K.Sul,“Digitally controlled AC drives,”IEEE Ind.Applicat.Mag.,vol.6,pp.51–62,July/Aug.2000.[11]T.Ohmae et al.,“A microprocessor-controlled high accuracywide-range speed regulator for motor drives,”IEEE Trans.Ind.Electron.,vol.29,pp.207–211,Aug.1982.Bon-Ho Bae(S’99–M’03)was born in Korea in1966.He received the Ph.D.degree in electricalengineering from Seoul National University,Seoul,Korea,in2002.He joined Rotem Company(formerly DaewooHeavy Industries Ltd.)in1992and worked for eightyears on the development of propulsion systems forelectric trains.His recent research projects were thedevelopment of the1.2-MV A IGBT inverter for thetraction system of a subway train,sensorless vectordrive system with130-kW70000-r/min PMSM for turbo-compressor,and the42-V ISG system using the high-saliency-ratio IPMSM.He is currently with the General Motors Corporation Advanced Technology Center,Torrance,CA,and his research interests are electric machine drives and automotiveapplications.Seung-Ki Sul(S’78–M’87–SM’98–F’00)was bornin Korea in1958.He received the B.S.,M.S.,andPh.D.degrees in electrical engineering from SeoulNational University,Seoul,Korea,in1980,1983,and1986,respectively.From1986to1988,he was an Associate Re-searcher with the Department of Electrical andComputer Engineering,University of Wisconsin,Madison.He then was with Gold-Star IndustrialSystems Company as a Principal Research Engineerfrom1988to1990.Since1991,he has been a member of the faculty of the School of Electrical Engineering,Seoul National University,where he is currently a Professor.His current research interests are power electronic control of electric machines,electric vehicle drives,and power convertercircuits.Jeong-Hyeck Kwon(S’96–M’99)received theB.S.degree in electronic engineering in1996fromYeungnam University,Taegu,Korea,and the M.S.degree in electronic engineering in1999fromChangwon National University,Changwon,Korea,where he is currently working toward the Ph.D.degree.Since1996,he has been a Researcher in indus-trial control in the Department of Electronics,PowerSystem R&D Center,Samsung Techwin Company,Changwon,Korea.His current research is focused onapplied super-high-speed motors anddrivers.Ji-Seob Byeon received the B.S.and M.S.degreesin electronic engineering from Changwon NationalUniversity,Changwon,Korea,in1998and2001,respectively.Since2001,he has been a Researcher in indus-trial control in the Department of Electronics,PowerSystem R&D Center,Samsung Techwin Company,Changwon,Korea.His current research is focused onDSP applications and digital control.。

变频器的控制方式

V/F曲线
▪ 变频器保持电压和频率成线性关系 460 Vac / 60 Hz = 7.667 V/Hz 230 Vac / 60 Hz = 3.833 V/Hz
为什么比例这么重要？
欧姆定律
▪ 什么是欧姆定律? ——V=IR
▪ 什么是电机的 “R”? ▪ R = 感抗(XL)
▪ 阻值微不足道 ▪ 什么是XL? ▪ XL = 2πFL ▪ V = I (2πFL)
▪ Field Oriented Control ▪ 精确的控制一个电机的励磁和转矩的目的是更加目的是更加精确的实现转矩控制和功率控制。
▪ FOC是罗克韦尔取得专利的控制技术，比FVC有更宽的带宽。
FOC控制图——闭环
• 1336 Force • 1336 Impact • PF700S I and II
FOC控制图——闭环
• 1336 Force • 1336 Impact • PF700S I and II
FOC 机械特性曲线
PowerFlex 家族电机控制方式
V/HZ Sensorless
PF4 & 4M yes
no
PF40 & 40P yes
yes
PF400
yes
no
PF70
yes
yes
PF70 ec yes
yes
PF700 Std yes
yes
PF700 VC yes
yes
PF700H yes
yes
PF700S I no
yes
PF700S II yes
yes
PF700L* yes
yes
PF755
yes
yes
PF753

无感应矢量控制无感应矢量型号变频器的控制原理

无感应矢量控制无感应矢量型号变频器的控制原理无感应矢量控制（Sensorless Vector Control）是一种高级的电机控制技术，通过变频器对电机的控制，实现精确的矢量控制。

在无感应矢量控制下，电机可以实现高效率、高性能的运行。

一、无感应矢量控制的基本原理无感应矢量控制的基本原理是通过对电机的电压和电流进行测量和分析，从而实现对电机的控制。

该控制方法不需要额外的传感器来获得转子位置和速度信息，从而减少了成本和复杂性。

在无感应矢量控制中，变频器根据电机的电压和电流信息，实时计算出电机的转矩和速度。

首先，通过对电机的电流进行矢量分解，得到电流的矢量分量。

然后，根据电压和电流之间的关系，计算出电机的转矩和速度。

最后，通过对电机的电压进行控制，实现对电机的精确控制。

二、无感应矢量控制的优势1. 无需使用传感器：无感应矢量控制不需要额外的传感器来获取电机的转子位置和速度信息，减少了设备的复杂性和成本。

2. 控制精度高：通过对电机的电压和电流进行实时测量和分析，无感应矢量控制可以实现对电机的高精度控制，提高了电机的性能和效率。

3. 适用性广：无感应矢量控制可以应用于不同类型的电机，包括异步电机和永磁同步电机，提高了其适用性和灵活性。

4. 运行平稳：无感应矢量控制可以实现对电机的平稳运行，减少了电机的振动和噪音，提高了设备的可靠性和稳定性。

三、无感应矢量型号变频器的控制原理无感应矢量型号变频器是一种专门用于实现无感应矢量控制的设备。

它通过内部的算法和控制模块来实现对电机的高精度控制。

无感应矢量型号变频器内部包含了电流传感器、电压传感器和控制模块。

首先，电流传感器用于对电机的电流进行测量，获取电流的矢量分量。

然后，电压传感器用于对电机的电压进行测量，实时获取电压的信息。

控制模块是无感应矢量型号变频器的核心部分，它根据电流和电压的信息，实时计算出电机的转矩和速度。

通过对电机的电压进行精确控制，控制模块能够实现对电机的平稳运行和高效率工作。

真正的FOC型-开环矢量控制理论

different in low-speed operation. Since voltage error caused by
dead time is as much as 1.5% 3%, the dead-time voltage error
overwhelms the required phase voltage, for example, in 0.5-Hz
.
Actual (estimated) speed of motor.
-axis (estimated) stator flux in the stator flux
reference frame (SFRF) [1].
-axis voltage in the SFRF.
-axis voltage decomposed to the axis of .
and angular frequency of the flux. Specifically, the polarity and
magnitude of the compensation terms are chosen differently de-
pending upon rotational direction and whether the motor is in the
mance of sensorless field-oriented control. The stator flux is simply
estimated by integrating vs
is. However, this integration
method deteriorates easily in the low-frequency region due to a

无编码器的矢量控制 (SLVC)

被动负载
矢量控制 5.1 无编码器的矢量控制 (SLVC)
异步电机在起动点上拖动的是被动负载时，可以在0频率点（静止）前一直在闭环控制中稳定运行，不会中途切换到开环控制。为此可进行以下设置： 1. 设置 p0500 = 2（工艺应用 = 无编码器闭环控制中的被动负载，直至 f = 0）。 2. 设置 p0578 = 1（计算工艺相关参数）。
电机额定电流实际电机励磁电流/短路电流工艺应用(Application) 静态转矩设定值（无编码器）加速附加转矩（无编码器）电机模型配置电机模型无编码器模式切换转速电机模型无编码器模式切换转速的回差电机模型闭环/开环控制切换延迟时间调制模式最大占空比
驱动功能
206
功能手册, (FH1), 07/2016, 6SL3097-4AB00-0RP5
此时会自动设置以下参数： – p1574 = 2 V（使用异步电机时） – p1574 = 4 V（使用他励同步电机时） – p1750.2 = 1，被动负载条件下在0 Hz前一直在闭环控制中运行 – p1802 = 4（RZM/FLB，不进行过调制） – p1803 = 106 %（出厂设置）经过这些设置后，被动负载功能自动激活。
● 在闭环控制中没有切换操作（工作稳定、无频率骤降、转矩保持恒定） ● 0 Hz前可一直保持无编码器的转速闭环控制 ● 0 Hz 前可以使用被动负载 ● 在进入0 Hz左右的低频区前可一直保持稳定的转速闭环控制 ● 相对于开环控制来说，动态响应更快
说明
请注意，当电机在0 Hz频率闭环控制中启动或反向时，不能此频率附近保持太长时间，即超过2秒或超过p175 8，否则会自动从闭环控制切换到开环控制。
说明再生运行在此运行方式中，频率接近为0 Hz时不可进行静态再生运行。

基于磁链叠加高频信号的PMSM速度位置观测法

［1］
width modulation， SVM）法，可以很好地抑制转矩波动，同时确保开关周期恒定，该方法称为 SVM-DTC 法［5］。另外，由于直接转矩控制系统
与微机控制技术迅猛发展，永磁同步电动机快速占领市场，其控制方法成为电机研究领域的热门。直接转矩控制（direct-torque control， DTC）就是一种高性能调速控制策略，由于其控制方法简单、不需要电机精准模型等优点
Abstract: A novel sensorless control method of permanent magnate synchronous motor （PMSM） under zero-low
electromagnetic torque and stator flux linkage， the load angle and the position error angle could be observed. Finally，
Key words: permanent magnate synchronous motor； direct-torque control； sensorless control； high frequency
永磁同步电动机依靠转子永磁体励磁，具有体积小、功率密度高以及功率因数可控等优点。 20 世纪 80 年代后汝铁硼永磁体出现，电力电子
电气传动 2017 年第 47 卷第 4 期
ELECTRIC DRIVE 2017 Vol.47 No.4
基于磁链叠加高频信号的 PMSM 速度位置观测法
段家珩，张昆仑（西南交通大学磁浮技术与磁浮列车教育部重点实验室，四川成都 610031）
摘要：针对永磁同步电动机能够在零低速条件下实现无传感器运行，提出了一种高频定子磁链注入方法来对电机运行中的速度与位置信号进行提取。在控制系统本身要对转矩以及定子磁链观测的基础上，利用该优势建立负载角观测器并推导位置误差角计算方法，最后使用锁相环原理提取出转子的位置及运行转速。仿真表明，该方法可以实现在永磁同步电机低速下转子速度位置的准确观测，同时在转速与负载突变的情况下也有稳定的观测结果。关键词：永磁同步电动机；直接转矩控制；无位置传感器；高频信号注入法中图分类号： TM351 文献标识码： A DOI： 10.19457/j.1001-2095.20170405

变频器变频率时保持电压的方法

变频器变频率时保持电压的方法There are several methods to maintain voltage while varying the frequency of a variable frequency drive (VFD). One common method is to use a sensorless vector control strategy, which adjusts the VFD output voltage and frequency simultaneously to ensure that the motor receives a constant voltage at varying frequencies. 还有一种常见的方法是使用传感器矢量控制策略，该方法可以同时调节变频器的输出电压和频率，以确保电动机在不同频率下获得恒定的电压。

Another approach is to implement an open-loop control system, where the VFD adjusts the voltage and frequency based on pre-defined motor characteristics and load conditions. 另一种方法是实施开环控制系统，其中变频器根据预定义的电动机特性和负载条件调节电压和频率。

In addition, some VFDs utilize closed-loop control systems, which incorporate feedback mechanisms to continuously monitor and adjust the voltage and frequency to maintain optimal motor performance. 此外，一些变频器还采用闭环控制系统，其中包括反馈机制，可以连续监测和调节电压和频率，以保持最佳电机性能。

低定子频率下消除电流测量误差的磁链观测器

低定子频率下消除电流测量误差的磁链观测器周二磊;符晓;伍小杰;戴鹏【摘要】随着控制性能要求的提高,电流测量误差对电励磁同步电动机控制性能的影响愈加显著。

从电流测量路径看,产生误差不可避免,直流偏移和比例不平衡误差会造成转速周期性地波动。

本文采用一种简单的谐振式观测器对低定子频率下存在的电流测量误差进行补偿,并且为了消除残余误差,纯积分磁链观测器采用了残余误差补偿器以精确观测磁链。

最后,基于Matlab/Simulink对电流两相传感器的电励磁同步电动机调速系统进行仿真,仿真结果证明了该方案的有效性。

%As higher performance of control system is required,current measurement error seriously affected the control performance of electrically excited synchronous motor.The errors generated from the current measurement path are inevitable,such as offset currents and gain deviations,which causes the periodic rotor speed ripples.This paper presents a simple resonant type observer to compensate current measurement errors for the pure-integration-based flux estimation at a low stator frequency.The technique further contains a residual error compensator to eliminate miscellaneous residual error of the integrator.Finally,this paper uses the Matlab/Simulink to simulate excited synchronous motor with two phase current sensors.The simulation results show the effectiveness of the proposed scheme.【期刊名称】《电工技术学报》【年(卷),期】2011(026)006【总页数】6页(P67-72)【关键词】电流测量误差;谐振式观测器;纯积分磁链观测器;电励磁同步电动机【作者】周二磊;符晓;伍小杰;戴鹏【作者单位】中国矿业大学信息与电气工程学院,徐州221008;中国矿业大学信息与电气工程学院,徐州221008;中国矿业大学信息与电气工程学院,徐州221008;中国矿业大学信息与电气工程学院,徐州221008【正文语种】中文【中图分类】TM3411 引言电励磁同步电动机以其效率高、功率因数高且可以调节等优点，在机械传动，特别是在大功率传动中广泛应用[1]。

德尔塔无传感器矢量控制迷你驱动VFD-EL-W系列产品说明书

Delta Sensorless Vector Control Compact Drive VFD-EL-W SeriesAutomation for a Changing WorldSensorless Vector Control Compact Drive VFD-EL-W SeriesSimple Speed ControlHorizontal MovementFixed Load ApplicationsFrame A1Frame BFrame A2Natural cooling (Frame A1): no maintenance requiredFan cooling (Frame A2, Frame B): easy fan installation, reliable design, fast dust removal150 % / 60 secs overload capability150% 60sCE certificationEnergy-savingSingle / multi-pump control:constant pressure mode & alternative operationProtection: overload, over voltage / over current stall preventionBuilt-in PID feedback controlP I DSafety and ReliabilityEasy MaintenanceComplete FunctionsApplicationsEdge Banding Machine• Communication isolation reduces the interference of HMI• One drive for two motors in parallel• Optimized accel. / decel. improve system eﬃciency• Small size, lightweight, easy maintenance and installationLogistics Conveyor• Built-in RS-485 COM port for high-speed communication• Fast and stable tension control• Small and compact design to save installation spaceMaterial Handling Machine• Multiple speed adjustment modes for diﬀerent applications• AVR function to ensure the stability and reliability• Speed tracking function for continuing operation afterpower resumes from an instantaneous power failureConstant Pressure Pump• Build-in PID pressure control; no need for external PID device to save system cost• Built-in automatic inspection and restoration functionsin case of water outage; no external PLC needed• System leakage control function• Multi-pump control: alternates pump operation in cycle(One drive supports max. 4 pumps)Wiring230V / 460V ModelsControl TerminalsMI1MI2MI3MI4SG+SG-PIN3: GND PIN4: SG-PIN5: SG+U(T1)V(T2)W(T3)+10V AVI/ACI ACM 123Fuse/NFB (No Fuse Breaker)FactorySetting Run/StopMulti-step1Multi-step2Digital Signal CommonFactory setting is malfunction indication IM 3～Motor Multi-function Contact Output +24V AVI ACISwitchNPN PNPMain Circuit (power) Terminals Control Circuit Terminals Shielded Leads & CableEERA RB RC R(L1)S(L2)T(L3)(R)L N (S)(T)ETerminal SG+, SG- are joined to PIN5, PIN4 of RJ45 ConnectorDCM Multi-step38 1Switch(Default)(Default)RARBRCMI1MI2MI3DCMSG+SG -ACM +10V AVIRS-485Control Terminal LocationNPN AVI ACIPNP+24V MI4Frames and AppearancesInput Terminals (R / L1, S / L2, T / L3)Digital KeypadControl Board Cover Output Terminals (U / T1, V / T2, W / T3) ■Frame A1 / A 2Digital Keypad Choose NPN / PNP Control Terminals RS-485 Terminal(RJ45)■Frame BInput Terminals (R / L1, S / L2, T / L3)Digital Keypad Control Board CoverOutput Terminals (U / T1, V / T2, W / T3)CaseGrounding TerminalsFrequency Control Potentiometer Frequency Control Potentiometer Choose ACI / A VIA B C D E Digital KeypadChoose NPN / PNP Control Terminals RS-485 Terminal(RJ45)Choose AVI / A CIA B C D E CABDECABDEGrounding Terminals Cooling Fan* Frame A1 does not include a cooling fanModel ExplanationVFD 007 EL 21 W -1Minimal & Economic Input Voltage21: 230 V 1 PH 43: 460 V 3 PHVFD-EL-W SeriesApplicable Motor Capacity002: 0.25 HP (0.2 kW)004: 0.5 HP (0.4 kW)007: 1 HP (0.75 kW) 015: 2 HP (1.5 kW)022: 3 HP (2.2 kW)040: 5 .5 HP (4.0 kW)Variable Frequency DriveMethods of Packing-1: Individual Package N/A: 12 pcs / CartonVoltage 230 V460 V FrameA1BA1A2BModelVFD-__EL21W(-1) VFD-__EL43W(-1)002004007015022004007015022040Max. Applicable Motor Output (kW)0.20.40.75 1.5 2.20.40.75 1.5 2.2 4.0Max. Applicable Motor Output (HP)0.250.5 1.0 2.0 3.00.5 1.0 2.0 3.0 5.5Output RatingRated Output Capacity (kVA)0.6 1.0 1.6 2.9 4.2 1.2 2.0 3.3 4.47.4Rated Output Current (A) 1.62.54.27.511.01.52.54.25.59.0Maximum Output Voltage (V)3-Phase Proportional to Input VoltageOutput Frequency (Hz)0.1 ~ 400Carrier Frequency (kHz) 2 ~ 12 (Default 8 kHz)Input Rating Rated Input Current (A)4.96.59.315.724.01.83.24.37.110.0Rated Voltage / Frequency Single Phase, AC 200 V ~ 240 V ，50 / 60 HzThree Phase, AC 380 V ~ 480 V ，50 / 60 HzVoltage Tolerance±10 % (180 V ~ 264 V)±10 % (342 V ~ 528 V)Frequency Tolerance±5 % (47 Hz ~ 63 Hz)Weight (kg) 1.0 1.4 1.01.4Cooling Method Natural CoolingFan CoolingNatural CoolingFan CoolingBrake Unit N/A DC Choke N/A EMI FilterN/AProduct SpecificationsSpecificationsC o n t r o l C h a r a c t e r i s t i c sControl SystemSPWM (Sinusoidal Pulse Width Modulation) control (V / F Control, Vector Control)Frequency Setting Resolution 0.01 Hz Output Frequency Resolution0.01 HzTorque Characteristics Including the auto-torque, auto-slip compensation; starting torque can be 150 % at 5 Hz Overload Endurance 150 % of rated current for 1 minuteSkip Frequency Three zones, setting range 0.1 ~ 400 HzAccel. / Decel. Time 0.1 to 600 secs (2 independent setting for accel. / decel. time)Stall Prevention LevelSetting 20 to 250 % of rated currentDC Brake Operating frequency 0.1 ~ 400 Hz, Output 0 ~ 100 % rated current Start time 0 ~ 60 secs, stop time 0~60 secs V / F PatternAdjustable V / F pattern O p e r a t i n g C h a r a c t e r i s t i c sFrequency SettingKeypadSetting by▲▼External Signal Potentiometer : 5k Ω / 0.5 W, 0 to +10 VDC, 4 to 20 mAMulti-function input MI2 ~ MI4 (8 steps : Including the main speed, jog, up / down); RS-485 serial interface Operating Setting SignalKeypad Setting by RUN and STOPExternal SignalRUN / STOP by MI1 (default) or 2-wire / 3-wire control (MI1, MI2, MI3), jog operation, RS-485 serial interface (Modbus)Multi-function Input Signal8-speed switch (including the main speed): ban commands for acceleration / deceleration, 2-speed switch for accel. / decel., counter, jogging (inching), external base block, driver reset, NPN / PNP inputs, AVI / A CI analog inputs Switch to a speed as the default.Multi-function Output Signal(only Relays) AC drive operating, frequency attained, zero speed, counter, over-torque inspection, external base block, operating modes, anomaly alarm, overheating alarm, emergency stopProtection FunctionsOver voltage, over current, under voltage, anomalies, overload, overheating, electronic thermal relays, PTC overheating protectionOperation FunctionsBuilt-in voltage regulators, accel. / decel. S-curve, over-voltage / over-current stall prevention, 5 anomalous logs, reverse ban, restart for instantaneous power outage, DC brake, automatic toque / slip compensation and motor parameter adjustment, carrier frequency setting, output frequency limits, parameter reset, PID control, external counter, Modbus protocol, reset and restart for anomalies, energy-saving, fan control (for models with fans), 1st / 2nd frequency sources and combination, NPN / PNP inputsDisplay Keypad (optional) 6 function keys, 4-digit 7-segment LED, 4 status LEDs, adjustable frequency, self-defined units, parameter settings and lock function, anomaly alarms, Run / Stop / Reset buttons E n v i r o n m e n t a l C o n d i t i o n sEnclosure Rating IP20Pollution Degree 2Installation Location Altitude 1,000 m or lower, keeping from corrosive gases, liquids and dust Operating Temperature -10°C to 50°C (VFD007EL21W(-1) requires fan accessories)Storage / TransportationTemperature -20°C to 60°CAmbient HumidityBelow 90 % RH (non-condensing)Vibration 1.0 mm, peak to peak 2–13.2 Hz; 0.7–1.0 G, 13.2–55 Hz; 1.0 G, 55–512 Hz; compliant with IEC 60068-2-6 Certification, RoHS, GB 12668.3SpecificationsGeneral Specifications■Frame A1■Frame A2ModelVFD002EL21W(-1)VFD004EL21W(-1)VFD004EL43W(-1)VFD007EL21W(-1)VFD007EL43W(-1)Frame W W1H H1D D1S A1mm 92.082.0162.0152128.7 2.00 5.4inch3.62 3.23 6.38 5.98 5.070.080.21DimensionsModelVFD015EL43W(-1)Frame W W1H H1H2D D1S A2mm 92.082.0180.5162.0152128.7 2.00 5.4inch3.62 3.237.11 6.38 5.98 5.070.080.21SA(Mounting Hole )W1W DSee AH 1HD1W1W H 1HH 2SADSee AD1(Mounting Hole )Frame W W1H H1D D1S1S2Bmm 100.089.0174.0162.9136.0 4.0 5.9 5.4inch3.94 3.50 6.85 6.42 5.350.160.230.21ModelVFD015EL21W(-1)VFD022EL21W(-1)VFD022EL43W(-1)VFD040EL43W(-1)Ordering InformationFrameCooling MethodOperating TemperaturePower RangeModels Frame A1Natural Cooling-10°C ~ 50°C 230 V: 0.2 ~ 0.75 kW 460 V: 0.4 ~ 0.75 kWVFD002EL21W(-1)VFD004EL21W(-1)VFD004EL43W(-1)VFD007EL21W(-1)* VFD007EL43W(-1)Frame A2Fan Cooling460 V: 1.5 kW VFD015EL43W(-1)Frame B230 V: 1.5 ~ 2.2 kW 460 V: 2.2 ~ 4.0 kWVFD015EL21W(-1)VFD022EL21W(-1)VFD022EL43W(-1)VFD040EL43W(-1)■Frame BVFDxxxELxxW-1 and VFDxxxELxxW share the same electrical specifications.*VFD007EL21W(-1): to reach 50℃operating temperature, a fan kit MKEL-AFKM1 is required (without derating). *VFD007EL21W(-1): to reach 40℃operating temperature, no need for a fan kit (without derating).W1H 1W HDS1D1AS2BSee ASee B(Mounting Hole )(Mounting Hole )10AccessoriesKeypadCableFan kitReactorVFD-PU06VFD-PU08RJ45 CableVFD-PU08VRF220X00AMKEL-AFKM1• 5 digits• Parameter duplication and recording • RJ11 connector • RS-485 communication• 4 digits• RJ45 connector• RS-485 communication• 4 digiits• RJ45 connector• RS-485 communicationNo.ModelLengthmm inch 1UC-CMC003-01A 30011.82UC-CMC005-01A 50019.63UC-CMC010-01A 100039.04UC-CMC015-01A 150059.05UC-CMC020-01A 200078.76UC-CMC030-01A 3000118.17UC-CMC050-01A5000196.8* RJ45 cable is not included for VFD-PU08 & VFD-PU08V.DELTA_IA-MDS_VFD-EL-W_C_EN_20200716Industrial Automation HeadquartersDelta Electronics, Inc.Taoyuan Technology CenterNo.18, Xinglong Rd., Taoyuan District, Taoyuan City 33068, TaiwanTEL: 886-3-362-6301 / FAX: 886-3-371-6301AsiaDelta Electronics (Shanghai) Co., Ltd.No.182 Minyu Rd., Pudong Shanghai, P .R.C.Post code : 201209TEL: 86-21-6872-3988 / FAX: 86-21-6872-3996Customer Service: 400-820-9595Delta Electronics (Japan), Inc.Tokyo OfficeIndustrial Automation Sales Department 2-1-14 Shibadaimon, Minato-ku Tokyo, Japan 105-0012TEL: 81-3-5733-1155 / FAX: 81-3-5733-1255Delta Electronics (Korea), Inc.Seoul Office1511, 219, Gasan Digital 1-Ro., Geumcheon-gu, Seoul, 08501 South KoreaTEL: 82-2-515-5305 / FAX: 82-2-515-5302Delta Energy Systems (Singapore) Pte Ltd.4 Kaki Bukit Avenue 1, #05-04, Singapore 417939TEL: 65-6747-5155 / FAX: 65-6744-9228Delta Electronics (India) Pvt. Ltd.Plot No.43, Sector 35, HSIIDC Gurgaon, PIN 122001, Haryana, IndiaTEL: 91-124-4874900 / FAX : 91-124-4874945Delta Electronics (Thailand) PCL.909 Soi 9, Moo 4, Bangpoo Industrial Estate (E.P .Z), Pattana 1 Rd., T.Phraksa, A.Muang, Samutprakarn 10280, ThailandTEL: 66-2709-2800 / FAX : 662-709-2827Delta Electronics (Australia) Pty Ltd.Unit 20-21/45 Normanby Rd., Notting Hill Vic 3168, Australia TEL: 61-3-9543-3720AmericasDelta Electronics (Americas) Ltd.Raleigh OfficeP .O. Box 12173, 5101 Davis Drive,Research Triangle Park, NC 27709, U.S.A.TEL: 1-919-767-3813 / FAX: 1-919-767-3969Delta Greentech (Brasil) S/ASão Paulo OfficeRua Itapeva, 26 – 3˚ Andar - Bela Vista CEP: 01332-000 – São Paulo – SP - Brasil TEL: 55-11-3530-8643 / 55-11-3530-8640Delta Electronics International Mexico S.A. de C.V.Mexico OfficeGustavo Baz No. 309 Edificio E PB 103Colonia La Loma, CP 54060Tlalnepantla, Estado de México TEL: 52-55-3603-9200*We reserve the right to change the information in this catalogue without prior notice.EMEAHeadquarters: Delta Electronics (Netherlands) B.V.Sales:*************************Marketing:*****************************TechnicalSupport:******************************CustomerSupport:****************************Service:***************************TEL: +31(0)40 800 3900BENELUX: Delta Electronics (Netherlands) B.V.De Witbogt 20, 5652 AG Eindhoven, The Netherlands Mail:****************************TEL: +31(0)40 800 3900DACH: Delta Electronics (Netherlands) B.V.Coesterweg 45, D-59494 Soest, Germany Mail:*************************TEL: +49(0)2921 987 0France: Delta Electronics (France) S.A.ZI du bois Challand 2, 15 rue des Pyrénées, Lisses, 91090 Evry Cedex, France Mail:***********************TEL: +33(0)1 69 77 82 60Iberia: Delta Electronics Solutions (Spain) S.L.UCtra. 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检测感应电机磁链的闭环方法

检测感应电机磁链的闭环方法作者：马鑫陈增禄马和平来源：《电子世界》2013年第10期【摘要】要实现电磁转矩和磁链的解耦控制，必须得到足够精确的磁链检测值。

计算磁链不可避免地受到电机参数不准确以及测量干扰的影响。

磁链的开环检测方法缺少对各种干扰的抑制，通过引入反馈来抑制干扰和变参数的影响是简单而有效的，称为检测磁链的闭环方法。

介绍四种检测磁链的闭环方法：（1）使用电流反馈的改进的电压模型法；（2）现代控制理论的观测器法；（3）滑模观测器法；（4）人工神经网络法。

【关键词】感应电机；磁链检测；观测器；滑模观测器；神经网络一、引言近年来，在交流调速领域矢量控制和直接转矩控制获得了巨大的发展并得到广泛应用，这两种控制策略都能使感应电机中耦合的电磁转矩和磁链达到与直流电机类似的解耦状态。

这样可以将感应电机结构上的可靠性与直流电机控制上的简单有效结合起来，使传统上可靠耐用但控制性能较差的感应电机在很多高性能应用场合代替了传统上控制性能最好的直流电机。

要实现电磁转矩和磁链的解耦控制，必须得到足够精确的磁链检测值。

最初矢量控制曾采用在电机槽内埋设线圈或在定子内表面设置磁敏元件的方式来直接检测气隙磁链信号，间接推算转子磁链或定子磁链，这种方式有不少工艺和技术问题，转速越低齿槽造成的检测信号脉动影响越严重，磁敏元件输出信号受温度影响大[1]。

实际中大多采用间接计算的方法，即根据感应电机数学模型导出磁链与较容易获得的电压、电流及转速间的数学关系，以此实时计算磁链。

由于磁链在控制中处于被检测反馈的地位，一般也将间接计算磁链称为检测磁链，在本文中，检测磁链就是指间接计算磁链。

由于使用电机的数学模型来计算磁链不可避免地用到电动机的定子参数和转子参数，所以使用的电机参数是否准确决定磁链估计是否准确。

而感应电机的定子和转子参数是时变的，比如与启动时相比，感应电机运行时定子电阻的变化量能超过50%，而转子电阻的变化量能超过100%[2]，因此怎样准确检测磁链，或者减小甚至消除不准确的参数对计算磁链造成的影响是十分重要的问题。

一种PMLSM的速度辨识控制方法

一种PMLSM的速度辨识控制方法李奇军;牛永江;时立民【摘要】针对永磁同步直线电机的矢量控制问题,提出了基于模型参考自适应的速度辨识方法.通过分析永磁同步直线电机的数学模型,设计了并联模型参考自适应模型,并利用Popov超稳定性定理推导了模型参考自适应律,得到了速度辨识自适应算法,实现了对电机速度和位置的辨识.将速度辨识算法嵌入永磁同步直线电机的双闭环矢量控制模型中,建立了基于模型参考自适应速度辨识的双闭环矢量控制系统.在MATLAB中对电机及其控制系统进行建模,并在负载突变和速度突变两种工况条件下对系统模型进行了仿真,仿真结果表明,该方法能实现对速度和位置的高精度估算,稳定性较好.【期刊名称】《机械设计与制造》【年(卷),期】2018(000)009【总页数】4页(P221-223,227)【关键词】直线电机;矢量控制;模型参考自适应;速度辨识【作者】李奇军;牛永江;时立民【作者单位】天水师范学院机电与汽车工程学院,甘肃天水 741000;天水师范学院机电与汽车工程学院,甘肃天水 741000;天水师范学院机电与汽车工程学院,甘肃天水 741000【正文语种】中文【中图分类】TH16;TM359.41 引言永磁同步直线电机（permanent magnet linear synchronous motor，PMLSM）体积小、效率高、推力大，而且在速度和定位精度方面优于其他直线电机，但PMLSM的矢量控制需要对速度进行专门测量反馈，在一定程度上对设备结构性能提出了更高的要求[1-3]。

因此，采用简单有效的速度辨识算法对直线电机的速度进行估计，具有实际工程意义。

模型参考自适应法（model reference adaptive system，MRAS）相对拓展卡尔曼滤波等算法具有结构简单、运算量小等优点被广泛应用于伺服系统，对系统参数进行辨识[4-7]。

采用模型参考自适应辨识算法，对PMLSM的速度进行辨识，以此构建PMLSM的双闭环矢量控系统并对其进行仿真验证。