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新型EBG光子晶体结构UWB天线

新型EBG光子晶体结构UWB天线0 引言超宽带(Ultra Wide Band, UWB)无线通信技术有着高带宽、低功耗、低复杂度等优点,因此其成为目前短距离无线通信的热点之一。

UWB 通信系统的天线必须具有体积小、宽带宽、全向覆盖等特点。

而一般将比带宽不小于10∶1 的天线,称为UWB 天线。

UWB 天线随着高速电子集成电路的快速发展,为适应小型集成化的需求,超宽带平面天线的研究与应用也渐渐称为研究热点。

2002 年美国联邦通信委员会(FCC)批准将3.1～10.6GHz 频段划作UWB 技术的商业应用频段,将广泛应用于新兴短程通信中。

UWB 技术规定的室内UWB 通信的实际频谱比带宽最高频率fu 与最低频率fl 之比,即fu/fl 为3.42∶1;同时规定对中心频率大于2.5GHz的UWB 系统需要拥有至少-10dB 的500MHz 带宽,而对中心频率低于2.5GHz 的UWB 系统,带宽至少应达到20%. 目前已研究了不少新型的UWB 天线,新型平面印刷单极子天线[1~3]不需要另加与之垂直的地板,其辐射UWB 与地板在一个介质面上,中间通过CPW 进行馈电。

采用宽缝隙结构UWB 天线[4]则是由最早出现的渐变式缝隙结构发展为现在的边射式宽带印刷缝隙结构,工作带宽可达到2～10 GHz.在工作频带内具有良好的近似全向辐射特性,因此,是具有实用价值的UWB 天线。

电磁带隙结构简称EBG（Electromagnetic Band-gap）光子晶体结构，它是由光子晶体应用于微波波段产生的一种称谓。

它们都是周期性介质结构，光子晶体在这种介质结构传播时，周期性势场造成的能量分级而出现禁带，对光子来说他的能量与频率的平方成正比，禁带意味着禁止某频段光子在其中传播。

这种特性称为光子带隙。

同样在电磁带隙结构中，由于电子和光子都满足薛定谔和麦克斯韦方程组，它们的同源性使得原子核周期性的势场同样会造成电子能量的分散，也表现为通带或者禁带。

金属挡板对共面天线隔离效应的试验研究

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目前可以采用的新的隔离方法有涂敷吸波材料［2~4］，EBG 电磁材料阻隔［5~7］，属挡板阻隔等［8~10］方法。本文主要研究金属挡板对阵面天线的隔离情况，金属挡板在对电磁波产生遮挡，改变传播方向和特性的同时，也会让电磁波产生绕射效应。国内
外针对金属挡板的电磁效应已经做了一定的研究，黄龙水［8］研究了金属平板对阵列天线端射特性的影响，研究表明由于金属挡板的反射效应，使其在 H 面的端射小于 E 面的，并且这种特性与阵列天线的扫描特性的相关性较小。宋东安等对自由空间下双叶挡板［9］和菲涅尔圆盘的［11］遮挡效应进行了理论和试验研究。随后又在此基础上通过理论和仿真研究［10］了阵面天线间垂直安装挡板时，在平面波照射的挡板后方区域阴影区域场强的影响因素。国防大学吕波［12］采用 U 形金属挡板，有效地抑制了阵面天线间的耦合，给应用提供了指导。
3 试验及结果分析
为了验证理论分析，采用试验的方法来研究高度对共面天线间隔离度影响，本文中隔离度通过传输系数 S21 来衡量。在金属挡板隔离试验中，采用两副喇叭天线模拟收发共面天线，发射方向与天线的安装面垂直（天线安装面为金属结构），喇叭口与天线安装面处于同一平面上。金属挡板垂直于天线安装面，金属挡板为与接收天线上方 90cm，天线间间距为 4.5m。试验测量收发天线间传输系数 S21，用于衡量天线间的隔离情况。试验期间整个

质粒图谱查询方法

3.google scholar: / 有些质粒是经过改造的，所以通过上述方法不能查询到相应信息。这时，可以在google scholar中输入质粒名称，可以直观地看哪些学者在何文章中使用了该质粒，从而可了解到质粒的来源;或者籍此向作者咨询或索取。 4.尝试从各大生物公司，例如invitrogen网站查询. 5. 这个网站收录了大量图谱： http://www.embl-hamburg.de/~geerlof/webPP/vectordb/bact_vectors/table.html
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0099--pGE-1—Stratagene--RNAi载体 0100--pSUPER.p53—OligoEngine--RNAi载体 0101--palter-ex1--promega 0102--pACYCDuet-1--NOVAGEN 0103--pEX lox(+) Vector—NOVAGEN--原核表达 0104--质粒名称：pBACgus-8 Transfer Plasmid—NOVAGEN--CHUANSUO 0105--pSCREEN?-1b(+) Vector Map—novagen--筛选 0106--PGEX-2T--BD Co--pDsRed2--Clontech 0107--pbgal-Basic—Clontech--mammalian reporter vector 0108—pBI—Clontech--express two genes of interest from a bidirectional tet-responsive promoter 0109--质粒名称：pbgal-Control—Clontech--mammalian reporter vector 0110-- pGEX-5X-1--原核表达 0111--pBI-EGFP—Clontech--pBI-EGFP-- coexpress 0112--pBI-G—Clontech--pBI-G--express b-galactosidase 0113--pBI-GL—Clontech--pBI-GL --express luciferase and b-galactosidase 0114--pCMS-EGFP—Clontech--mammalian expression vector 0115--pd2EYFP-1—Clontech--启动子测定 0116--质粒名称--pd2EYFP-N1—Clontech--融合表达 0117--pd4EGFP-Bid—Clontech--融合表达 Bid 0118--pDNR-CMV—Clontech--pDNR-CMV 0119--pDNR-EGFP Vector—Clontech 0120--pDNR-LacZ –Clontech 0121--pECFP-Endo—Clontech--真核表达0122--pECFP-ER—Clontech--真核表达0123--pEGFP-Actin—Clontech--真核表达0124--pGAD GH--Clontech--酵母表达 0125--pGADT7-Rec –Clontech--酵母表达 0126--pGADT7-RecAB—Clontech--酵母表达 0127--pGADT7-Rec2—Clontech--酵母表达 0128--pGBKT7—Clontech--酵母表达 0129--pHAT 10/11/12—Clontech 0130--pHAT20—Clontech 0131—pHygEGFP—Clontech 0132—pLacZi—Clontech 0133—pM—Clontech--pM is used to generate a fusion of the GAL4 DNA-BD 0134--pPKCa-EGFP—Clontech 0135--pPKCb-EGFP—Clontec 0136--pSIREN-DNR Vector—Clontech--RNAi 0137--pSIREN-DNR-DsRed-Express Vector—Clontech--RNAi 0138--pSIREN-RetroQ—Clontech--RNAi 0139--pIRES-EYFP—Clontech--RNAi 0140--pSRE-Luc—Clontech--RNAi 0141--pTK-neo—novagen--原核表达 0142--pZsGreen Vector—Clontech--pZsGreen is a pUC19-derived prokaryotic expression vector 0143--pTandem-1—novagen--原核表达 0144--pZsGreen1-C1Vector—Clontech----真核表达 0145--质粒名称：M13mp18—novagen--原核表达 0146--pZsGreen1-DR Vector—Clontech--真核表达 0147--PZsGreen1-N1 Vector—Clontech --真核表达 0148--T7Select415-1b—novagen----真核表达 0149--pZsYellow Vector—Clontech --真核表达 0150—pTimer—Clontech --真核表达 0151--pTA-Luc—Clontech --真核表达 0152--pTAL-Luc—Clontech --真核表达 0153--pTA-SEAP—Clontech --真核表达 0154--pTAL-SEAP—Clontech --真核表达 0155--pTet-On—Clontech --真核表达 0156--pTet-Off—Clontech --真核表达 0157--pTet-ATF—Clontech --真核表达 0158--pTet-CREB—Clontech --真核表达

小型化超宽带叶型微带单极子天线设计

现代电子技术Modern Electronics TechniqueJul.2023Vol.46No.142023年7月15日第46卷第14期0引言随着时代的发展，向大容量、高速率方向发展的无线通信技术成为了该领域的主要目标[1‐2]。

作为通信系统中的关键模块，超宽带[3‐4]（UWB ）天线可以极大提高无线通信系统的信道容量、频谱效率和工作带宽范围，有着广阔的应用前景。

具有三维结构的倒锥天线，结构对称性高，能够实现43∶1的阻抗带宽[5]，但是其体积大，馈电结构稳定性差。

因此，具有低成本、易小型化及易加工等优势的微带单极子天线，逐渐成为无线通信领域的焦点[6]。

基于印刷电路板（PCB ）的微带单极子天线，在贴片上采用分形结构，比如六边型[7]、雪花型[8]或者勋章型[9]等，增加贴片的周长来提升带宽。

相比于线形结构，圆形结构周长更大，且对称性高，带宽更宽。

文献[10]中，利用椭圆型辐射贴片实现了24.1∶1的宽带阻抗匹配。

DOI ：10.16652/j.issn.1004‐373x.2023.14.001引用格式：李想，曹建银，姚晨阳，等.小型化超宽带叶型微带单极子天线设计[J].现代电子技术，2023，46（14）：1‐6.小型化超宽带叶型微带单极子天线设计李想1，3，曹建银2，姚晨阳2，3，丁振东2，王昊2，3，陶诗飞2（1.电磁空间认知与智能控制技术实验室，北京100191；2.南京理工大学，江苏南京210094；3.南湖实验室，浙江嘉兴314002）摘要：针对目前超宽带（UWB ）微带单极子天线带宽较窄以及尺寸较大等缺点，文中提出一种基于共面波导（CPW ）馈电的小型化超宽带微带单极子天线。

该天线由叶型的辐射贴片（其上挖去3个圆形贴片）、梯形地板和环形三叉戟共面馈电组成，可实现1~18GHz 的超宽带频率覆盖。

使用HFSS 软件对天线的结构和尺寸进行分析，得出最终的天线尺寸仅为40mm×75mm×0.5mm 。

A Compact, High Isolation and Wide Bandwidth Antenna Array for Long Term Evolution Wireless Devices

A Compact,High Isolation and Wide Bandwidth AntennaArray for Long Term Evolution Wireless Devices Mina Ayatollahi,Qinjiang Rao,and Dong WangAbstract—A compact dual-port,multiple input-multiple output(MIMO) antenna array for handheld devices is introduced.The antenna structure consists of two quarter wavelength monopole slots etched on the ground plane of a printed circuit board(PCB)and a meandered slot cut between them.The meandered slot not only reduces the coupling between the two slot antennas,but also improves the bandwidth and efﬁciency of the array by acting as a radiating parasitic element.Simulated and measured results show that the meandered isolating slot allows the antennas to achieve wider bandwidth,higher efﬁciency,higher isolation and better diversity perfor-mance,compared to other types of isolating slots.Index Terms—Antenna mutual coupling,MIMO antennas,mobile de-vice,slot antennas.I.I NTRODUCTIONWith the emergence of new wireless standards such as long-term evolution(LTE),multiple-input-multiple output(MIMO)technology which uses multiple antennas,has become a very promising technique for enhancing the performance of wireless communication systems [1]–[3].Optimal MIMO performance requires low correlation between signals received by each of the antennas.This requires low mutual coupling between the antenna ports which is not normally possible in a compact device,because the antennas are closely spaced.The high mutual coupling,which is due to the surface waves induced in the ground plane,increases the received signal correlation and decreases diversity gain and channel capacity[4].To reduce the mutual coupling between the antenna elements,var-ious approaches have been used,including neutralization technique[5], simultaneous matching[6],etching slits in the middle of the ground plane[7]and using EBG substrates[8].These techniques either occupy a considerable space on the PCB or need special fabrication techniques. Another approach is etching an isolating slot between the antenna el-ements.For example a vertical slot[9],a T-shaped slot[10],or two L-shaped slots[11]have been used.Although these isolating slots re-duce the mutual coupling between the antennas,they do not improve their bandwidth.In this communication,a compact multiband MIMO antenna array for handheld devices is presented.The structure consists of two monopole radiating slots and a new meandered isolating slot, all etched along an edge of the ground plane of the PCB[12].As a design example,the proposed antenna system has been designed to operate in the2.6GHz LTE band(2.5–2.7GHz),as well as the2.5 GHz(2.4–2.5GHz)WLAN band.Simulated and experimental results, including S-parameters,radiation patterns,radiation efﬁciency and signal correlations,are presented and discussed.The results show that the meandered isolating slot reduces the mutual coupling between the two slot antennas and also acts as a radiating parasitic element, Manuscript received October25,2010;revised May30,2011;accepted May 04,2012.Date of publication July05,2012;date of current version October02, 2012.The authors are with the Research in Motion Limited(RIM),Waterloo,ON N2V2P1,Canada(e-mail:mayatollahi@).Color versions of one or more of theﬁgures in this communication are avail-able online at .Digital Object Identiﬁer10.1109/TAP.2012.2207312Fig.1.MIMO slot antenna array.which introduces an additional resonance frequency and increases thebandwidth of the antennas.The performance of the proposed antennasystem has also been compared to the one with other isolating slotshapes such as T-shaped,vertical,a pair of L-shaped and also withoutan isolating slot.The comparisons show that the bandwidth of theproposed antenna system is4times the one when no isolation slot isused,and more than three times the bandwidth using other slot types.This communication is organized as follows.Section II presents thedesign and layout of the proposed antenna array.In Section III,the sim-ulated and measured results of the array are presented and compared tothe system without the isolating slot.The diversity parameters and per-formance of the proposed MIMO system are discussed in Section IV.Section V compares the performance of the proposed array to a similararray with other isolating slot shapes.Finally,Section VI provides theconcluding remarks.II.A NTENNA A RRAY S TRUCTUREAs shown in Fig.1,the proposed antenna structure consists of twoquarter wavelength radiating slots cut close to an edge of a groundplane,on one side of a FR4substrate with a thickness of1.5mm anda relative permittivity of4.4.The length and width of the substrate andthe ground plane are95mm and55mm,respectively.The antennasystem is designed to operate at2.6GHz LTE application band.Basedon the required impedance bandwidth and resonance frequency,the di-mensions of the radiating slots and their distance to the edge of theground plane,which are denoted by l,w,and d in Fig.1,has been op-timized using a Finite Difference Time Domain commercial software.These parameters have been obtained as20mm,1mm and3mm,re-spectively.Each antenna is fed with a50 impedance feed at a distanceof3mm from its closed end.A meandered isolating slot is cut between the two antennas,as shownin Fig.1.The width of the meandered slot is1mm and its total length isoptimized at about quarter of the wavelength at the center frequency of2.6GHz,which is around30mm.The lengths of the three arms of themeandered slot parallel to the top edge of the PCB are6mm,11mmand5mm,respectively.Other dimensions are shown in Fig.1.Basedon the optimized dimensions,the antenna array is prototyped and theantennas are fed by coaxial cables,as shown in Fig.2.0018-926X/$31.00©2012IEEEFig.2.Prototype of the antenna array with the meandered isolatingslot.Fig.3.Simulated and measured S parameters (dB)for antenna P1of Fig.1with and without (w/o)the meandered isolating slot.III.S IMULATED AND M EASURED R ESULTSTo investigate the effect of the meandered slot on the performance of the antenna array,antenna P1in Fig.1is excited at the frequency of 2.6GHz while the other antenna is terminated to a 50 load.The simulated S parameters of the array with and without (w/o)the meandered slot,and the measured S parameters of the prototype of Fig.2,are shown in Fig.3.The results obtained for port 2are similar and not presented here.A very good agreement between the simulated and measured S parameters is observed.As shown,the meandered slot has increased the isolation between the two ports from 8dB to 15dB at 2.6GHz,and the isolation is more than 15dB across the entire bandwidth.It is also observed that the meandered slot has increased the bandwidth of the antennas at 10dB return loss more than 4times,from 100MHz to more than 400MHz The bandwidth which is from 2.4–2.84,covers two application bands,LTE 2.6GHz and W ALN 2.5GHz.The S11plots show that in the structure with the meandered slot,there are two resonance frequencies close to each other,resulting in a wider band-width.Since the meandered slot has branches close to the excited an-tenna,there is a strong coupling between the two slots.The meandered slot is then parasitically fed through the excited antenna and acts as a parasitic radiator,contributing to the total radiation and improving the bandwidth.To demonstrate the effect of the meandered slot on the performance of the antennas,the current distribution on the ground plane with and without the meandered slot are obtained at the frequency of 2.6GHz and shown in Fig.4.As shown in Fig.4(a),a high concentration of current is observed on the ground plane close to the second antenna,and on the top edge of the ground plane,which demonstrates the high mutual coupling between the two slot antennas when the isolatingslotFig.4.Current distribution for the antenna array of Fig.1,when P1is excited.(a)Without the isolating slot.(b)With the meandered isolatingslot.Fig.5.Measured impedance curves for Port 1in the 2.4–2.7GHz frequency range.(a)Without the meandered slot.(b)With the meandered slot.is not used.As seen in Fig.4(b),adding the meandered slot reduces the current around the second antenna considerably.This is because the surface waves are suppressed from reaching the second antenna,which improves the isolation between the two antennas.Also,a strong current distribution around the meandered slot is ob-served,especially around the portion which is adjacent the excited an-tenna.This shows a strong coupling between the meandered slot and the excited antenna.The meandered slot acts as a parasitic radiating element coupled to the excited antenna and contributes to the total ra-diation.In addition,this coupling creates an additional resonance fre-quency for the excited antenna and improves the impedance bandwidth of the excited antenna considerably.This is shown in the measured input impedance of antenna P1,with and without the meandered slot,in Fig.5for the 2.4-2.7GHz frequency range.The simulated radiation patterns at the frequency of 2.6GHz are shown in Fig.6for antenna P1with and without (W/O)the isolating slot.As seen,the radiation pattern with the isolating slot is more omni-directional in the horizontal XY plane in the direction of the secondFig.6.Simulated gain patterns at 2.6GHz for radiating slot P1of Fig.1.(a)X -Y plane.(b)Y -Z plane.(c)X -Zplane.Fig.7.Measured radiation efﬁciency for antenna P1.antenna,compared to when the slot is not used.The simulated radiation efﬁciency of the antenna is also obtained at the frequency of 2.6GHz for both cases.The radiation efﬁciency without the meandered slot is obtained as 81.7%.The isolating slot increases the efﬁciency to 86.4%,which is due to the reduced mutual coupling between the slot antennas and also radiation from the meandered slot which acts as a parasitic radiating element.The measured radiation efﬁciency of antenna P1is shown in Fig.7.The measured radiation efﬁciency is around 78%at the frequency of 2.6GHz and more than 70%over the entire bandwidth.The measured radiation patterns of antenna P1is shown in Fig.8for the frequency of 2.6GHz.It should be noted that the simulated gain patterns and efﬁ-ciency are obtained by considering the conductor and dielectric losses of the structure,but with the assumption of an ideal feeding arrange-ment.The measured results include the insertion loss of the actual feed network and connector and cable loss.Therefore there are some dis-crepancies between the measured and simulated radiation patterns and efﬁciency as a result of the physical feeding arrangement.IV .D IVERSITY P ERFORMANCE OF THE A NTENNA A RRAYThe envelope correlation coefﬁcient (ECC)is used to evaluate the diversity performance of multi antenna systems.The envelope correlation coefﬁcient can be calculated using the far-ﬁeld pattern data [13].Diversity gain is obtained when the envelope correlation coefﬁcient is less than 0.5,and in uniform environment,when the radiation efﬁciencies of the two antennas are close toeachFig.8.Measured radiation patterns at the frequency of 2.6GHz for antenna P1.(a)X -Y plane.(b)Y -Z plane.(c)X -Z plane.TABLE ID IVERSITY P ARAMETERS OF THE MIMO A RRAY OF F IG .1.TABLE IIP ERFORMANCE C OMPARISON OF THE S TRUCTURE OF F IG .1W ITH V ARIOUSI SOLATING S LOT SHAPESother.The envelope correlation coefﬁcient of the antenna array of Fig.1has been computed and shown in Table I for various frequencies in the operating bandwidth.The uniform angular power spectrum and isotropic environment is considered for this calculation.It is observed that the envelope correlation is close to zero over the bandwidth,which means that the patterns of the two antennas are de-correlated and demonstrates excellent diversity condition.V .C OMPARISON W ITH O THER I SOLATING S LOT S HAPESThe radiation and diversity performance of the antenna array of Fig.1is simulated and compared to the performance of the array when other isolating slot shapes are used in place of the meandered slot.The slot shapes that are considered are T-shaped,dual L and a quarter wavelength vertical slot.The isolating slot in each case is also designed for a center frequency of 2.6GHz using the commercial FDTD software,and the antenna parameters have been obtained using the same software.Table II shows the simulated results for each case.As seen above,the meandered isolating slot provides a signiﬁcantly broader bandwidth,higher gain and higher efﬁciency compared to other slot shapes and when no isolating slot is used.The variation of the gainof the antenna structure with meandered isolating slot is from3.3dB at 2.4GHz to2.87dB at2.8GHz with a maximum of3.6dB at2.6GHz.VI.C ONCLUSIONSA compact low mutual coupling MIMO antenna array for mobile handsets has been presented.The radiating elements are quarter wave-length slot antennas and a meander shaped slot has been used between the two antennas to isolate them.The measured and simulated S param-eters and the impedance Smith chart show that the meandered slot not only improves the isolation of the radiating elements,but also improves the bandwidth of the antennas signiﬁcantly by coupling to the excited antenna and introducing additional resonance frequency for it.The an-tenna structure covers a broad bandwidth between2.4–2.84GHz,suit-able for LTE2.6GHz and WLAN2.5GHz.The diversity parameters of the array have been evaluated,which show a very good diversity performance.The measured and simulated radiation performance of the proposed array has been evaluated.The simulated performance has been compared with the ones of a similar two element slot array,but with other shapes of isolating slot.The results show that the proposed design has obvious advantages over other isolating slot shapes in terms of bandwidth,efﬁciency,isolation and diversity performance.R EFERENCES[1]W.C.Y.Lee,Mobile Communications Engineering.New York:Wiley,1982.[2]R.G.Vaughan and J.B.Andersen,“Antenna diversity in mobile com-munications,”IEEE Trans.Veh.Technol.,vol.36,pp.149–172,Nov.1987.[3]J.S.Colburn,Y.Rahmat-Samii,M.A.Jensen,and G.J.Pottie,“Eval-uation of personal communications dual antenna handset diversityperformance,”IEEE Trans.Veh.Technol.,vol.47,pp.737–746,Aug.1998.[4]S.Lu,T.Hui,and M.Bialkowski,“Optimizing MIMO channel capac-ities under the inﬂuence of antenna mutual coupling,”IEEE AntennasWireless Propag.Lett.,vol.7,pp.287–290,2008.[5]A.Diallo,C.Luxey,P.Le Thuc,R.Staraj,and G.Kossiavas,“En-hanced two-antenna structures for universal mobile telecommunica-tions system diversity terminals,”IET Microw.,Antennas Propag.,vol.2,pp.93–101,Feb.2008.[6]J.Rahola and J.Ollikainen,“Analysis of isolation of two-port antennasystems using simultaneous matching,”in Proc.Eur.Conf.on An-tennas and Propagation:EuCAP,Edinburgh,U.K.,Nov.2007,pp.11–16.[7]C.-Y.Chiu,C.-H.Cheng,R.D.Murch,and C.R.Rowell,“Reductionof mutual coupling between closely packed antenna elements,”IEEETrans.Antennas Propag.,vol.55,pp.1732–1738,Jun.2007.[8]F.Yang and Y.Rahmat-Samii,“Microstrip antennas integrated withelectromagnetic band-gap(EBG)structures:A low mutual couplingdesign for array applications,”IEEE Trans.Antennas Propag.,vol.51,pp.2936–2946,Oct.2003.[9]M.Karaboikis,C.Soras,G.Tsachtsiris,and V.Makios,“Compactdual-printed inverted F antenna diversity systems for portable wire-less devices,”IEEE Antennas Wireless Propag.Lett.,vol.3,pp.9–14,2004.[10]H.-T.Chou,H.-C.Cheng,H.-T.Hsu,and L.-R.Kuo,“Investigationsof isolation improvement techniques for multiple input multiple output(MIMO)WLAN portable terminal applications,”Progr.Electromagn.Res.,vol.PIER85,pp.349–366,2008.[11]K.Kim,W.Lim,and J.Yu,“High isolation internal dual band planarinverted-F antenna diversity system with band-notched slots for MIMOterminals,”in Proc.36th Eur.Microwave Conf.,2006,pp.1414–1417.[12]M.Ayatollahi,Q.Rao,and D.Wang,“Wideband High Isolation TwoPort Antenna Array for Multiple Input Multiple Output Handheld De-vices,”U.S.patent8085202.[13]T.Taga,“Analysis for mean effective gain of mobile antennas in landmobile radio environments,”IEEE Trans.Veh.Technol.,vol.39,pp.117–131,May1990.Experimental Characterization of a BroadbandTransmission-Line Cloak in Free SpacePekka Alitalo,Ali E.Culhaoglu,Andrey V.Osipov,Stefan Thurner,Erich Kemptner,and Sergei A.Tretyakov Abstract—The cloaking efﬁciency of aﬁnite-size cylindrical transmis-sion-line cloak operating in the X-band is veriﬁed with bistatic free space measurements.The cloak is designed and optimized with numerical full-wave simulations.The reduction of the total scattering width of a metal ob-ject,enabled by the cloak,is clearly observed from the bistatic free space measurements.The numerical and experimental results are compared re-sulting in good agreement with each other.Index Terms—Scattering,scattering cross section.I.I NTRODUCTIONThe transmission-line cloak concept has been recently introduced [1],[2]as an alternative to the transformation-optics[3]–[7]and scat-tering cancellation approaches[8]–[10].In addition to these,there exist several other cloaking techniques and variations of these concepts.A detailed overview can be found,e.g.,in recent review papers[2],[6], [7],[9].Instead of utilizing anisotropic(and often resonant)metamaterials [6],[7]or plasmonic materials[9],the transmission-line cloak enables the electromagnetic wave to smoothly travel through the cloaked ob-ject inside a volumetric network of transmission lines,resulting in a simple and cheap way to obtain broadband cloaking of objects with se-lected geometries.It should be emphasized that the transmission-line cloak can only“hide”objects thatﬁt inside the volumetric network of transmission lines[2],i.e.,these objects cannot be bulky and electri-cally large objects.The technique allows cloaking of arrays of electri-cally small objects or mesh-like objects that let transmission lines go through them.A clear distinction should be made between cloaks that can hide an object in free space and the so-called ground-plane cloaks that can be used to hide an object above a boundary[11].In ground-plane cloaks the complexity of the material parameters is not as demanding as in cloaks operating in free space.Recent developments in ground-plane cloaks show that it is possible to realize such devices even for large objects operating within the visible frequency spectrum[12]–[15].In this work we study aﬁnite-size,three-dimensional transmission-line cloak that can hide a three-dimensional metallic object from elec-tromagnetic waves in free space.The basic cloak geometry is known from previous results[2]and the dimensions of the cloak are here op-timized for operation in the X-band(8GHz–12GHz).The previous realizations of the cylindrical transmission-line cloak utilized a cou-pling layer made of widening metal strips to couple the electromagnetic Manuscript received October07,2011;revised January18,2012;accepted May11,2012.Date of publication July10,2012;date of current version October 02,2012.This work was supported in part by the Academy of Finland and Nokia through the centre-of-excellence program.The work of P.Alitalo was supported by the Academy of Finland via post-doctoral project funding.P.Alitalo and S.A.Tretyakov are with the Department of Radio Science and Engineering/SMARAD Centre of Excellence,Aalto University School of Elec-trical Engineering,FI-00076Aalto,Finland(e-mail:pekka.alitalo@aalto.ﬁ).A.E.Culhaoglu,A.V.Osipov,S.Thurner and E.Kemptner are with the Microwaves and Radar Institute,German Aerospace Center(DLR),82234 Wessling,Germany.Color versions of one or more of theﬁgures in this communication are avail-able online at .Digital Object Identiﬁer10.1109/TAP.2012.22073390018-926X/$31.00©2012IEEE。

TE 品牌 4900-5875MHz单频带天线（适用于IEEE 802.11 a g n ac等Wi

Dimensions are in inches and milli-meters unless otherwise specified. Values in brackets are metric equivalents.Dimensions are shown for reference purposes only.Specifications subject to A: 1-800-522-6752Canada: 1-905-475-6222Germany: +49 (0) 6251-133-1999For other country numbers go to:UK: +44 (0) 800-267666Netherlands: +31 (0) 73-6246-999China: +86 (0) /supportcenterCatalog: 1-1773726-7Revised 02-14Part Number: 2118326-1Product FactsSmall and lightweight PCB antenna • assembly.RoHS compliant.• RecommendationsSpecificationsFor best performance follow Mounting • Guide and Keep Out Area on next page.Frequency Range (MHz)4900–5875Peak Gain +2.4 dBi VSWR< 2.5:1PolarizationLinearAzimuth Beamwidth Omni-directional Power Handling3 Watt cwFeed Point Impedance 50 Ohms unbalancedSize 15.0 mm x 10.0 mm x 1.0 mm Weight 1.0 g.Mounting Adhesive. See diagram on page 2.Keep Out Area See diagram on page 2.Cable / Connector120 mm length. 1.13 mm dia. with U.FL connectorTest Orientation in Free Space AzimuthElevationEfficiencyVSWR15dBi 0dBi -5dBi -10dBi -15dBi -20dBi -25dBi-30dBi90º270º180º0ºLegend:5470 MHz270ºLegend:5470 MHzDimensions are in inches and milli-meters unless otherwise specified. Values in brackets are metric equivalents.Dimensions are shown for reference purposes only.Specifications subject to A: 1-800-522-6752Canada: 1-905-475-6222Germany: +49 (0) 6251-133-1999For other country numbers go to:UK: +44 (0) 800-267666Netherlands: +31 (0) 73-6246-999China: +86 (0) /supportcenterCatalog: 1-1773726-7Revised 02-14Part Number: 2118326-1© 2014 TE Connectivity family of companies. All Rights Reserved.1-1773726-7–CSD–PDF–02-14TE (logo) and TE Connectivity are trademarks.Wi-Fi is trademark. Other logos, products and company names mentioned herein may be trademarks of their respective owners.While TE has made every reasonable effort to ensure the accuracy of the information in this catalog, TE does not guarantee that it is error-free, nor does TE make any other representation,warranty or guarantee that the information is accurate, correct, reliable or current. TE reserves the right to make any adjustments to the information contained herein at any time without notice.TE expressly disclaims all implied warranties regarding the information contained herein, including,but not limited to, any implied warranties of merchantability or fitness for a particular purpose.The dimensions in this catalog are for reference purposes only and are subject to change without notice. Specifications are subject to change without notice.Consult TE for the latest dimensions and design specifications.9565 Soquel Drive Aptos, CA 95003(+ 831) Mounting Guide and Keep Out AreaNOTES:Dimensions: mm Diagram is not to scaleAntenna designed to be mounted on plastic cover.1. Area in 2. blue above indicates Keep Out Area.For more inform 3. ation please call TE.2。

Belkin_et_al-2011-Microwave_and_Optical_Technology_Letters

deviation is observed.This acceptable difference can be attrib-uted to the inﬂuence of the SMA connectors,the imperfect sol-dering as well as the nonuniformity of the substrate’s properties that normally affect theﬁlter’s performance.4.CONCLUSIONSA simple tunable bowtie-like EBG-based device has been designed and engineered in this article.Modiﬁcation of the resulting stopband can be conducted by properly conﬁguring the EBG unit cell and desired levels of reconﬁgurability are viable through RF-MEMS switches.Numerical investigation proves a controllable bandstop behavior extending to DC,whereas the easily contrived prototype allows the extraction of promising ex-perimental data that successfully verify the simulation results. ACKNOWLEDGMENTSThis work was supported by the National Scholarships Foundation of Greece(IKY),under Grant No.5282.REFERENCES1.H.J.de Los Santos,RF MEMS circuit design for wireless communi-cations,Artech House,Boston,MA,2002.2.L.Li and D.Uttamchandani,MEMS microwave device with switch-able capacitive and inductive states,IET Micro Nano Lett3(2008), 77–81.3.Z.Brito-Brito,I.Llamas-Garro,and L.Pradell,Precise frequencyand bandwidth control of microstrip switchable bandstopﬁlters, Microwave Opt Technol Lett51(2009),2573–2578.4.F.Yang and Y.Rahmat-Samii,Electromagnetic band gap structuresin antenna engineering,Cambridge University Press,London,UK, 2008.5.J.J.Yu,B.T.Tan,and S.T.Chew,CPW EBG using butterﬂy-radialslot(BRS)for low-pass widebandﬁltering,Microwave Opt Technol Lett41(2004),320–323.6.S.Y.Huang and Y.H.Lee,A tapered small-size EBG microstripbandstopﬁlter design with triple EBG structures,Microwave Opt Technol Lett46(2005),154–158.7.H.Liu,L.Sun,and T.Yoshimasu,Slot resonator-based electromag-netic bandgap coplanar waveguide and itsﬁlter application,Phys Lett A359(2006),171–174.8.M.F.Karim,A.-Q.Liu,A.Yu,and A.Alphones,Micromachinedtunableﬁlter using fractal electromagnetic bandgap(EBG)struc-tures,Sens Actuators A133(2007),355–362.9.L.Liang,C.H.Liang,L.Chen,and X.Chen,A novel broadbandEBG using cascaded mushroom-like structure,Microwave Opt Technol Lett50(2008),2167–2170.10.N.Entesari and G.M.Rebeiz,A differential4-bit6.5–10-GHz RFMEMS tunableﬁlter,IEEE Trans Microwave Theory Tech53 (2005),1103–1110.11.A.Takacs,D.Neculoiu,D.Vasilache,A.Muller,P.Pons,L.Bary,P.Calmon,H.Aubert,and R.Plana,Tunable bandstop MEMSﬁlter for millimetre-wave applications,Electron Lett43(2007),675–677.12.C.Y.Ong and M.Okoniewski,Low-loss MEMS switchable micro-stripﬁlters,Microwave Opt Technol Lett50(2008),2557–2561. 13.R.Kaunisto,K.Reimann,K.Boyle,J.Krogerus,Z.Liu,P.G.Stee-neken,and J.Ollikainen,2.0–2.7GHz programmable bandpassﬁlter with RF-MEMS capacitance matrices,Electron Lett45(2009), 738–739.14.M.A.El-Tanani and G.M.Rebeiz,High-performance1.5–2.5-GHzRF-MEMS tunableﬁlters for wireless applications,IEEE Trans Microwave Theory Tech58(2010),1629–1637.15.D.I.Karatzidis,T.V.Yioultsis,and E.E.Kriezis,Fast analysis ofphotonic crystal structures with mixed-order prism macroelements, IEEE J Lightwave Technol26(2008),2002–2009.V C2011Wiley Periodicals,Inc.TUNABLE RF-BAND OPTOELECTRONIC OSCILLATOR AND OPTOELECTRONIC COMPUTER-ADDED DESIGN MODEL FOR ITS SIMULATIONMikhail E.Belkin,1Alexei V.Loparev,1Yuliya Semenova,2 Gerald Farrell,2and Alexander S.Sigov11Moscow State Institute of Radio-Engineering,Electronics and Automation,Technical University,Moscow,Russia;Corresponding author:belkin@mirea.ru2Photonics Research Centre at the Dublin Institute of Technology, Dublin8,IrelandReceived10February2011ABSTRACT:A cost-effective tunable version of optoelectronic oscillator(OEO)that overcomes the well-known design tradeoffs in a traditional radio frequency(RF)oscillator is proposed.To validate the technique,a simple computerized model of the proposed OEO is used on the basis of commercial optoelectronic computer-added design tool.The model correctly simulates the phase noise close to the carrier.As an example,we demonstrate a3-GHz OEO with simulation results that are experimentally conﬁrmed by prototype measurements.A parametric comparison with an advanced wideband synthesizer integrated circuits (IC)is also presented.V C2011Wiley Periodicals,Inc.Microwave Opt Technol Lett53:2474–2477,2011;View this article online at .DOI10.1002/mop.26304Key words:RF-band optoelectronic oscillator;phase noise;OE-CAD simulation1.INTRODUCTIONThe optoelectronic oscillator(OEO)considered in this letter is one of the most attractive functional examples of microwave photonics devices aspiring to become a new class of highly sta-ble RF oscillators in a frequency range from hundreds of mega-hertz to hundreds of gigahertz[1].The key advantage of an OEO by comparison with traditional RF and microwave oscilla-tors,that is being focused on recently by most researchers,is the achievable remarkably high spectral purity[2].The latter increases with delay time[1]and long delays can be achieved easily with an opticalﬁber delay line length of several kilo-meters.Such OEOs have a wide area of potential applications inFigure6Simulation results versus measurement data(000and111 cases).[Colorﬁgure can be viewed in the online issue,which is avail-able at ]theﬁelds of measurement equipment[3],telecommunications[4], and precise sensors[5].It is these applications that are the main motivation for research and development of OEOs,given that the fundamental frequency operation limit of optoelectronic devices is hundreds of gigahertz[6],which is much higher than that of modern state of the art integrated microwave oscillators[7].In an early model of OEOs,a steady-state mode analysis was performed by a quasi-linear interpretation[7].This model has the advantage of simplicity but is valid only in a small-signal mode and does not allow accurate simulation of dynamic fea-tures in a nonlinear circuit such as OEO.To study the dynamics of an OEO,an accurate time-domain model taking account of the interaction between nonlinearity and time delay that allowed investigating analytically the stabil-ity properties of an OEO,has been developed[8].As an exten-sion of the work presented in Ref.8,the same group has per-formed a phase noise spectral simulation using the Langevin approach with experimental veriﬁcation of the results for an OEO with aﬁber length of4km[9].The state of the art for OEOs is currently considered to be a single-frequency(within the X-band)OEO product from OEwaves with a class leading phase noise level ofÀ163dBc/ Hz at a10-kHz offset from the carrier[10].In this work,we investigate another important beneﬁt of OEOs that allows one to overcome the well-known tradeoff in a tradi-tional RF oscillator between two key parameters:a wider tuning band and lower phase noise[11].Speciﬁcally,the former parame-ter is inversely proportional to the oscillator’s external Q-factor, whereas the latter one is directly proportional to the same parame-ter.On the other hand,the tuning band of an OEO is determined solely by the tuning capabilities of band-passﬁlter(BPF)and phase noise levels,which as noted above,are deﬁned by the length of the opticalﬁber.This feature of an OEO offers in principle a straight-forward way to produce super-high stability RF and microwave oscillators with multioctave tuning using discrete oscil-lation modes.However,there is one issue of importance,the trade-off between free spectral range(FSR)and phase noise.The FSR of an OEO is a function of the speed of light c,theﬁber core re-fractive index n,and the length of theﬁber loop L by Ref.5:FSR%c:(1)Therefore,increasing L will result not only in a reduction of the OEO’s phase noise but also in a reduction of the FSR so that there will be a larger number of very closely spaced oscilla-tion frequencies.As a result,it will be difﬁcult to isolate only one operating frequency with an RFﬁlter.This problem has been solved experimentally in Ref.3for high-precision mea-surement equipment using a multipleﬁber loop approach.In this scheme,three loops are used(with three microwave-band photo-diodes)resulting in a one-octave band OEO that can be tuned using a yttrium iron garnet(YIG)-ﬁlter from6to12GHz in steps of3MHz and which exhibited a phase noise as low as À128dBc/Hz at a10-kHz offset from the carrier.In our opinion,a more cost-effective remedy to mitigate this issue,in particular for applications in the area of telecommunica-tions,is the use of an OEO constructed using a single-loop of ﬁber with a moderate noise and a frequency tuning step of a few megahertz.To validate this technique,a simple computerized OEO model that provides the OEO’s spectral and phase noise characteristics is proposed and highlighted in sections which fol-low.The simulation results are also successfully validated by measurements on a prototype of the proposed OEO.A compari-son with the advanced wideband RF synthesizer that constitutes an alternative to the proposed solution is also presented.2.OPTOELECTRONIC OSCILLATOR COMPUTERIZED MODELThe typical layout of an OEO[7]with some realistic elements is presented in Figure1.The system consists of two principalFigure1The layout of OEO.[Colorﬁgure can be viewed in the online issue,which is available at ]Figure2The OEO schematic using the VPItransmission Maker OE-CAD tool(a)and test bench for OEO phase noise simulation(b).Time window coupler(TWC)is an auxiliary element allowing one to incre-ment the spectral resolution.[Colorﬁgure can be viewed in the online issue,which is available at ]Figure3Simulated OEO output spectrum(a)and spectra of the single-sideband OEO phase noises powers(in bandwidth of10Hz)at offsets from the carrier of10,100kHz,and1MHz.[Colorﬁgure can be viewed in the online issue,which is available at ]subsystems:optical and electrical.The optical subsystem includes a semiconductor laser module (SLM),electro-optical modulator (EOM),optical ﬁber line (OFL),and photodetector module (PDM).To ensure reliable operation of the OEO with low laser noise levels and low loss for the optical section,an optical isola-tor and polarization controller are introduced into the system.The electrical subsystem includes a low-noise RF ampliﬁer (LNA),BPF,power ampliﬁer (PA),and electrical coupler.The operating principle of the OEO is based on the conversion of the continu-ous optical radiation energy emitted from the SLM into energy inthe RF-band.To achieve this,the EOM is controlled by an optoe-lectronic positive feedback loop (Fig.1).The OEO carrier fre-quency in the RF-band is determined by the BPF’s central fre-quency,while the overall energy storage time of OEO depends on the ﬁber delay.By managing the ﬁber length and gain of the LNA and PA,one can ensure,for a given circuit,the amplitude and phase balance required for self-sustained oscillation.To our knowledge,to date,there is no speciﬁc optoelectronic computer-added design (OE-CAD)tool for microwave photonic device simulation.In the case of an OEO,the critical issue is the realization of the feedback loop and for this,simultaneous simulation of both the optical and RF elements is necessary.A comparative analysis showed that the most suitable tool for OEO modeling is VPI System’s VPItransmission Maker TM [12].A schematic layout of a computerized model is depicted in Fig-ure 2(a)for the single-loop OEO as shown in Figure 1.The modeling is performed with a periodic boundary conditions that allow concurrent simulation of the RF and optical elements of the OEO in an object-oriented environment.TABLE 1OEO and PLL Synthesizer ComparisonFrequency Range (GHz)Output Power (dBm)Parasitic Suppression(dB)Phase Noise (dBc/Hz)at 3GHz;Offset of 10kHz 100kHz 1MHz OEO prototype 2.5À15847À104À120À138ADF4350(fundamental mode)2.2À4.4513À92À111À134Figure 4PSA-measured OEO prototype spectrum (a)and phase noise (b)characteristics.[Color ﬁgure can be viewed in the online issue,which is available at ]In the schematic,all the elements of Figure1have a speciﬁc interpretation.For example,the SLM is represented by single-mode Fabri–Perot rate equations-based laser model,the EOM model is based on a differential Mach–Zehnder interferometer model,and the OFL is modeled by a combination of an optical attenuator and a delay line.The spectrum of the OEO output signal is extracted by a spectrum analyzer model.The speciﬁc test bench realization for simulating OEO phase noise character-istics in the bandwidth of10Hz at offsets of10,100kHz,and 1MHz from the carrier,is presented in Figure2(b).3.SIMULATION RESULTSThe oscillation spectrum in the steady-state mode for the OEO schematic depicted in Figure2(a)has been simulated for the fol-lowing parameters of the model elements:the optical power of the SLM is11.5mW at a wavelength of1550nm,RIN¼À150 dB/Hz;the EOM’s optical insertion loss is5dB,voltage bias is 2.5V,modulation bandwidth is30GHz;the OFL’s delay time is s¼314.4ns corresponding to aﬁber length of65m;the PDM responsivity is0.8A/W;the LNA’s noiseﬁgure is3dB. The BPF is represented by a fourth-order Butterworthﬁlter with a centre frequency of3GHz and3-dB bandwidth of12MHz.The simulated OEO output spectrum is presented in Figure 3(a),and the spectra of the single-sideband phase noise compo-nents at offsets from the carrier of10,100,and1MHz are shown in Figure3(b).Note that from Figure3(a),the OEO out-put power is close toþ5dBm,the suppression ratio of the adja-cent modes is close to47dB.From Figure3(b),the relative phase noise levels of the simulated OEO are S10kHz¼À105 dBc/Hz,S100kHz¼À121dBc/Hz,and S1MHz¼À139dBc/Hz.4.EXPERIMENTAL VERIFICATIONFor an experimental veriﬁcation of simulated results,an OEO prototype has been constructed with the frequency tuning range of2.5–15GHz with a step value about2.5MHz.The major pro-totype’s elements are SLM:MQW DFB laser diode LDI H-DFB-1550(1.55l m,power output up to17mW,RIN¼À155 dB/Hz)from IIT;EOM:LiNbO3Mach–Zehnder optical modula-tor MATH-005-40(1.55l m,typical insertion loss4.5dB,3-dB bandwidth of35GHz)from COVEGA;PDM:pin-photodiode XPDV2120R(wavelength range1480–1620nm,responsivity 0.6A/W,3-dB bandwidth50GHz)from u2t Photonics.For the electrical section,a tunable band-pass YIG-ﬁlter(tuning range 1.5–15GHz,insertion loss less than5dB,3-dB bandwidth is11 MHz at1.5GHz and20MHz at15GHz)from Magneton,is used as a BPF.Removable sets of two ampliﬁers(total gain of near50dB,noiseﬁgure3.5dB)for the frequency bands2.5–8 and8–15GHz are used.A sample of the measured OEO output spectrum and phase noise characteristics at the oscillation frequency of3GHz is presented in Figure4.As seen from Figure4(a),P osc¼8dBm, the adjacent mode suppression ratio is47dB and the experimen-tal phase noises are[Fig.4(b)]S10kHz¼À104dBc/Hz,S100kHz ¼À120dBc/Hz,and S1MHz¼À138dBc/Hz.These experi-mental results closely match the simulation results and conﬁrm the feasibility of the proposed OEO model.PARISON WITH A MICROWAVE SYNTHESIZERA wholly electronic microwave counterpart to the proposed so-lution is a phase locked loop(PLL)synthesizer.To validate the beneﬁts of the OEO design presented here,Table1lists a brief technical comparison of the OEO prototype with IC ADF4350,an advanced wideband synthesizer with integrated voltage-con-trolled oscillator from Analog Devices[13].It is clear from Table1,that the OEO prototype provides an extremely wide tuning range(2.5octaves)with a comparable output power,very much stronger parasitic suppression and a 10-dB lower phase noise on average.6.CONCLUSIONA novel concept of an OEO design for telecommunication appli-cations is proposed and is validated using a simple computerized model,successfully realized using the OE-CAD VPItransmission Maker TM tool.The experimental data for a prototype of the OEO show a goodﬁt to the simulation results that conﬁrms the validity of the proposed model and the feasibility of the pro-posed OEO.The developed model can be also being applied to the simulation of different OEO types,for example,multiple-loop OEOs.ACKNOWLEDGMENTSThe authors would like to express their thanks to Cristina Arellano, Dr.Hadrien Louchet,and Jim Farina of VPI Systems for fruitful discussions regarding OEO phase noise simulation singularities. This work was supported by the Russian Federation Ministry of Education and Science program‘‘Progress of scientiﬁc potential of the higher school(2009–2011).’’REFERENCES1.X.S.Yao and L.Maleki,Optoelectronic oscillator for photonic sys-tems,IEEE J Quantum Electron32(1996),1141–1149.2.L.Maleki,Recent progress in opto-electronic oscillator,In:Micro-wave Photonics International Topical Meeting,October12–14, 2005,pp.81–84.3.D.Eliyahu and L.Maleki,Tunable,ultra-low phase noiseYIG based opto-electronic oscillator,In:IEEE/MTT-S Digest, Philadelphia,PA,2003,pp.2185–2187.4.M.Shin and P.Kumar,1.25Gbps optical data channel up-conver-sion in20GHz-band via a frequency-doubling optoelectronic oscil-lator for radio-over-ﬁber systems,In:IEEE/MTT-S,Honolulu,HI, June2007,pp.63–66.5.N.L.Duy,B.Journet,I.Ledoux-Rak,et al.,Opto-electronic oscilla-tor:Applications to sensors,In:International Topics’Meeting on 2008Asia–Paciﬁc Microwave Photonics Conference,Hong Kong, China,MWP/APMP,2008,pp.131–134.6.A.J.Seeds and K.J.Williams,Microwave photonics,IEEE/OSA JLightwave Technol241(2006),4628–4641.7.X.S.Yao,Opto-electronic oscillators,In:W.S.C.Chang(Ed.),Book:RF photonic technology in opticalﬁber links,Cambridge University Press,Cambridge,England,2002,pp.255–292.8.Y.K.Chembo,rger,H.Tavernier,et al.,Dynamical instabil-ities of microwaves generated with optoelectronic oscillators,Opt Lett32(2007),2571–2573.9.Y.K.Chembo,K.Volyanski,rger,et al.,Determination ofphase noise spectra in optoelectronic microwave oscillators:A Lan-gevin approach,IEEE J Quantum Electron45(2009),178–186. 10./downloads/Adv%20OEO%20PB_042610.pdf.11.M.E.Belkin and L.Belkin,Microwave opto-electronic oscillatorresearch,In:Proc.SIBIRCON-2010,Listvyanka,Russia,July2010, pp.589–593.12..13./static/imported-ﬁles/data_sheets/ADF4350.pdf.V C2011Wiley Periodicals,Inc.。

基于特征模理论的低剖面MIMO立方体天线

基于特征模理论的低剖面MIMO立方体天线作者：于琪陈益凯杨仕文来源：《南京信息工程大学学报（自然科学版）》2019年第01期摘要基于特征模理论，给出了一种适用于室内环境的16端口多输入多输出（MIMO）立方体天线设计.首先利用特征模理论在对一个矩形金属片进行模式分析的基础上，同时激励金属片的不同模式，设计了一款工作于5.150～5.875 GHz的高隔离度的4端口MIMO天线单元，并引入人工磁导体（AMC）表面代替原天线的地板，大大降低了天线的剖面.进一步地，围绕立方体环绕一周组成4×4端口的MIMO立方体天线，在较小的空间内实现了天线的多端口与多极化.仿真和测试结果表明：天线在5.150～5.875 GHz频段内端口反射系数Sii< -10 dB，端口间隔离度Sij>20 dB.关键词特征模理论;MIMO立方体天线;人工磁导体;低剖面中图分类号TN820文献标志码A0引言随着无线通信系统的快速发展，有限的频谱资源与无线通信系统对信道容量不断增长的需求之间的矛盾日益加深.而多输入多输出（MIMO）天线由于其在不增加额外功率及频谱资源的前提下可通过多径传输提升系统信道容量的特性得到广泛应用[1-3]，因此具有良好的应用前景.现在由于有限空间资源的限制，MIMO天线的小型化受到了越来越高的重视.传统的较多端口的MIMO天线一般是平面阵列[4]，这样的MIMO天线占用空间较大且极化种类较少.而MIMO立方体天线可以在较小的空间内实现多个天线的集成以及极化多样性，在天线小型化方面有着明显的优势.文献[5]将12个偶极子安置在立方体的12条边上，在0.5λ×0.5λ×0.5λ的空间内实现了12端口的MIMO天线，文献[6]则在0.76λ×0.76λ×0.76λ的空间内实现了工作于2.40～2.48 GHz的18端口的MIMO天线，但它们所提出的天线带宽均很窄.MIMO天线的小型化会增强其端口间的耦合，所以在MIMO天线的設计中，在有限空间内实现各个单元间的去耦尤其重要.在现有的文献中提出了很多去耦技术：可通过正交排布天线单元[7]，减小端口间的近场耦合，从而提高端口间的隔离度，但是这种方法具有较大的局限性;也可通过引入去耦网络达到去耦效果[8]，但引入去耦网络的同时还需引入匹配网络，这无疑会增加天线的复杂度.比较常见的一种方法是在地板上开适当长度的槽来充当滤波器[9]，滤除部分耦合分量，或是引入电子带隙（EBG）结构[10]，抑制表面波的传播，提高天线的隔离度，当然这种方法也是引入其他结构，也会增加设计难度.本文采用了基于特征模理论的去耦方法.首先通过分析天线自身的模式电流，得到了5种显著的相互正交的模式，通过同时激励起这5种相互正交的模式得到一个高隔离度的4端口的MIMO天线单元.该去耦合方法基于天线自身特性，不引入其他结构，大大简化了天线的设计难度.然后，为了降低天线剖面加入AMC结构将天线剖面降为原来的34%，并将该天线组成立方体结构，而各单元空间位置的正交性确保端口间的隔离度没有恶化，从而得到了16端口的小型化MIMO立方体天线.1矩形金属片的模式分析特征模理论是Garbacz[11]在1965年首次提出的.一个物体的特征模指的是用于描述物体表面电流的一系列完全正交的电流，即特征电流Jn.同时这些电流所对应的辐射远场也是相互正交的，而特征电流Jn可由下面的矩阵方程计算得出：式中，X和R分别表示的是电场积分方程阻抗矩阵的虚部和实部，λn是Jn所对应的特征值.λn的范围从-到+，而它的模值λn决定了一个辐射系统所存储的能量，λn越大，系统所储存的能量越多，反之亦然.在辐射与散射问题中，λn=0时的情况尤其重要，此时的模式称为谐振模式，对应系统储存能量为零时的情况，而本文中所需关注的也就是λn接近于零的模式.模式显著性（MS，其量值记为SM）是用于描述当物体受到外部激励时，若每一个模式均被理想地激励时，它们对整体电流分布的贡献率，可用以下方程表示：定义SM≥12，即λn≤1的模式为显著模式，反之为不显著模式.本文中对一个矩形金属片在5～6 GHz进行特征模分析.该金属片尺寸为0.85λ0×0.85λ0（λ0为5.5 GHz所对应的真空中的波长），在其正下方距离0.38λ0处放置一地板，得到的矩形金属片特征模的模式显著性随频率变化的曲线如图1所示，可看出该矩形金属片的显著模式，即SM≥12的模式有5个，为图1中所标示的Mode1—Mode5.图2给出了该矩形金属片5个显著模式随频率变化的模式电流分布和辐射方向图，可以看出在5～6 GHz频段间，这5个显著模式的模式电流分布和辐射方向图均只有微小的变化.为了激励出Mode1—Mode5这5个模式，在金属片电流最大处进行开T形槽馈电，即在矩形金属片4个边的中心处以及4个角处进行开槽馈电，所得到的矩形金属片电流分布如图3所示.值得注意的是，Mode3和Mode5的馈电位置发生重叠，为了简化设计，可将这2种模式合在一起构成新的Mode3.2天线单元的设计图4给出了天线单元馈电端口的设计方案，给其中每一组端口以图中所示方向电流馈以等幅同相的电流，即可激励出所需的4种模式.以此原则设计馈电网络，采用一分二Wilkinson功分器连接Port11与Port12馈以等幅同相的电流可激励Mode1;采用0°和180°环形电桥连接Port21和Port22（Port31和Port32），分别馈以等幅同相和等幅反相的电流，即可分别激励出Mode2和Mode3;采用一分四Wilkinson功分器连接Port41、Port42、Port43和Port44，馈以等幅同相的电流，便可激励出Mode4.本文所设计的天线在HFSS中完成仿真设计，仿真模型如图5a所示.在同轴线处以理想集总端口激励，将所得全波仿真数据导入HFSS自带的2D电路中，利用其中自带的功分器与0°和180°环形电桥进行馈电，所得到的天线单元4个端口的反射系数与隔离度如图5b所示.由图5b可以看出，在5.150～5.875 GHz整个频段内，4个端口的反射系数Sii<-10 dB，阻抗匹配良好，而各个端口之间的隔离度|Sij|>25 dB，端口间的互耦很小，这正是由特征模的相互正交性所带来的.3低剖面天线单元设计前文所设计的天线单元采用理想导电体平面作为反射板，天线距离反射板高度为0.38λ0，约为21 mm.人工磁导体表面是一种周期性结构，具有反射同相的特性，可以代替传统的理想导电体平面作为天线的反射板，非常适用于低剖面天线的设计[12].为了降低天线剖面，本文采用图6a所示的AMC表面单元，为一“井”字形贴片敷在厚度为3.175 mm、介电常数为4.5的TP-2的介质板上，上方放置一厚度为1.58 mm、介电常数为2.17的TLY-5A的介质板.由图6b可以看出，该AMC结构在5.150～5.875 GHz频段内反射相位均在+90°～-90°之间，可作为天线的反射板.最终采用由8×8个AMC单元构成的AMC表面作为地板，其中为了避免同轴线对AMC表面的影响，去掉了Port4的4根同轴线所穿过的4个AMC单元，所得到的天线距离地板尺寸为7.175 mm（含介质板厚度），天线剖面缩减了66%.由前文给出的天线馈电网络设计原理设计了一分二功分器、一分四功分器与0°和180°环形电桥.为了缩减天线的尺寸，将地板尺寸缩减为1.05λ0×1.05λ0，同时将馈电网络放置在2层厚度为0.508 mm、介电常数为3.5的RF-35的介质板上，第1层放置一个0°和180°环形电桥，第2层放置一个一分二功分器与一个一分四功分器（如图7所示）.最终所得天线单元结构如图8所示，其端口S参数仿真结果如图9所示，在工作频段内端口反射系数Sii< -10 dB，隔离度Sij>20 dB.44×4端口的MIMO立方体天线设计将4个最终得到的低剖面天线单元环绕立方体一周组成一个尺寸为1.26λ0×1.26λ0×1.05λ0的MIMO立方体天线，如图10a所示.这样的设计在较小的空间内实现了16端口的MIMO天线设计，并且增加了极化多样性，非常适合室内环境.基于天线仿真模型，加工了实物模型，如图10b所示，测试结果如图11所示.由于该立方体天线的对称特性，在图11中只给出了A面天线单元端口的S参数和A、B面间端口隔离度以及A、C面间端口隔离度的测试结果.由测试结果可看出，A面端口的反射系数在5.150～5.875 GHz频段间均小于-10 dB，隔离度均大于20 dB，而A面与B面、C面之间端口隔离度均大于30 dB，因此可认为该立方体天线在5.150～5.875 GHz频段内所有端口反射系数均小于-10 dB，端口隔离度大于20 dB.同时图12给出了该立方体天线其中一个单元在xoz面和yoz面上的辐射方向图的测试结果与仿真结果，可看出一个天线单元的4个端口可以激励出多种极化，而将单元环绕一周所得到的立方体天线则会具有更多的极化，可以更好地接收来自各个方向的信号.为了衡量本文所设计的MIMO天线在使用时的分集性能，计算了天线端口间的包络相关系数[13]，计算所用的公式如下：式中ρi，j为端口i和端口j间的包络相关系数，Ei（θ，φ）为端口i激励其他端口匹配时的辐射远场，Ej（θ，φ）为端口j激励其他端口匹配时的辐射远场.由图11的测试结果可以看出，各单元间的端口耦合非常小，单元间几乎没有影响，故本文中只计算了单元内部端口间的包络相关系数.同时，由于用现有测试设备测试3D方向图比较困难且精度不高，而天线测试结果与仿真结果较为贴合，所以采用仿真结果计算包络系数，所得结果如图13所示.一般情况下MIMO天线各端口之间的包络相关系数小于0.5时，才有较为显著的分集效果[14]，由图13可以看出该天线端口间包络相关系数均小于0.5，即可认为它有良好的分集性能.5结束语本文基于特征模理论设计了一款工作于5.150～5.875 GHz、尺寸为1.26λ0×1.26λ0×1.05λ0的16端口MIMO立方体天线.该天线利用特征模理论实现了天线单元的多极化与各端口间的高隔离度，并通过加载AMC表面实现了天线单元的低剖面，从而在较小空间内实现了多端口、高隔离度、多极化的MIMO立方体天线.实验结果表明该天线在工作频段内反射系数小于-10 dB，端口間隔离度大于20 dB，满足MIMO天线的设计要求.参考文献References[1]Foschini G J，Gans M J.On Limits of wireless communications in a fading environment when using multiple antennas[J].Wireless Personal Communications，1998，6（3）：311-335[2]帅吉莉.电脑MIMO天线应用简介[J].科技创新导报，2011（1）：22SHUAI Jili.Introduction to computer MIMO antenna application[J].Science and Technology Innovation Herald，2011（1）：22[3]Bae H，Harackiewicz F J，Park M J，et pact mobile handset MIMO antenna for LTE700 applications[J].Microwaveand Optical Technology Letters，2010，52（11）：2419-2422[4]Yang BQ，Yu Z Q，Dong Y Y，et pact tapered slot antenna array for 5G millimeter-wave massive MIMO systems[J].IEEE Transactions on Antennas and Propagation，2017，65（12）：6721-67273低剖面天线单元设计前文所设计的天线单元采用理想导电体平面作为反射板，天线距离反射板高度为0.38λ0，约为21 mm.人工磁导体表面是一种周期性结构，具有反射同相的特性，可以代替传统的理想导电体平面作为天线的反射板，非常适用于低剖面天线的设计[12].为了降低天线剖面，本文采用图6a所示的AMC表面单元，为一“井”字形贴片敷在厚度为3.175 mm、介电常数为4.5的TP-2的介质板上，上方放置一厚度为1.58 mm、介电常数为2.17的TLY-5A的介质板.由图6b可以看出，该AMC结构在5.150～5.875 GHz频段内反射相位均在+90°～-90°之间，可作为天线的反射板.最终采用由8×8个AMC单元构成的AMC表面作为地板，其中为了避免同轴线对AMC表面的影响，去掉了Port4的4根同轴线所穿过的4个AMC单元，所得到的天线距离地板尺寸为7.175 mm（含介质板厚度），天线剖面缩减了66%.由前文给出的天线馈电网络设计原理设计了一分二功分器、一分四功分器与0°和180°环形电桥.为了缩减天线的尺寸，将地板尺寸缩减为1.05λ0×1.05λ0，同时将馈电网络放置在2层厚度为0.508 mm、介电常数为3.5的RF-35的介质板上，第1层放置一个0°和180°环形电桥，第2层放置一个一分二功分器与一个一分四功分器（如图7所示）.最终所得天线单元结构如图8所示，其端口S参数仿真结果如图9所示，在工作频段内端口反射系数Sii< -10 dB，隔离度Sij>20 dB.44×4端口的MIMO立方体天线设计将4个最终得到的低剖面天线单元环绕立方体一周组成一个尺寸为1.26λ0×1.26λ0×1.05λ0的MIMO立方体天线，如图10a所示.这样的设计在较小的空间内实现了16端口的MIMO天线设计，并且增加了极化多样性，非常适合室内环境.基于天线仿真模型，加工了实物模型，如图10b所示，测试结果如图11所示.由于该立方体天线的对称特性，在图11中只给出了A面天线单元端口的S参数和A、B面间端口隔离度以及A、C面间端口隔离度的测试结果.由测试结果可看出，A面端口的反射系数在5.150～5.875 GHz频段间均小于-10 dB，隔离度均大于20 dB，而A面与B面、C面之间端口隔离度均大于30 dB，因此可认为该立方体天线在5.150～5.875 GHz频段内所有端口反射系数均小于-10 dB，端口隔离度大于20 dB.同时图12给出了该立方体天线其中一个单元在xoz面和yoz面上的辐射方向图的测试结果与仿真结果，可看出一个天线单元的4個端口可以激励出多种极化，而将单元环绕一周所得到的立方体天线则会具有更多的极化，可以更好地接收来自各个方向的信号.为了衡量本文所设计的MIMO天线在使用时的分集性能，计算了天线端口间的包络相关系数[13]，计算所用的公式如下：式中ρi，j为端口i和端口j间的包络相关系数，Ei（θ，φ）为端口i激励其他端口匹配时的辐射远场，Ej（θ，φ）为端口j激励其他端口匹配时的辐射远场.由图11的测试结果可以看出，各单元间的端口耦合非常小，单元间几乎没有影响，故本文中只计算了单元内部端口间的包络相关系数.同时，由于用现有测试设备测试3D方向图比较困难且精度不高，而天线测试结果与仿真结果较为贴合，所以采用仿真结果计算包络系数，所得结果如图13所示.一般情况下MIMO天线各端口之间的包络相关系数小于0.5时，才有较为显著的分集效果[14]，由图13可以看出该天线端口间包络相关系数均小于0.5，即可认为它有良好的分集性能.5结束语本文基于特征模理论设计了一款工作于5.150～5.875 GHz、尺寸为1.26λ0×1.26λ0×1.05λ0的16端口MIMO立方体天线.该天线利用特征模理论实现了天线单元的多极化与各端口间的高隔离度，并通过加载AMC表面实现了天线单元的低剖面，从而在较小空间内实现了多端口、高隔离度、多极化的MIMO立方体天线.实验结果表明该天线在工作频段内反射系数小于-10 dB，端口间隔离度大于20 dB，满足MIMO天线的设计要求.参考文献References[1]Foschini G J，Gans M J.On Limits of wireless communications in a fading environment when using multiple antennas[J].Wireless Personal Communications，1998，6（3）：311-335[2]帅吉莉.电脑MIMO天线应用简介[J].科技创新导报，2011（1）：22SHUAI Jili.Introduction to computer MIMO antenna application[J].Science and Technology Innovation Herald，2011（1）：22[3]Bae H，Harackiewicz F J，Park M J，et pact mobile handset MIMO antenna for LTE700 applications[J].Microwaveand Optical Technology Letters，2010，52（11）：2419-2422[4]Yang BQ，Yu Z Q，Dong Y Y，et pact tapered slot antenna array for 5G millimeter-wave massive MIMO systems[J].IEEE Transactions on Antennas and Propagation，2017，65（12）：6721-6727。

基于EBG结构的天线阵降耦合效应研究

基于EBG结构的天线阵降耦合效应研究辛毅;刘运林【摘要】Based on EBG(Electromagnetic Band Gap) structure, a micro-strip array antenna is designed. Its periodical structure and band-gap characteristics could effectively reduce the coupling of between antenna arrays. Due to adoption of single EBG or single periodical EBG, the Antenna isolation ratio is guitely sensitive to the patch size and the distance of between the elements. Based on this feature, the effects of EBG structure on array antenna respectively in unit five and six are compared, finally reaching the optimal solution. In addition, the EBG structure in units six, cycles three is applied to remarkably improving the isolation degree of between the antennas, thus laying certain foundations for future researchof compact antenna. For this reason,this paper is of certain reference value.%设计了一款基于电磁带隙结构EBG（Electromagnetic Band Gap）的微带阵列天线，利用其周期性结构和禁带的特性，有效降低天线阵列单元间的互耦。

应用于感知物联网的方向图可重构超高频RFID标签天线

应用于感知物联网的方向图可重构超高频RFID标签天线徐守辉;邱方;王子旭【摘要】A hinge⁃type ultrahigh frequency RFID(radio frequency identification)antenna with reconfigurable radiation pat⁃tern is proposed for cognitive Internet of Things(CIOT). The antenna is a dipole antenna applied to metal object. It was ana⁃lyzed with FEM(finite element method). The hinge⁃type structure enables the dipole antenna to be rotatable andby rotating the angle between the two antenna arms from 0 degree to 90 degree. Its radiation pattern can be reconfigured form 40° to -90° (assuming the axis vertical to ground is 0°). Moreover,EBG(electromagnetic band⁃gap)structure is applied to realize further reconfigurable radiation patters. Specifically,taking 60° as an example,by tuning the size of rectangular hole of EBG,its radia⁃tion patterns could be tuned ±30°. The reconfigurable radiation pattern is qualitatively verified in an experimentby performance measurement system.%提出了一款新颖的应用于感知物联网的方向图可重构的RFID超高频电子标签天线。

单细胞测序筛选原发和复发胶质母细胞瘤中恒定存在的胶质瘤细胞

单细胞测序筛选原发和复发胶质母细胞瘤中恒定存在的胶质瘤细胞范琴;王俊杰;孙鹏【期刊名称】《临床医学进展》【年(卷),期】2024(14)5【摘要】目的:胶质母细胞瘤是成人最常见、侵袭性最强的原发性高致死性脑肿瘤。

基因表达似乎是预测胶质母细胞瘤患者生存和治疗反应的重要因素。

然而,不同患者之间肿瘤细胞的异质性可能会阻碍找到一种癌症的恒定治疗靶点,因为不仅ndGBM和rGBM之间的遗传谱不同,不同患者之间甚至同一患者的肿瘤细胞亚群之间的遗传谱也不同。

因此,在不同的ndGBM和rGBM患者中发现一种恒定的肿瘤细胞类型可能是更好地治疗胶质母细胞瘤的一种潜在方法。

本文通过单细胞测序(single-cell RNA sequencing, scRNA-seq)筛选恒定的肿瘤细胞类型,比较新诊断的胶质母细胞瘤(ndGBM)和复发性胶质母细胞瘤(rGBM)组的遗传机制。

方法:从Gene Expression Omnibus (GEO)数据库(GSE182109)下载3例ndGBM患者(3例样本)和2例rGBM患者(7例样本)胶质母细胞瘤样本的基因表达谱,并使用R包进行scRNA-seq。

结果:共有59,400个胶母细胞瘤组织细胞及12,172个胶质瘤细胞最终通过质控,本研究成功筛选出了恒定的胶质瘤细胞亚型。

此外,本研究还进行了拟时序分析、基因本体(Gene Ontology, GO)和京都基因与基因组百科全书(Kyoto Encyclopedia of Genes and Genomes, KEGG)富集分析、细胞–细胞相互作用和单细胞调控网络推测聚类。

本研究还鉴定了ndGBM和rGBM组的关键基因基因和各组的不同功能。

结论:本文筛选了WHO IV级、IDH野生型和甲基化的胶质母细胞瘤中存在于每个样本和每个发展阶段的一类肿瘤细胞亚型。

进一步的研究需要通过分子实验来验证分子机制,并针对这些靶点开发诊断方法和药物治疗ndGBM和rGBM。

EB病毒核抗原1羧基端的原核表达、纯化及其免疫学特性

EB病毒核抗原1羧基端的原核表达、纯化及其免疫学特性王卫东;齐建国;谷淑燕;桑建利【期刊名称】《中国生物化学与分子生物学报》【年(卷),期】2004(20)6【摘要】为进一步研究EB病毒核抗原 1(EBNA1)的功能及提高EB病毒 (EBV)相关疾病辅助诊断的特异性 ,对EBNA1基因3′端的部分片段进行了原核表达、纯化并初步研究其免疫学特性 .采用PCR法扩增了EBNA1基因编码区 ,经酶切鉴定、序列分析后 ,将其3′端 5 73bp片段克隆至原核表达载体pET30a中 ,得到重组质粒pET30a SS5 80 .该重组质粒转化大肠杆菌BL2 1(DE3)感受态细胞并经异丙基β D 硫代半乳糖苷 (IPTG)诱导表达出分子量约 2 5kD的融合蛋白 (2 5 kDEBNA1) .该蛋白以包涵体和可溶形式存在 ,均可用Ni2 + 离子亲和柱纯化 .Western印迹结果显示 ,该蛋白能与鼻咽癌 (NPC)病人血清发生特异性反应 .纯化的 2 5kDEBNA1蛋白免疫BALB c小鼠后 ,经ELISA检测获得了高效价的多克隆抗体 .免疫印迹和间接免疫荧光结果显示制备的免疫小鼠血清能够与HeLa细胞中瞬时表达的EBNA1蛋白发生特异性反应 ,且特异性优于鼻咽癌病人血清 .以上结果表明成功构建了EBNA1羧基端的原核表达质粒 ,并在大肠杆菌中高效表达了 2 5kDEBNA1蛋白。

【总页数】7页(P798-804)【关键词】EB病毒核抗原1;原核表达;纯化;抗原性;免疫原性【作者】王卫东;齐建国;谷淑燕;桑建利【作者单位】北京师范大学生命科学学院;中国疾病预防控制中心病毒病预防控制所,北京100052;中国疾病预防控制中心病毒病预防控制所【正文语种】中文【中图分类】Q789【相关文献】1.EB病毒潜伏膜蛋白2多表位的原核表达及其抗原特性分析 [J], 陆丽君;李玲玲;刘建晓;王佳;朱珊丽;陈筱菲;张丽芳2.EB病毒核抗原1羧基端原核表达产物的纯化方法 [J], 王卫东;齐建国;谷淑燕3.EB病毒诱发淋巴瘤中EB病毒核抗原－1的表达 [J], 路素丽;何洁;唐运莲;甘润良4.猪圆环病毒2型Cap羧基端多肽原核表达及抗原性分析 [J], 俞莉俐;姜平;冯志新;5.EB病毒LMP1蛋白CTAR23结构域的原核表达及纯化 [J], 张雪雁;杨英;马桂璋因版权原因，仅展示原文概要，查看原文内容请购买。