Introduction Radio Ad Filtering with Machine Learning 1

合集下载

radio propagation

radio propagation

Outline
• Introduction and some terminology • Propagation Mechanisms • Propagation models
– Large scale propagation models – Small scale propagation (fading) models
• Diffraction occurs when waves hit the edge of an obstacle
– “Secondary” waves propagated into the shadowed region – Excess path length results in T a phase shift – Fresnel zones relate phase shifts 1st Fresnel zone to the positions of obstacles
– Conductors & Dielectric materials (refraction)
• Diffraction
– Fresnel zones
• Scattering
– “Clutter” is small relative to wavelength
17 March 1999 Radio Propagation 8
• Nearby metal objects (street signs, etc.)

• Large distant objects
– Analytical model: Radar Cross Section (RCS)
17 March 1999 Radio Propagation 15

亚赫英氏SQ-6混音控制台指南说明书

亚赫英氏SQ-6混音控制台指南说明书

IntroductionSafetyBefore powering on the SQ, read the safety instructions sheet (AP9240/CL1-1) that is supplied along with this guide. For your own safety and that of the operator, technical crew and performers, follow all instructions and heed all warnings included in these documents and printed directly on the equipment. RegistrationTo be kept informed of updates, the latest firmware and new releases for the SQ range, register your SQ-6 at /registerFirmware and Reference GuideThis introduction is intended to give you an overview of the SQ-6 hardware and outline operating principles. Visit to obtain the latest version of firmware and reference guide. The latest firmware is required if you intend to use any SQ Apps with your SQ.VentilationThe SQ uses fans for cooling. Adequate space must be left for air flow around fans and vents when in use.FeaturesThe SQ is a high resolution 96kHz audio mixing console. It has been designed using the latest technology to provide the most detailed and accurate sound quality, along with a range of options for expandability and integration.AP11349 Issue 2AccessoriesSQ-BRACKET Detachable Metal Bracket for iPad/tabletAP11333 Water repellent polyester dustcover with printed logoAR84 8 XLR input, 4 XLR output, dSnake Remote AudioRack (Rackmount) AR2412 24 XLR input, 12 XLR output dSnake Remote AudioRack (Rackmount)AB168 16 XLR Input, 8 XLR Output, dSnake Remote AudioRack (StageBox/Rackmount) DX168 16 XLR Input, 8 XLR Output, 96kHz DX Remote AudioRack (StageBox/Rackmount) DX164-W 16 XLR Input, 4 XLR Output, 96kHz DX Wall Mount Audio Expander DX-HUB Remote Audio Hub with 4 DX Link ports (Rackmount kit available) AH9650 100m drum of EtherFlex Cat5e with locking Neutrik EtherCon connectors AH9981 50m drum of EtherFlex Cat5e with locking Neutrik EtherCon connectors AH965120m of Neutrik EtherFlex Cat5e with locking Neutrik EtherCon connectorsSLink Port Compatibility Sample Rate Protocol Max LengthDX168, DX164-W, DX Hub 96kHz DX 100m Cat5e or higher AR2412, AR84, AB168 48kHz dSnake 120mCat5e or higher ME-U, ME-1, ME-50048kHzdSnakeCat5e or higherSQ Range48 input channels with preamp, HPF, PEQ, gate, comp, delay 32 output channels (LR, 12 mono/stereo Mix, 3 Stereo Matrix) 8 stereo FX with dedicated return channels 8 Mute groups, 8 DCA groupsSource patching (Local, SLink remote, Option card, USB) Output socket and Insert I/O patchingMulti-channel USB streaming and direct to USB drive recording Talkback mic input, dual footswitch control, wireless controlSQ-6 Specific144 fader strips (24+1 faders, 6 layers) 24 local mic/line input sockets 3 local stereo line input sockets 14 XLR + 2 TRS output sockets 16 assignable SoftKeys4 assignable Soft Rotaries with LCD DisplaysLocal Mic/Line Inputs Local Stereo Line Inputs Talkback Mic Input Local XLR OutputsLocal TRS Jack OutputsAES Digital OutputMono/Dual Footswitch Connection Mains Power Input and Switch I/O Port - Option CardMulti-format multi-channel digital audioUSB-B PortConnection to a computer for multi-channel audio and MIDI I/O Network Port Connect to a router for network/wireless controlSLink PortFor connection to Allen&Heath remote audio racks, including AB, AR and DX ranges, as well as the ME personal monitoring systemTouch Screen, Screen Select Keys and Screen EncoderView processing and access the routing and setup menus using keys below. Touch to select a parameter and use the rotary to adjust values.Fader Strips and Layer Select Keys6 layers of 24 faders provide 144 assignable strips for access to any combination of channels, returns,masters and DCAs. Each strip has fader, mute, select and PAFL keys, peak and signal meter.Ident StripLCD displays show channel name and colour for each of the 24 strips. Press the‘View’ key to see secondary information such as input source.Channel(Pre/HPF/Gate/Comp)Physical controls for the selected channel. Preamp, HPF frequency, Gate threshold, Comp threshold.Channel (PEQ/GEQ)Physical controls for the selected channel. EQ band select keys and parametric controls. Use the ‘Fader Flip’ key to present selected mix GEQ on faders. Pan ControlMaster Strip and Mix Select KeysPress a blue ‘Mix’ key to present its sends on the 24 faders and its master on the master fader strip. Select ‘LR’ to work with the main LR mix and channel faders.FX Send Select KeysPress a blue ‘FX’ key to present its sends on the 24 faders and its master send on the master fader strip. Headphone Output and Level Control Main MeterDisplays the LR Mix or selected PAFL signal level.Talk KeyMomentary or latching switch for the talkback microphone.SQ-Drive PortRecord/play audio direct to/from a USB drive. Transfer scene, show and library data using a USB key. Update SQ firmware.ST3 Input3.5mm stereo jack input, can be used for connection to an external background music device.Pre Fade and Assign KeysHold ‘Pre-Fade’ and press ‘Sel’ to toggle channels pre or post fade to the mix. Hold ‘Assign’ and press ‘Sel’ to route channels to the selected mix.CH to All Mix KeyPress and hold to present all sends to mixes for the currently selected channel. The ident strip displays mix names. Copy/Paste/Reset KeysUsed to copy, paste or reset processing blocks or channel parameters.Library KeyOpens different libraries to enable save and recall of presets for channel/mix/FX processing.Assignable SoftKeysUse Setup screen to assign functions such as mutes, tap tempo, scene recall, SQ-Drive control and more.Assignable EncodersUse Setup screen to assign functions for quick access to often used parameters.i. Power off any connected amplifiers or powered speakers. ii. Navigate to the ‘Home’ screen and select ‘Shut Down’ iii.Switch off the unit using the push switch (27).Press a blue ‘LR’, ‘Mix’ or ‘FX’ Key to present send levels for the selected Mix on the 24 Fader Strips. Use the Layer Keys (2) to move through the 6 layers of faders and adjust individual levels. The Master strip (7) controls the master send level of the selected Mix/FX.Select a strip by pressing the green ‘Sel’ Key on a Fader Strip (2) or the Master Strip (7).The physical controls (4), (5) and (6) can now be used to adjust parameters for the selected strip.Go to the ‘Processing’ screen to see an overview of the processing for the selected strip.Tap on any part of the processing to see a detailed view, then touch a parameter on-screen and use the touch screen encoder (1) to adjust.Mute Keys are illuminated when a strip is muted.By default, PAFL (Pre/After Fade Listen) Keys allow you to route one channel at a time to the PAFL bus/Phones output. PAFL settings can be changed in the ‘Setup’ screen.Mix sends set to ‘Post Fade’ follow the LR send levels. To toggle channels between ‘Pre Fade’ and ‘Post Fade’ for the selected Mix, hold the ‘Pre Fade’ Key and use ‘Sel’ Keys.To assign or un-assign a strip from the currently selected mix, hold the ‘Assign’ Key and use ‘Sel’ Keys.Pressing and holding the ‘CH to All Mix’ Key will display the send levels for the currently selected strip across the main fader strips.Press the ‘FX’ Key to see and adjust FX engines.Use the ‘Library’ Key (17) to recall FX types and presets - change parameters by selecting on-screen and using the touch screen encoder.FX busses 1 to 4 (8) send to FX engines 1 to 4 by default.FX Return channels can be routed to Mixes in the same way as stereo input channels.Hold the ‘Copy’ Key and press an ‘In’ Key (4) (5), a ‘Sel’ Key (2) (7), to copy parameters.Hold the ‘Paste’ Key and press a ‘Sel’ Key (2) (7) to paste the copied processing to another channel. Hold the ‘Reset’ Key and press an ‘In’ Key (4) (5), a ‘Sel’ Key (2) (7), or on-screen to reset parameters.A ‘Scene’ is used to store or recall a mix. A ‘Show’ comprises multiple scenes and all settings. Press the ‘Scenes’ Key to access the list of scenes in the current show.Use a combination of scene filters and ‘Safes’ to decide which settings/parameters/strips are affected when a scene is recalled.i. Connect power lead (27).ii. Connect input sources using (20), (21) and (22).iii. Connect outputs (23) and (24) to amplifiers, speakers or line level inputs on other equipment. iv. If required, connect digital I/O such as AudioRacks or Computers using (25), (28), (29) and (31). v. If you are using a footswitch, connect this (26). vi. Switch on the SQ using the push switch (27).vii.Power on any connected amplifiers or powered speakers.To reset all mix, parameter and routing settings go to the ‘Scenes’ screen (1), then press and hold the ‘Reset Mix Settings’ button. This will ‘zero’ the desk without deleting saved scenes or libraries.To check or alter patching, go to the ‘I/O’ screen (1) and use the matrix to patch from Local/Digital Inputs to SQ input channels, and to patch SQ outputs [LR/Mix/Group/Matrix/DirectOut] to Local/Digital Outputs.Balanced mono/stereo inputs Mic or line level XLR 1=Gnd, 2=+, 3= -ST1 and ST2 Inputs Line level ¼” TRS Jack Tip= +, Ring= -, Sleeve=GndST3 Input Line level 3.5mm Jack Tip=Left, Ring=Right, Sleeve=Gnd Balanced XLR Outputs Line level XLR 1=Gnd, 2= +, 3= -Balanced Jack Outputs Line level ¼” TRS Jack Tip= +, Ring= -, Sleeve=GndSLink RJ45/EtherCON. Use Cat5e or higher. Refer to individual expansion unit instructions.AES Stereo Digital Output Digital XLR Use 110Ω AES CableRear USB Connection USB-B, Conforms to USB 2.0 standardNetwork Connection RJ45, Use Cat5e or higherFootswitch ¼” TRS (dual) or TS (mono) JackThere are many support resources available through our website including user guides, knowledgebase articles and access to the Allen & Heath Digital Community.For local language support, please contact the Allen & Heath distributor for your region.Limited One Year Manufacturer’s WarrantyAllen & Heath warrants the Allen & Heath -branded hardware product and accessories contained in the original packaging ("Allen & Heath Product”) against defects in materials and workmanship when used in accordance with Allen & Heath's user manuals, technical specifications and other Allen & Heath product published guidelines for a period of ONE (1) YEAR from the date of original purchase by the end-user purchaser ("Warranty Period").This warranty does not apply to any non-Allen & Heath branded hardware products or any software, even if packaged or sold with Allen & Heath hardware.Please refer to the licensing agreement accompanying the software for details of your rights with respect to the use of software/firmware (“EULA”).Details of the EULA, warranty policy and other useful information can be found on the Allen & Heath website: /legal.Repair or replacement under the terms of the warranty does not provide right to extension or renewal of the warranty period. Repair or direct replacement of the product under the terms of this warranty may be fulfilled with functionally equivalent service exchange units.This warranty is not transferable. This warranty will be the purchaser’s sole and exclusive remedy and neither Allen & Heath nor its approved service centres shall be liable for any incidental or consequential damages or breach of any express or implied warranty of this product.Conditions of WarrantyThe equipment has not been subject to misuse either intended or accidental, neglect, or alteration other than as described in the User Guide or Service Manual, or approved by Allen & Heath. The warranty does not cover fader wear and tear.Any necessary adjustment, alteration or repair has been carried out by an authorised Allen & Heath distributor or agent. The defective unit is to be returned carriage prepaid to the place of purchase, an authorised Allen & Heath distributor or agent with proof of purchase. Please discuss this with the distributor or the agent before shipping. Units returned should be packed in the original carton to avoid transit damage.DISCLAIMER: Allen & Heath shall not be liable for the loss of any saved/stored data in products that are either repaired or replaced.Check with your Allen & Heath distributor or agent for any additional warranty information which may apply. If further assistance is required please contact Allen & Heath Ltd.Any changes or modifications to the equipment not approved by Allen & Heath could void the compliance of the product and therefore the user’s authority to operate it.。

cognitiveRadio[1]答辩PPT

cognitiveRadio[1]答辩PPT

5
Cognitive Radio Overview
Software Radio
Programmable Wideband Wideband RF Processor(s) A/D-D/A* Conversion
HF LVHF VHF-UHF Cellular
PCS Indoor & RFLAN VHDRBaseband Modem
Back End Control
Equalizer Algorithm
Software
Antenna
Hardware
RF
Modem
INFOSEC
Baseband
User Interface
Secure Downloads, Pro-Active Radio Resource Management
contains RKRL j
Micro-world j := {<Frame>}*
<Frame>
<Handle><Model><Body><Context><Resource-specification>
Syntax
<Root><Source><Time><Place> <Resources><Depth><Breadth><Sub-Elements><Sub-Frames>
RF IF
Bits
Bits
Aux
Aux or
Aux
CT
Fill
PT
BB
Air
I/O

QPSK

QPSK

WIRELESS, RF, AND CABLE Application Note 686: Oct 13, 2000QPSK Modulation DemystifiedReaders are presented with step by step derivations showing the operation of QPSK modulation and demodulation. The move from analog communication to digital has advanced the use of QPSK. Euler's relation is used to assist analysis of multiplication of sine and cosine signals. A SPICE simulation is used to illustrate QPSK modulationof a 1MHz sine wave. A phasor diagram shows the impact of poor synchronizationwith the local oscillator. Digital processing is used to remove phase and frequency errors.Since the early days of electronics, as advances in technology were taking place, the boundaries of both local and global communication began eroding, resulting in a world that is smaller and hence more easily accessible for the sharing of knowledge and information. The pioneeringwork by Bell and Marconi formed the cornerstone of the information age that exists today and paved the way for the future of telecommunications.Traditionally, local communication was done over wires, as this presented a cost-effective wayof ensuring a reliable transfer of information. For long-distance communications, transmissionof information over radio waves was needed. Although this was convenient from a hardware standpoint, radio-waves transmission raised doubts over the corruption of the information and was often dependent on high-power transmitters to overcome weather conditions, large buildings, and interference from other sources of electromagnetics.The various modulation techniques offered different solutions in terms of cost-effectivenessand quality of received signals but until recently were still largely analog. Frequency modulation and phase modulation presented a certain immunity to noise, whereas amplitude modulation was simpler to demodulate. However, more recently with the advent of low-cost microcontrollers and the introduction of domestic mobile telephones and satellite communications, digital modulation has gained in popularity. With digital modulation techniques come all the advantages that traditional microprocessor circuits have over their analog counterparts. Any shortfalls in the communications link can be eradicated using software. Information can now be encrypted, error correction can ensure more confidence in received data, and the use of DSP can reduce the limited bandwidth allocated to each service. As with traditional analog systems, digital modulation can use amplitude, frequency, or phase modulation with different advantages. As frequency and phase modulation techniques offer more immunity to noise, they are the preferred scheme for the majority of services in use today and will be discussed in detail below.Digital Frequency ModulationA simple variation from traditional analog frequency modulation (FM) can be implemented by applying a digital signal to the modulation input. Thus, the output takes the form of a sine wave at two distinct frequencies. To demodulate this waveform, it is a simple matter of passing the signal through two filters and translating the resultant back into logic levels. Traditionally, this form of modulation has been called frequency-shift keying (FSK).Digital Phase ModulationSpectrally, digital phase modulation, or phase-shift keying (PSK), is very similar to frequency modulation. It involves changing the phase of the transmitted waveform instead of the frequency, these finite phase changes representing digital data. In its simplest form, a phase-modulated waveform can be generated by using the digital data to switch between two signals of equal frequency but opposing phase. If the resultant waveform is multiplied by a sine wave of equal frequency, two components are generated: one cosine waveform of double the received frequency and one frequency-independent term whose amplitude is proportional to the cosine of the phase shift. Thus, filtering out the higher-frequency term yields the original modulating data prior to transmission.This is difficult to picture conceptually, but mathematical proof will be shown later.Quadraphase-Shift ModulationTaking the above concept of PSK a stage further, it can be assumed that the number of phase shifts is not limited to only two states. The transmitted "carrier" can undergo any number of phase changes and, by multiplying the received signal by a sine wave of equal frequency, will demodulate the phase shifts into frequency-independent voltage levels.This is indeed the case in quadraphase-shift keying (QPSK). With QPSK, the carrier undergoes four changes in phase (four symbols) and can thus represent 2 binary bits of data per symbol. Although this may seem insignificant initially, a modulation scheme has now been supposed that enables a carrier to transmit 2 bits of information instead of 1, thus effectively doubling the bandwidth of the carrier.The proof of how phase modulation, and hence QPSK, is demodulated is shown below.The proof begins by defining Euler's relations, from which all the trigonometric identities can be derived.Euler's relations state the following:which gives an output frequencyby any phase-shifted sine wave.To prove this,Thus, the above proves the supposition that the phase shift on a carrier can be demodulated into a varying output voltage by multiplying the carrier with a sine-wave local oscillator and filtering out the high-frequency term. Unfortunately, the phase shift is limited to two quadrants;a phase shift of /2. Therefore, to accurately decode phase shifts present in all four quadrants, the input signal needs to be multiplied by bothsinusoidal and cosinusoidal waveforms, the high frequency filtered out, and the data reconstructed. The proof of this, expanding on the above mathematics, is shown below. Thus,A SPICE simulation verifies the above theory.Figure 1A shows a block diagram of a simple demodulator circuit. The input voltage, QPSK IN, is a 1MHz sine wave whose phase is shifted by 45°, 135°, 225°, and then 315° every 5µs.Figures 2 and 3 show the "in-phase" waveform, Vi, and the "quadrature" waveform, V q, respectively. Both have a frequency of 2MHz with a dc offset proportional to the phase shift, confirming the above mathematics.Figure 1B is the phasor diagram showing the phase shift of QPSK IN and the demodulated data.The above theory is perfectly acceptable, and it would appear that removing the data from the carrier is a simple process of low-pass filtering the output of the mixer and reconstructing the 4 voltages back into logic levels. In practice, getting a receiver local oscillator exactly synchronized with the incoming signal is not easy. If the local oscillator varies in phase with the incoming signal, the signals on the phasor diagram will undergo a phase rotation, its magnitude equal to the phase difference. Moreover, if the phase and frequency of the local oscillator are not fixed with respect to the incoming signal, there will be a continuing rotation on the phasor diagram.Therefore, the output of the front-end demodulator is normally fed into an ADC and any rotation resulting from errors in the phase or frequency of the local oscillator are removed in DSP.With the advances in monolithic silicon germanium (SiGe) technology, all of the above front-end circuitry can be integrated to reduce the problems outlined. A good example of how much of the front-end circuitry can be integrated is illustrated in the MAX2450, ultra-low-power quadrature modulator/demodulator IC. This is one of many devices from Maxim Integrated Products that incorporates the quadraphase shifter, the on-chip oscillator, and the mixer. Once the data has been demodulated, the output can be applied to a high-frequency dual-channel ADC (such as the MAX1002 or the MAX1003) before processing the signal in DSP.As the MAX2450 is designed to be used at an IF of 35MHz to 80MHz, RF signals up to2.5GHz can be downconverted using the MAX2411A. This is a high-frequencyup/downconverter with a low-noise amplifier (LNA) local oscillator, and it has access to the output of the LNA for image-reject filtering.Alternatively, an effective way of converting straight to baseband is using a direct-conversion tuner IC. The MAX2102 is designed to take RF inputs from 2150MHz and convert directly down to baseband I and Q signals, thus providing cost savings over multiple-stage devices. The above devices are part of the rapidly expanding RF chipsets from Maxim Integrated Products. With five high-speed processes, more than 70 high-frequency standard products, and 52 ASICs in development, Maxim is committed to being a major player in the RF/wireless, fiber/cable, and instrumentation markets.MORE INFORMATIONMAX1002:QuickView -- Full (PDF) Data Sheet (120k)-- Free Sample MAX1003:QuickView -- Full (PDF) Data Sheet (128k)-- Free Sample MAX2102:QuickView -- Full (PDF) Data Sheet (160k)-- Free Sample MAX2361:QuickView -- Full (PDF) Data Sheet (40k)-- Free Sample MAX2411A:QuickView -- Full (PDF) Data Sheet (144k)-- Free Sample MAX2450:QuickView -- Full (PDF) Data Sheet (120k)-- Free Sample。

600_electrical_engineering_books

600_electrical_engineering_books

這600本書幾乎包括了電氣工程專業的所有內容。

例如:電子學最基礎的《Circuit.Analysis.Theory.And.Practice.》(電路分析)、哈佛大學的經典教材《The.Art.of.Electronics》(電子學的藝術)、DSP.Facts.and.Equipment。

詳細書籍名:Wireless.Securit.PrivacyBest.Practices.and.Design.Techniques.Artech-Interference.Analysis.and.Reduction.for.Wireless.Systems.munications.works.munications.Network.Design._20-_20.Wiley._.Sons.802.11.Security.N.Fundamentals.Cisco.Press.eBookwork.Site.Surveying.and.Installation.Cisco.Press.Nov.2004.eBookA.First.Course.in.Corporate.Finance.b.in.Circuits.and.Electronics.munication.er_27s.Guide.to.Aspect.Ratio.Conversion.A.wavelet.tour.of.signal.processing.Mallat.S..draft_.2005.MNw.ponent.Modeling.Morgan.Kaufmann.eBook.-.LiB. Abstract.Harmonic.Analysis.of.Continuous.Wavelet.Transforms.Adaptive.Digital.Filters.Second.Edition.putational.Intelligence.Perspective.Adaptive_20Control_20Systems.Addison.Wesley._20-_20.RTP..Audio.and.Video.for.the.Internet.Advanced.Digital.Signal.Processing.and.Noise.Reduction.2nd.Edition.Advanced.Techniques.in.RF.Power.Amplifier.Design.works.Springer.eBook.Advanced_20Control_20Engineering.Advances.in.Fingerprint.Technology.Second.Edition.eBookworks.Artech.House.Publishers.Jun.2005.eBook. Aerials..Air.and.Spaceborne.Radar.Systems.An.Introduction.2001.WilliamAndrewPublishing.RR. munication.Systems.And.Their.Applications.Alternative.Breast.Imaging.Kluwer.Academic.Publishers.eBook.An.Introduction.To.Statistical.Signal.Processing.An.Introduction.to.Digital.Audio.An.Introduction.to.Pattern.Recognition.An_20Introduction_20to_20the_20Theory_20of_20Microwave_20Circuits_20_Kurokawa_. Analog.BiCMOS.Design.Practices.and.Pitfalls.Analog.Circuit.Design.Analog.Circuits.Cookbook.Analog.Integrated.Circuit.Design.Analog.and.Digital.Circuits.for.Electronic.Control.System.Applications..Analog_20And_20Digital_20Control_20System_20Design.Analysis.And.Design.Of.Analog.Integrated.Circuits.Analysis_20and_20Design_20of_20Integrated_20Circuit-Antenna_20Modules.Antenna_20Arraying_20Techniques_20In_20The_20Deep_20Space_20Network.Antenna_20handbook.rmation.Super.Skyways.Institute.of.Physics.Publishing.Feb.2004.eBook-DDU. Application.-.Specific.Integrated.Circuits.-.Addison.Wesley.Michael.John.Sebastian.Smith. munications.2002.Art.And.Business.Of.Speech.Recognition.Addison.Wesley.eBook.yout.Artech..Radio.Frequency.Integrated.Circuit.Design.Artech.House.GPRS.for.Mobile.Internet.rmation.theory.Asynchronous.Circuit.Design..Audel.Electrical.Course.for.Apprentices.and.Journeymen.eBook.Automated.Fingerprint.Identification.Systems..AFIS..Academic.Press.eBookAutomotive_20Computer_20Controlled_20Systems_20Diagnostic_20Tools_20And_20Techniques. Bandwidth.efficient.digital.modulation.in.deep.munications.ponents._.Hardware.-.I.CFS.ponents._.Hardware.-.II.CFS.Basic.Theory.and.Application.of.Transistors.Bebop.to.the.Boolean.Boogie.Bluetooth.Application.Developers.Guide.Bluetooth.Demystified.Bluetooth.Security.2004.BluetoothGuide.Broadband.Bible.John.Wiley.and.Sons.eBook.Broadband.Bringing.Home.the.Bits.Broadband.Microwave.Amplifiers.Artech.House.eBook-TLFeBOOK.Building.Financial.Models.McGraw-Hill.2004.works.with.802.11.eBook.C.Algorithms.for.Real._20-_20.time.DSP.1995.CAD_20of_20Microstrip_20Antennas_20for_20Wireless_20Applications.CDMA.Capacity.and.Quality.Optimization.CDMA.Mobile.Radio.Design.Artech.House.CDMA.RF.System.Engineering.CDMA.Systems.Capacity.Engineering.Artech.House.Publishers.eBook._20-_20.kB.CMOS.Analog.Circuit.Design.CMOS.Electronics.How.It.Works.How.It.Fails.yout.CMOS.Integrated.ADC.and.DAC.2ndEd..CMOS.PLL.Synthesizers.Analysis.and.Design.Springer.Nov.2004.eBook.-.LinG.CMOS.memory.circuits.CRC.Press.munications.Facility.Design.Handbook.CRC_20Press_20-_20Intelligent_20Control_20Systems_20Using_20Soft_20Computing_20Metho dologies.Cellular.Mobile.Radio.Systems.Designing.Systems.For.Capacity.Optimization.Circuit.-.techniques-for-low-voltage-high-speed-ADCs.Circuit.Analysis.Theory.And.Practice.Circuit.Design.for.RF.Transceivers.munications.Circuits.for.the.Hobbyist.Closed.Circuit.Television.Closing.The.Gap.Between.ASIC.and.Custom.Tools.And.Techniques.of.High.Performance.ASIC.Desig n.work.Test.and.Measurement.Handbook.works._20-_20.Fundamental.Concepts.-.McGraw.Hill.-.Leon-Garcia_.Widjaja. Communications.Receivers.DSP_.Software.Radios_.and.Design_.Third.Edition.Compact_20and_20Broadband_20Microstrip_20Antennas.Complete.Wireless.Design.Computer.Explorations.in.Signals.and.Systems.Computer.imaging.recipes.in.C.Myler.H.R._.Weeks.A.R..PH_.1993pi.T.munication.Consumer_27s.Guide.to.Cell.Phones.and.Wireless.Service.Plans.Continuous.-.Time.Active.Filter.Design.Control_20EngineeringGuide_20For_20Beginners.Coplanar_20Waveguide_20Circuits__20Components__20and_20Systems.Crane.R..Simplified.approach.to.image.processing.in.C.PH_.1997.T.ISBN.0132264161.DOE.Fundamentals.Handbook_.Electrical.Science.vol.1.DOE.Fundamentals.Handbook_.Electrical.Science.vol.2.DOE.Fundamentals.Handbook_.Electrical.Science.vol.3.DOE.Fundamentals.Handbook_.Electrical.Science.vol.4.DSP.Facts.and.Equipment.DSP.Realtime.Operating.Systems.for.Embedded.Systems.DSP.for.In.Vehicle.and.Mobile.Systems.Springer.eBook-YYePG.working.Devices._20-_20.Fourth.Edition.Data.Conversion.Handbook.Elsevier.eBook.-.LinG.Deep.Submicron.CMOS.Circuit.Design.Simulator.In.Hands.Delmar.Digital.Signal.Processing._20-_20.-Filtering.Approach.Delmar.Fiber.Optics.Technician_27s.Manual.2nd.Ed..Design.Of.Linear.RF.Outphasing.Power.Amplifiers.Artech.House.eBookNs.Springer.Sep.2005. Design.of.Analog.CMOS.Integrated.Circuits.Design_20of_20RF_20And_20Microwave_20Amplifiers_20And_20Oscillators..Designing.Analog.Chips.work.works.Developments.in.Speech.Synthesis.John.Wiley.Sons.Apr.2005.eBook._20-_20.LinG. Dictionary.of.Video.Television.Technology.Dielectric_20Resonator_20Antennas.Digital.Audio.Broadcasting.munication.Over.Fading.Channels.munications.Design.for.the.Real.World.Digital.Design.Fundamentals.Digital.Design.Principles.and.Practices.Digital.Electronics.Digital.Frequency.Synthesis.Demystified.Digital.Integrated.Circuits.wo02_8.munication.Digital.Logic.And.Microprocessor.Design.With.VHDL.Digital.Signal.Processing.Handbook.VK.Madisetti_DB.Williams_CRC.ing.C.bVIEW.Newnes.Jun.2005.eBook._20-_20.D DU.munications.Ieee.Digital.Switching.Systems.System.Reliability.and.Analysis.Digital.Synthesizers.and.Transmitters.for.Software.Radio.Springer.Jul.2005.eBook._20-_20.DDU. Digital.Systems.Engineering..Digital.Video.Quality.Vision.Models.and.Metrics.John.Wiley.and.Sons.Mar.2005.eBook._20-_20.D DU.Digital.Video.for.Dummies.Wiley..2003._.3Ed.Digital.image.processing._20-_20.B.Jahne.Digital.signal.Processing.Digitally.Assisted.Pipeline.ADCs.Theory.and.Implementation.Discovering.Bluetooth.Sybex.Discrete.Time.Signal.Processing._20-_20.Oppenheim.Distortion.Analysis.of.Analog.Integrated.Circuits.Distortion.in.rf.power.amplifiers.ebook._20-_20.lib.Duda.R.O._.Hart.P.E._.Stork.D.G..Pattern.classification.02ed._.Wiley.C.738s.EDGE.for.Mobile.Internet.ESD.In.Silicon.Integrated.Circuits.Electrical.Circuits.plante_CRC.Electrical._.Electronic.Principles._.Technology.-.0750665505.Newnes.John.Bird.Electrician_27s.Exam.Question.and.Answers.Electromagnetic_20Waves_20and_20Antennas.Electronics.for.Dummies.John.Wiley.and.Sons.eBook.-.LinG.Electronics.for.Hobbyists.1.Electronics.for.Hobbyists.2.Electronics.for.Hobbyists.3.Electronics.for.Hobbyists.4.Electronics.for.Hobbyists.5.Electronics.for.Hobbyists.6.Electronics.for.Hobbyists.7.work.Technologies.Springer.Sep.2004.eBook._20-_20.LinG. working.Engineer_27s.Mini.-._5bNotebook.-.555_5d.-.Timer.IC.Circuits.Engineer_27s.Notebook.II.A.Handbook.Of.Integrated.Circuit.Applications.-.Forrest.Mims. Engineering.Digital.Design.rmation.Theory.Error.control.coding..From.theory.to.practice.Sweeney.P..Wiley_2002.Essentials.of.Managing.Corporate.Cash.-.John.Wiley.Sons.Experimental.Approach.CDMA._.Interference.From.Architecture.Through.VLSI.Fast.Forward.MBA.in.Finance.Feedback.Amplifiers.Theory.and.Design.Feedback.Circuit.Analysis.Feedback.Linearization.of.RF.Power.Amplifiers.Feedbackcontroltheory.munication.Systems.Fiber.Optic.Sensors.Fiber.to.the.Home.The.New.Empowerment.Wiley.Interscience.Oct.2005.eBook._20-_20.LinG. Fibre.Channel.for.Mass.Storage._20-_20.Prentice.Hall.Fibre.Channel.for.SANs.Filter.Handbook.a.Practical.Design.Guide.-.S..Niewiadomski.Finance.for.Non.-.Financial.Managers.Financial.Engineering.Principles.A.Unified.Theory.Financial.Risk.Manager.Handbook.Wiley.Second.Edition.Financial.modeling.with.jump.processes.Finite_20Antenna_20Arrays_20and_20FSS.First.course.on.wavelets.Hernandez_.Weiss..CRC_.1996.T.ISBN.0849382742.Fixed.Broadband.Wireless.System.Design._20-_xxuss.For.Dummies.HDTV.For.Dummies.Nov.2004.eBook._20-_20.DDU.Fundamental_20Limitations_20In_20Filtering_20And_20Control.Fundamentals.Of.Electric.Circuits..Fundamentals.Of.RF.Circuit.Design.With.Low.Noise.Oscillators.munication.Fundamentals.of.Global.Positioning.System.Receivers.Fundamentals.of.Telecommunications.Fundamentals.of.wavelets..Theory_.algorithms_.and.applications.Goswami_.Chan..Wiley.T.319s. Fuzzy_20Control_20Systems_20-_20Design_20and_20Analysis.munications.works..Protocols.Terminology.and.Implementation.GSM.Switching.Services.and.Protocols.Getting.Started.As.a.Financial.Planner.Rev.and.Updated.Guide.To.Budgets.And.Financial.Management.Guide.To.Digital.Signal.Processing.HF_20Antenna_20Cookbook.HF_20Filter_20Design_20and_20Computer_20Simulation.Handbook.Of.Time.Series.Analysis_.Signal.Processing_.And.Dynamics.Handbook.of.Multisensor.Data.Fusion.puting.munications.works.Harjani.Design.Of.Modulators.For.Oversampled.Converters.Wang.-.1998.High.-.Speed.Signal.Propagation.Advanced.Black.Magic.Prentice.eBook-LiB.High.-.speed.Digital.Design.-.Johnson._.Graham.High.Frequency.Techniques.An.Introduction.to.RF.and.Microwave.Engineering.Wiley-IEEE.Press.. High_20Performance_20Control.IEE.Tutorial.Meeting.on.Digital.Signal.Processing.for.Radar.and.Sonar.Applications_.1990. IEEE.._20-_20..Telecommunications.Performance.Engineering.IEEE._20-_20.Adaptive.fuzzy.power.control.for.CDMA.mobile.radio.systems.IEEE._20-_work.Modeling_.Planning.and.Design.work.Design.Guide.IP.Routing.working_3b.Straight.to.the.Core.Ieee._20-_munication.Circuits.And.Systems.works.Springer.Sep.2005.eBook._20-_20.DDU. bVIEW.And.IMAQ.Vision.Prentice.eBook._20-_20.LiB.Image.Processing.in.C.Image.Recognition.and.Classification..algorithms-marcel.dekker.-.2002.-.isbn.0824707834.-.49. works.Newnes.Jul.2004.eBook._20-_20.DD U.Implementing.Bluetooth.in.an.Embedded.Device.Industrial.electronics.for.engineers_.chemists_.and.technicians.Industrial_20Control.Integrated.Electronics.Integrated.Fiber.Optic.Receivers.Buchwald.Intermodulation_20Distortion_20in_20Microwave_20and_20Wireless_20Circuits. Introduction.To.Error.Correcting.Codes.Introduction.To.Logic.Design.-.Shiva.S.G..-.M.Dekker.1998.2Ed.Introduction.To.Sound.Processing.work.Engineering.Introduction.to.03G_munications.Introduction.to.Airborne.Radar.Introduction.to.Bluetooth.Technology_.Market_.Operation_.Profiles_._.Services. Introduction.to.CPLD.and.FPGA.Design.Introduction.to.Fiber.Optics.Introduction.to.RF.Equipment.and.System.Design.Introduction.to.RF.Propagation.Wiley.Interscience.Sep.2005.eBook._20-_20.DDU. Introduction.to.Wireless.Local.Loop.Introduction_to_Wave_Propagation_Transmission_Lines_and_Antennas.John.Wiley.And.Sons.An.Introduction.To.Parametric.Digital.Filters.And.Oscillators.John.Wiley.And.Sons.Device.Modeling.For.Analog.And.RF.CMOS.Circuit.Design.John.Wiley.And.Sons.Digital.Logic.Testing.And.Simulation.John.Wiley._20-_20.Fundamentals.of.Digital.Television.Transmission.John.Wiley._20__20.Sons._20-_works.John.Wiley._20__20.Sons._20-_20.Mobile.and.Wireless.Design.Essentials.work.Design.Aug.2004.eBook._2 0-_20.DDU.John.Wiley.and.Sons.Multi.Carrier.and.Spread.Spectrum.Systems.works.Karu.J..Signals.and.systems_.made.ridiculously.simple.2001.L.T.ISBN.0964375214.Kay.S.M..Fundamentals.of.statistical.signal.processing...estimation.theory.PH.L.T.30.Ken.Martin.Digital.Integrated.Circuit.Design.300dpi.ponents.eBook.-.LiB. works.eBook._20-_20.LiB. Kluwer.Reuse.Methodology.Manual.for.System.-.on-a-Chip.Designs.3rd.Ed..LabVIEW.Digital.Signal.Processing.McGraw.Hill.Professional.May.2005.Layout.CMOS..Circuit.Design._.Li.Simulation.Baker._Boyce.-.1997.2.Linear_20Control_20System_20Analysis_20and_20Design_20Fifth_20Edition.Linear_20Optimal_20Control.Liquidity.Liabilities.Cash.Management.Balancing.Financial.Risks.Wiley.Low-Angle_Radar_Land_Clutter_-_Measurements_and_Empirical_Models.Lumped_20Elements_20for_20RF_20and_20Microwave_20Circuits.MPEG.7.Audio.and.Beyond.Audio.Content.Indexing.and.Retrieval.John.Wiley.and.Sons.Jan.2006. puter.Vision.Springer.Aug.2005.eBook._20-_20.DDU.McGraw.-.Hill.Teach.Yours.Electricity.and.ElectronicsEbook-FLY.McGraw.Hill.-.Principles.and.applications.of.Electrical.Engineering.McGraw.Hill.Financial.Analysis.Tools.and.Techniques.a.Guide.for.Managers.McGraw.Hill._20-_ponents.McGraw.Schaum_27s.Outlines.of.Digital.Signal.Processing.McGraw.Schaum_27s.Outlines.of.Signals._.Systems.McGraw._20-_20.Hill.-.Broadband.Crash.Course.-.2002.McGraw._20-_20.Hill.-.Wireless.A.to.Z.puter._20-_20._20T.266s_20.-.oriented.Approach.to.Pattern.Recognition.AP_.19 72.Microstrip_20Filters_20For_20RF_20Microwave_20Applications.Microwave_20Circuit_20Modeling_20Using_20Electromagnetic_20Field_20Simulation. Microwave_20Component_20Mechanics.Microwave_20Electronics_20Measurement_20and_20Materials_20Characterization. Microwave_20Resonators_20and_20Filters_20For_20Wireless_20Communication.Microwave_engineering_using_microstrip_circuits_.Microwaves.and.Wireless.Simplified.Artech.House.2nd.Edition.Apr.2005.Millimeter.-.wave.Integrated.Circuits.Springer.eBook-YYePG.Mixed.Signal.And.DSP.Design.Techniques.working._20-_20.John.Wiley._.Sons.-.IEEE.Press.munications.Engineering._20-_20.Theory.and.Applications_.Second.Edition. munications.Mobile.Location.Services.The.Definitive.Guide._20-_20.Prentice.Hall.works.Wiley._20-_20.eBOOK.Model.Based.Signal.Processing.Wiley.IEEE.Press.Oct.2005.eBook._20-_20.LinG.Modern.Antenna.Design.Jun.2005.eBook-DDU.munication.Circuits.Modern.Receiver.Front.Ends.Systems.Circuits.and.Integration.Wiley.Feb.2004.eBook-DDU. Modern.Signal.Processing.Modern_20Control_20Engeneering__203rd_20ed_5d._5bOgata_5d_5bPrentice_20Hall_5d. Morgan.Kaufmann.._20-_20..Digital.Video.And.Hdtv.Algorithms.And.Interfaces.2003.Multi.-.Standard.CMOS.Wireless.Receivers_.Analysis._.Design.Multicarrier.Techniques.for.04G_munications.Multivariable.Control.Systems.An.Engineering.Approach.Springer.eBook-TLFeBOOK.Nano.CMOS.Circuit.and.Physical.Design.Network.Calculus.A.Theory.of.Deterministic.Queuing.Systems.for.the.Internet.Networks_20and_20Devices_20Using_20Planar_20Transmissions_20Lines.Neural_20Systems_20For_20Control.New.technologies.for.WLAN.munications.Pocket.Book.Newnes.Guide.to.Television._.Video.Technology.Newnes.Radio.and.RF.Engineering.Pocket.Book.Newnes_20Industrial_20Control_20Wiring_20Guide.Next.Generation.Mobile.Systems.3G.and.Beyond.John.Wiley.and.Sons.May.2005.eBook._20-_20. DDU.Nixon_.Aguado..Feature.Extraction.and.Image.Processing.2002.Noise.In.Receiving.Systems.Nonlinear.Microwave.And.RF.Circuits.2nd.Edition.Nonlinear_20Microwave_20Circuit_20Design.ON.Analog.Integrated.Circuits.OReilly.Digital.Video.Hacks.May.2005.eBook._20-_20.DDU.OReilly.RFID.Essentials.Jan.2006.O_27Reilly._20-_20._20802._20-_works-.The.Definitive.Guide. Observers_20in_20Control_20Systems.Op.Amp.Applications..Op.Amps.Design.Application.and.Troubleshooting.Op.Amps.for.Everyone.Design.Reference.Operational.Amplifiers.Design.and.Applications.munications.Essentials.munications.Rules.of.Thumb.working.Handbook.Mcgraw._20-_20.Hill.Optical.System.Design.Optical.Through._20-_munications.Handbook.Optical.signal.processing.Vanderlugt.A..Wiley_.1991pi.L.T.180s.PEo.Optimal.Filtering.Optimal_20Control_20Linear_20Quadratic_20Methods.Optimal_20Sampled_20Data_20Control_20Systems.Optimizing.Wireless._20-_20.RF.Circuits.work.Handbook.Pattern.Classification.And.Learning.Theory.Lugosi.nguage.Processing.works.Polling_.Scheduling_.and.Traffic.Cont rol.munications.Phased.Array.Antenna.Handbook.Artech.House.Publishers.Second.Edition.eBook-kB.Phased_20Array_20Antennas_20Hansen_20R.C._20_Wiley_1998__ISBN_20047153076X__200dp i__T__504s__EE_.Photodetection._20__20.Measurement._20-_20.Maximizing.Performance.in.Optical.Systems. Practical.Analog.And.Digital.Filter.Design.Practical.Electronics.for.Inventors.Practical.FPGA.Programming.in.C.Prentice.Hall.PTR.Apr.2005.yout._20-_e.of.Stock.Lenses.Practical.Rf.Pcb.Design.Geoff.Smithson.Scanned.Practical.Rf.System.Design._20-_20.Egan.Practical_20Applications_20of_20Computational_20Intelligence_20for_20Adaptive_20Control. Practical_20Approach_20to_20Signals_20Systems_20and_20Control.Pragmatic.Introduction.to.Electronic.Engineering.0._v1_.works.John.Wiley.and.Sons.munication.system.simulation.with.wireless.applications._20-_20.Prentice.Hall. Principles.Of.Corporate.Finance.Principles.of.Asynchronous.Circuit.Design.-.A.Systems.Perspective.Principles.of.Digital.Transmission.With.Wireless.Applications.Principles.of.Sigma.Delta.Conversion.for.Analog.to.Digital.Converters.munication.Systems.eBook._20-_20.TLFeBOOK. Programmable.Digital.Signal.Processors.Architecture.Programming_.and.Applications. munication.System.Design.QoS.in.Integrated.03GNetworks.2002.Quantitative.Finance.for.Physicists.An.Introduction.Queueing.Theory.With.Applications.to.Packet.Telecommunication.Springer.eBook._20-_20.YYePG. RDS..The.Radio.Data.System.RF-Microwave_20Circuit_20Design_20for_20Wireless_20Applications.ponents.and.Circuits.munications.munications.RFID.Field.Guide.Deploying.Radio.Frequency.Identification.Systems.Feb.2005.eBook._20-_20.LiB. RFID.For.Dummies.Mar.2005.eBook._20-_20.LinG.RFID.Sourcebook.Prentice.Hall.PTR.RFID._20-_20.Read.My.Chips_.RF_20__20Microwave_20Radiation_20Safety_20Handbook.RF_20and_20Microwave_20Wireless_20Systems.Radar.Systems_.Peak.Detection.and.Tracking.Radar.Technology.Encyclopedia._20-_20.1998.Radar_20Principles.munication.and.Sensor.Applications.Radio.Engineers_27.Handbook._20-_20._2001e_20-_20.-.d.-.Terman.Radio.Frequency.Circuit.Design.Radio.Frequency.Transistors.Radio.Shack.-.Getting.started.in.electronics.Radio.Shack.Engineer_27s.Mini.-._5bNotebook.T.52s_5d.Radio._.Electronics.Cookbook.Radio_20Frequency_20and_20Microwave_20Communication_20Circuits.Radiometric.Tracking.Techniques.for.Deep.Space.Navigation.Radiosity.and.realistic.image.synthesis.Cohen.M.F._.Wallace.J.R..AP_.1995.Real.802.11.Security.Wi._20-_20.Fi.Protected.Access.And.802.11i.Addison.Wesley.eBook-LiB. Real.Analog.Solutions.for.Digital.Designers.Real.World.Digital.Audio.Peachpit.Press.No05._20v.200.Real._20-_pression--Techniques.And.Algorithms.Rf.Cmos.Power.Amplifier._20-_20.Ebook.Kluwer.Inter.Hella._.Ismall.Risk.Management.And.Capital.Adequacy.McGraw.Hill.SIP.Demystified.MUNICATIONS.HANDBOOK.munication.Engineering.eBook._20-_20.EEn.Satellite.Handbook.working.Principles.and.Protocols.John.Wiley.and.Sons.Oct.2005.eBook._20-_20.DDU. Schaums.Outline.Of.Theory.And.Problems.Of.Electric.Circuits.eBook.Secrets.of.RF.Circuit.Design._.Third.Edition.Securing.and.managing.WLAN.Shannon._20-_20.TheoryComm.munication.Fundamentals.of.RF.System.Design.and.Application. Signal.Analysis.Alfred.Mertins.Signal.Analysis.Time.Frequency.Scale.and.Structure.RL.Allen_ls.Signal.Detection.and.Estimation.munications.Handbook._20-_20.CRC.Press.-.2005.Signal.analysis.wavelets.filter.banks-Mertins.A..Wiley_.1999.Signals.And.Systems.Signals._20__20.Systems.with.MATLAB.Applications._20-_20.Orchard.Publications. munications.Sliding_20Mode_20Control_20in_20Engineering.Smart.Antennas.CRC.Press.Jan.2004.eBook-DDU.Some.Design.Aspects.on.RF.CMOS.LNAs.and.Mixers.Sonet.or.SDH.Demystified.Space._20-_20.Time.Coding.John.Wiley.And.Sons.eBook.Space._20-_munications.Specification.of.the.Bluetooth.System.Spectrum.Wars.Speech.Coding.Algorithms.Foundation.and.Evolution.of.Standardized.Coders.Wiley.eBook._20-_2 0.KB.works.Speech.Separation.By.Humans._20__20.Machines.Springer.eBook._20-_20.YYePG.Stability_20Analysis_20of_20Nonlinear_20Microwave_20Circuits.pression.to.Advanced.Video.Coding.IEEE.Standard.Handbook.of.Audio.and.Radio.Engineering.Standard.Handbook.of.Video.and.Television.Engineering_.4th.ed.Starting.Electronics.-.Elsevier.-.3rd.Edition.-.2005.Statistical.and.Adaptive.Signal.Processing.Supervised.and.Unsupervised.Pattern.Recognition.Synthesis.and.optimization.of.DSP.algorithms.Constantinides_.Cheung_.Luk..Kluwer_.2004.T.144s_20Bayesian.Approach.to.Image.Interpretation.Kopparapu_.Desai..Kluwer_.2002.T.181s_20Wavelets_.with.applications.in.signal.and.image.processing.Bultheel.A..2002.T.212s_20Brandwood..Fourier.transforms.in.radar.and.signal.processing.2003.T.359s_20Mann.S..Intelligent.image.processing.Wiley_.2002.T.406s_20Dudgeon.D._.Mersereau.R._.Merser.R._.Multidimensional.Digital.Signal.Processing.199 5.T.548s_20Ballard.D.H._.Computer.vision.Brown.C.M..PH_.1982.ISBN.0131653164.T.621s_20Image.analysis.and.mathematical.morphology.Serra.J..AP_.1982.300dpi.CsIp.TAB.Electronics.Guide.to.Understanding.Electricity.and.Electronics.eBook.-.EEn.Telecom.Crash.Course.Telecom.Dictionary.Telecommunication.Circuit.Design._20-_20.Second.Edition.Telecommunications.Essentials.CHM.Telecommunications.Regulation.Teletraffic.Engineering.Handbook.The.Art.and.Science.of.Analog.Circuit.Design.The.Art.of.Electronics.02ed.munications.Professional..A.Guide.for.Engineers.and.Managers. working.The.Engineer_27s.Guide.to.Decoding._.Encoding.The.Engineer_27s.Guide.to.Standards.Conversion.The.Great.Telecom.Meltdown.Artech.House.Jan.2005.eBook._20-_20.LiB.works.munications.Handbook.The.Mobile.Radio.Propagation.Channel._20-_20.Second.Edition.-.Wiley.The.Personal.Finance.Calculator.McGrawHill.munication.Applications.Handbook.The.Telecommunications.Handbook.The.Wireless.Data.Handbook._20-_20.Fourth.Edition.Thetrated.dictionary.of.electronics.Troubleshooting.Analog.Circuits.US.Navy._20-_20.Digital.Data.Systems.Ultra.Wideband.Radio.Technology.ing.Coded.Signals.Understanding.Cellular.Radio.munications.Understanding.Digital.Signal.Processing.Understanding.Digital.Terrestrial.Broadcasting.MAZ._20-_20.Artech.House. munications.Understanding.Telephone.Electronics.Understanding_20Microwaves_20_Scott_.rmation.Retrieval.IRM.eBook._20-_20.YYePG.Video.Demystified.A.Handbook.For.The.Digital.Engineer.munications.Voice.Over.802.11.W._20-_20._20for.03G_works.munications.System.Waveguide_20Handbook.Wavelets.For.Kids.A.Wavelets.For.Kids.B.Wideband.TDD.WCDMA.for.the.Unpaired.Spectrum.John.Wiley.Sons.May.2005.eBook._20-_20.Lin G.Wiley.-.Essentials.of.Financial.Analysis.Wiley._20-_works_.IP.and.the.Internet.-.Protocols_.Design.and.Operation.Wiley._20-_20.Digital.Image.Processing.WK.Pratt.-.Third.Edition.2001.munication.Systems._20-_20.Prentice.Hall.PTR.munication.Technologies.munication.Technology.munications.Wireless.Data.Demystified.McGraw.Hill.eBook._20-_20.LiB.Wireless.Data.Technologies.Reference.Handbook.John.Wiley.and.Sons.Wireless.Foresight.Scenarios.of.the.Mobile.World.in.2015.John.Wiley.and.Sons.eBook._20-_20.Li B.Wireless.Internet.Telecommunications.Artech.House.Publishers.eBook._20-_20.YYePG. working.with.ANSI._20-_20._2041__20-_20.-.Second.Edition.works.First._20-_20.Step..2005.munication.Systems.Springer.Verlag.Telos.Sep.2004.ISBN0387227849. Wireless.Technology.Protocols.Standards.and.Techniques.Young_.Gerbrands_.van.Vliet..Fundamentals.of.image.processing.Delft.U._.1998.T.11._5bT.270s_5dJohnson.D.H._.Wise.J.D..Fundamentals.of.electrical.engineering.1999._5bT.498s_5dGustafsson.F..Adaptive.Filtering.and.Change.Detection.Wiley_.2000._Delmar__20Modern_20Control_20Technology--Components_20__20Systems_20_2nd_20Ed._. dsp.algorithms.for.programmers.eWiley.Mobile.Fading.Channels._20-_20.-Modelling_.Analysis._.Simulation.electronics_20technician_20volume_201_20-_20safety.electronics_20technician_20volume_202_20-_20administration.electronics_20technician_20volume_203_20-_20communications_20systems.electronics_20technician_20volume_204_20-_20radar_20systems.electronics_20technician_20volume_206_20-_20digital_20data_20systems.electronics_20technician_20volume_207_20-_20antennas_20and_20wave_20propagation. low.power.asynchronous.DSP.numerical_20methods_20in_20electromagnetics.operational.amplifiers.-.2nd.edition.practical_aspects_of_feedback_control.structure.and.interpretation.of.signals.and.systems.下載地址:/file/f5ddfade86600_electrical_engineering_books.rar。

TL-WA850RE 300Mbps 通用 Wi-Fi 扩展器 使用说明书

TL-WA850RE 300Mbps 通用 Wi-Fi 扩展器 使用说明书

TL-WA850RE300Mbps Universal Wi-Fi Range ExtenderCOPYRIGHT & TRADEMARKSSpecifications are subject to change without notice. is a registered trademark of TP-LINK TECHNOLOGIES CO., LTD. Other brands and product names are trademarks or registered trademarks of their respective holders.No part of the specifications may be reproduced in any form or by any means or used to make any derivative such as translation, transformation, or adaptation without permission from TP-LINK TECHNOLOGIES CO., LTD. Copyright ©2015 TP-LINK TECHNOLOGIES CO., LTD.All rights reserved.FCC STATEMENTThis equipment has been tested and found to comply with the limits for a Class B digital device, pursuant to part 15 of the FCC Rules. These limits are designed to provide reasonable protection against harmful interference in a residential installation. This equipment generates, uses and can radiate radio frequency energy and, if not installed and used in accordance with the instructions, may cause harmful interference to radio communications. However, there is no guarantee that interference will not occur in a particular installation. If this equipment does cause harmful interference to radio or television reception, which can be determined by turning the equipment off and on, the user is encouraged to try to correct the interference by one or more of the following measures:∙Reorient or relocate the receiving antenna.∙Increase the separation between the equipment and receiver.∙Connect the equipment into an outlet on a circuit different from that to which the receiver is connected.∙Consult the dealer or an experienced radio/ TV technician for help.This device complies with part 15 of the FCC Rules. Operation is subject to the following two conditions:1) This device may not cause harmful interference.2) This device must accept any interference received, including interference that maycause undesired operation.Any changes or modifications not expressly approved by the party responsible for compliance could void the user’s authority to operate the equipment.Note: The manufacturer is not responsible for any radio or tv interference caused by unauthorized modifications to this equipment. S uch modifications could void the user’s authority to operate the equipment.FCC RF Radiation Exposure StatementThis equipment complies with FCC RF radiation exposure limits set forth for an uncontrolled environment. This device and its antenna must not be co-located or operating in conjunction with any other antenna or transmitter.“To comply with FCC RF exposure compliance requirements, this grant is applicable to only Mobile Configurations. The antennas used for this transmitter must be installed to provide a separation distance of at least 20 cm from all persons and must not be co-located or operating in conjunction with any other antenna or transmitter.”CE Mark WarningThis is a class B product. In a domestic environment, this product may cause radio interference, in which case the user may be required to take adequate measures.Canadian Compliance StatementThis device complies with Industry Canada license-exempt RSS standard(s). Operation is subject to the following two conditions:(1)This device may not cause interference, and(2)This device must accept any interference, including interference that may cause undesired operation of the device.Cet appareil est conforme aux norms CN R exemptes de licence d’Industrie Canada. Le fonctionnement est soumis aux deux conditions suivantes:(1)cet appareil ne doit pas provoquer d’interférences et(2)cet appareil doit accepter toute interférence, y compris celles susceptibles de provoquer un fonctionnement non souhaité de l’appareil.Industry Canada StatementComplies with the Canadian ICES-003 Class B specifications.Cet appareil numérique de la classe B est conforme à la norme NMB-003 du Canada.This device complies with RSS 210 of Industry Canada. This Class B device meets all the requirements of the Canadian interference-causing equipment regulations.Cet appareil numérique de la Classe B respecte toutes les exigences du Règlement sur le matériel brouilleur du Canada.Korea Warning Statements당해무선설비는운용중전파혼신가능성이있음.NCC Notice & BSMI Notice注意!依據低功率電波輻射性電機管理辦法第十二條經型式認證合格之低功率射頻電機,非經許可,公司、商號或使用者均不得擅自變更頻率、加大功率或變更原設計之特性或功能。

Radio Frequency (RF) 系列产品说明书

Radio Frequency (RF) 系列产品说明书


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Option (1): Prepared to connect FM Approved remote display type FHX40 from Endress+Hauser
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FMR232-
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1. Control room equipment may not use or generate over 250 Vrms. 2. Use FM Approvals Entity-Approved intrinsic safety barrier with Voc or Vt £ Vmax, Isc or It £ Imax, Ca ³ Ci + Ccable, La ³ Li + Lcable

Detection of the Intrinsic Size of Sagittarius A through Closure Amplitude Imaging (include

Detection of the Intrinsic Size of Sagittarius A through Closure Amplitude Imaging (include

a r X i v :a s t r o -p h /0404001v 1 31 M a r 2004Detection of the Intrinsic Size of Sagittarius A*through Closure Amplitude ImagingGeoffrey C.Bower,1∗Heino Falcke,2Robeson M.Herrnstein,3Jun-Hui Zhao,4W.M.Goss,5,Donald C.Backer 11Astronomy Department &Radio Astronomy Laboratory,University of California,Berkeley,CA 94720,USA 2Radio Observatory Westerbork,ASTRON,P.O.Box 2,7990AA Dwingeloo,The Netherlands 3Department of Astronomy,Columbia University,Mail Code 5246,550West 120th St.,New York,NY 10027,USA 4Harvard-Smithsonian Center for Astrophysics,60Garden Street,MS 78,Cambridge,MA 02138,USA 5National Radio Astronomy Observatory,Array Operations Center,P.O.Box O,Socorro,NM 87801,USA ∗To whom correspondence should be addressed;E-mail:gbower@.We have detected the intrinsic size of Sagittarius A*,the Galac-tic Center radio source associated with a supermassive black hole,showing that the short-wavelength radio emission arises from very near the event horizon of the black hole.Radio observations with the Very Long Baseline Array show that the source has a size of 24±2Schwarzschild radii at 7mm wavelength.In one of eight 7-mm epochs we also detect an increase in the intrinsic size of 60+25−17%.These observations place a lower limit to the mass density of SgrA*of1.4×104solar masses per cubic astronomical unit.Sagittarius A*(Sgr A*)is the compact,nonthermal radio source in the Galactic Center associated with a compact mass of4×106M⊙(1,2,3).It is the best established and closest supermassive black hole candidate and serves as the prime test case for the black hole paradigm.Emission at radio,near-infrared,and X-ray wavelengths traces processes in the environment of the event horizon(4,5,6,7,8,9).High resolution radio imaging of Sgr A*can ultimately distinguish between the many different models for the emission,accretion and outflow physics of the source as well as provide an important test of strong-field gravity(10).Sgr A*has been a target of such observations for the past30years(11).Its intrinsic size and structure have remained obscured,however,because radio waves from Sgr A*are scattered by turbulent interstellar plasma along the line of sight(12).The scatter-broadened image of Sgr A*is an ellipse with the major axis oriented almost exactly East-West and a quadratic size-wavelength relation.The turbulent plasma is parametrized with a power-law of turbulent energy density as a function of length scale with outer and inner scales that correspond to the scale on which turbulence is generated and damped,respectively.Scattering theory predicts that the scatter-broadened image will be a Gaussian when the inner length scale of the turbulent medium is larger than the longest baseline of the observing interferometer(13).Addition-ally,the scatter-broadened image size will scale quadratically as a function of wavelength. In the case of Sgr A*,the longest interferometer baseline used in our analysis b max∼2000 km corresponds to a length scale in the scattering medium D scattering/D source×b max∼25 km,where D source=8kpc is the distance from Sgr A*to the Earth and D scattering=100 pc is the distance from Sgr A*to the scattering screen(12).This scale is much less than the predicted and measured values of the inner scale,which fall in the range102to105.5km(14,15).The amplitude of turbulence in the Galactic Center scattering screen is∼2−3orders of magnitude greater than what is seen in the next most powerful scat-tering region,NGC6334B(16),however,suggesting that the Galactic Center case may be atypical.The presence of strong scattering has pushed observations to shorter and shorter wavelengths where scattering effects decrease and intrinsic source structure may dom-inate,creating a deviation from the measured size-wavelength law.On the basis of extensive observations with the National Radio Astronomy Observatory’s Very Long Baseline Array(VLBA),L98measure the index of the size-wavelength power-law to be α=1.99±0.03(17).L98also claim a deviation from the scattering law in the minor axis at7mm wavelength(43GHz),implying an intrinsic size of72Schwarzschild radii (R s)(18).Unfortunately,precise measurements of the size of Sgr A*are seriously hampered by calibration uncertainties related to the variable antenna gain and atmospheric opacity at the low antenna elevations necessary to observe Sgr A*from the northern hemisphere. Closure amplitudes have been used to constrain the size of Sgr A*with VLBI observations at3.4mm(19).The closure amplitude does not rely on calibration transfer from another source as traditional imaging methods do and is independent of all station-dependent amplitude errors.This method does not,however,eliminate baseline-dependent errors such as variable decorrelation(which also influence conventional calibration and imaging techniques).The closure amplitude is conceptually related to the closure phase,a more well-known quantity which is also independent of station-based gain errors.The principle drawback of closure amplitude analysis for simple source structures is the reduction in the number of degrees of freedom relative to a calibrated data set.The number of independent data points for a7-station VLBA experiment is reduced by a factor14/21.Additionally,the closure amplitude method can not determine the absoluteflux density for the source. These shortcomings are more than offset by the confidence that the result gives through its accurate handling of amplitude calibration errors.We describe here the analysis of new and archival VLBA data through closure am-plitude and closure phase quantities.We analyze3new experiments including data at 1.3cm,6new experiments including data at0.69cm,as well as10experiments from the VLBA archive including data at6,3.6,2.0,1.3,0.77,0.69and0.67cm wavelength.Observations and Initial Data ReductionSix new observations were made with the VLBA as part of our Very Large Arrayflux density monitoring program(20).Three observations were made in each of two sepa-rate epochs in July/August2001and April/May2002(STable1).In thefirst epoch, observations at1.3cm and0.69cm were interleaved over5hours.In the second epoch, observations were obtained only at0.69cm in order to maximize the signal to noise ra-tio(SNR)of thefinal result.All observations were dual circular polarization with256 Mbits/sec recording rate.We also analyzed a number of experiments from the VLBA archive over the wavelength range of6.0cm to0.67cm(STable1).The experiments BS055A,B and C were those analyzed by L98.The experiment BB113was previously analyzed(21).Initial data analysis was conducted with the NRAO Astronomical Imaging Processing System(22).Standard fringe-fitting techniques were employed to remove atmospheric and instrumental delays from the data(SOM text).High SNR fringes were detected for most stations on the compact source NRAO530(J1733-1302),indicating the overall quality of the data.Due to the relatively larger size of Sgr A*,fringes were obtained for a subset of5to8stations(STable1).Data were then averaged over wavelength and time for each experiment.The quality of thefinal result is dependent upon the visibility averaging time.The longer the averaging time,the higher the SNR of the closure amplitude calculation(23).On the other hand, as the averaging time approaches the phase decorrelation time,the closure amplitudes cease to be accurate.It is not necessary,however,to determine the best averaging time precisely,since neither of these effects is a strong function of time(23).The results that we give are for an averaging time of30seconds,but wefind that for averaging times of 15to120seconds the estimated intrinsic size of Sgr A*does not differ by more than10% (SOM text).No amplitude calibration was applied at any stage.The averaged data were then written to textfiles for analysis by our own analysis programs,external to AIPS.Closure Amplitude and Closure Phase Analysis of a Single GaussianWe form the closure amplitude from the measured visibilities and average the closure amplitudes over time.Closure amplitudes were averaged over scans,which were5to15 minutes in duration.The code uses the scatter in the closure amplitudes before averaging to determine the error in the closure amplitude.Only independent closure amplitudes were formed(24).We selected visibility data only with station elevations>10◦to reduce sensitivity to phase decorrelation,which is more significant at low elevations.We also excluded data at (u,v)distances greater than25Mλat6.0cm,50Mλat3.6cm,150Mλat2.0cm and 1.3cm,and250Mλat0.69cm.These sizes are comparable to the expected size of Sgr A*at each wavelength.Visibility amplitudes beyond the cutoffwere indistinguishable by inspection from noise.This(u,v)-distance limit reduced sensitivity to the noise bias or station-dependent differences in the noise bias.Results were not strongly dependent onthe value of this cutoff.Model visibilities for each baseline and time datum were computed for an elliptical source of a givenflux density S0,major axis size x,minor axis size y and position angle φ.In addition,a noise bias was added in quadrature to each model visibility.Our model visibility amplitude(squared)on baseline ij is thenA2ij=S20e−D0((u′ij x)β−2+(v′ij y)β−2)+N2ij,(1)where D0=2(π)2,N ij is the noise bias,and u′ij and v′ij are baseline lengths in log2units of wavelength in a coordinate system rotated to match the position angleφ.Model closure amplitudes were then formed from these model visibilities.We determine the best-fit parameters using a non-linearfitting method that minimizesχ2between the model and measured closure amplitudes(SFig.1,STable2).Wefind the reducedχ2for the amplitudesχ2A≈1for all experiments.In the case of an image produced by interstellar electron scattering on baselines longer than the inner scale of turbulence,βis the power-law index of electron densityfluctuations (13).The parameterβis related to the exponentαof the scattering law(size∝λα)as β=α+2,allowing an independent check of theλ2law(13,14,15).For the case of the Galactic Center scattering we expectβ=4,in which case Equation1is a Gaussian function and x and y are the FWHM in the two axes.Allowingβto be unconstrained in ourfits,wefindβ=4.00±0.03,which is consistent with the expectation of scattering theory(SFig.3).All remaining analysis is conducted with the assumption thatβ=4.The introduction of the noise bias to the model changes our calculation from a pure closure amplitude to a noise-biased closure amplitude.We found that our results did not require that we consider the noise bias as dependent on station or time(SOM text).Thus, we chose N ij(t)=N0because it is simpler computationally and has a smaller number ofindependent parameters.Errors in the model parameters were determined by calculatingχ2for a grid of models surrounding the solution andfitting constantχ2surfaces(SFig.2).Monte Carlo simula-tionsfind confidence intervals that are smaller by a factor of two than determined from the χ2analysis,suggesting that the dominant sources of error are baseline-based errors such as phase decorrelation,which were not included in the Monte Carlo simulations(SOM text).Closure phases were formed,averaged and analyzed in a manner similar to the closure amplitudes.We tested the closure phases against the hypothesis that they are all zero. This hypothesis is the case for a single elliptical Gaussian and other axisymmetric struc-tures with sufficiently smooth brightness distributions.An axisymmetric disk is a notable exception to this hypothesis since it induces ringing in the transform plane.The reduced χ2for this hypothesisχ2φ≈1for all experiments(STable2),indicating no preference for multiple components,non-axisymmetric structure or disk-like structure.Although the solutions for a single Gaussian component are sufficiently accurate,we did search the parameter space for two component models.To do this,we performed a minimization ofχ2with respect to closure amplitude and closure phase jointly.The reducedχ2for these models was roughly equal to the values for the single Gaussian component despite the addition of several degrees of freedom.We also calculated upper limits to theflux densities of secondary components that are in the range2-10%,typically (SFig.4,STable2).The absence of any improvement indicates that a single Gaussian component is sufficient and the simplest model of the data.This absence is particularly significant for the cases whereχ2>1and suggests,as noted before,that the results are dominated by closure errors rather than improperly modeled structure.Scattering Law and Intrinsic SizeWe determined the size of the major and minor axes of Sgr A*for each experiment(Fig.1 and2,STable2).The major axis is oriented almost exactly East-West.The major axis size is measured much more accurately than the minor axis size because of the poorer North-South resolution of the array.All major axis measurements at1.3and0.69cm are larger than the scattering size determined by L98(25)and the new scattering size that we determine below,although the difference is statistically significant in only one epoch at0.69cm.Minor axis measurements are distributed about the scattering result and no one differs significantly from the expected result.The L98scattering law is adequate for the minor axis measurements as a function of wavelength(Fig.3).All the measured minor axis sizes agree with the scattering law to better than3σ.The data are also consistent with a constant position angle of78.0+0.8−1.0 deg withχ2ν=2.2forν=6degrees of freedom.We determinefits to the major and minor axis sizes as a function of wavelength using subsets of the data with a minimum wavelengthλmin of2.0cm,1.3cm,0.6cm and0.3 cm(STable3).The lastfit includes the3.4mm circular Gaussianfits of for the major axis only(19).There are twofits for each subset,allowingαto vary andfixingα=2.χ2νis less than3for the minor axis case withλ≥0.6cm,confirming that the solution is adequate forα=2.The major axis data,however,are discrepant from the L98and the new scattering law (Fig.3).All of the7mm results fall above the L98scattering law.Two of these points are significantly different at greater than3σ.The L98scattering law predicts a size of690µarcsec at0.69cm,which is∼7σfrom the measured size(712+4−3µas)and smaller than any of the measured sizes(Fig.2).An attempt tofit a scattering law withαmajor=2toall data withλ>0.6cm givesχ2ν=24for6degrees of freedom,demonstrating that the hypothesis can be strongly rejected.In fact,the1.35cm major axis size is also discrepant with the best-fitαmajor=2scattering law,givingχ2ν=5.6for3degrees of freedom.We consider two alternative models for our resuls:case A,the scattering power-law exponentαmajor is not exactly2;or,case B,intrinsic structure in Sgr A*is distorting the size-wavelength relation at short wavelengths.For case A,wefind adequate solutions for all data at wavelengths≥0.3cm with αmajor=1.96±0.01.The result is clearly discrepant with scattering theory which re-quiresβ=4and marginally discrepant with our determination of the scattering theory parameterβ=4.00±0.03(SFig.3),since scattering theory predicts thatα=β−2.For case B,we determine a new scattering law from observations withλ≥2.0cmthat is even less than that of andαmajor=2.This solution has a scale parameterσ1cmmajorL98,increasing the discrepancy at short wavelengths.Removing this new scattering law in quadrature gives an intrinsic size of0.7±0.1mas at1.35cm,0.24±0.01mas at0.69 cm and0.06±0.05mas at0.35cm(Table1).On the basis of the disagreement betweenβandα,we reject case A and claim that we have determined the size of intrinsic structure in Sgr A*at1.35and0.69cm.The two cases predict substantially different sizes at20cm.For the major axis case A predicts541±2mas while case B predicts595±3.The20.7cm(ν=1450MHz) major axis size624±6mas measured with the VLA A-array(26)is discrepant with both of these cases,although more strongly with case A.These measurements are particularly difficult since the source is only partially resolved in the A-array:the synthesized beam is about2.6×0.9arcsec oriented North-South.Additionally,extended structure in the Galactic Center makes estimation of the size strongly dependent on the estimate of the zero-baselineflux density.We attempted to verify the20cm size with analysis of three VLA A-array observa-tions at21.6cm obtained originally for polarimetry(8).Results for each of the three experiments were similar and dominated by systematic errors that make an estimate of the intrinsic size difficult.We were unsuccessful at analyzing these experiments with our closure amplitude technique,possibly due to the poor resolution of Sgr A*and inability of our code to handle the large number of stations.In any case,the reliability of ampli-tude calibration of the VLA at20cm reduces the need for closure amplitude analysis. We imaged all baselines and measured the totalflux density of Sgr A*byfitting a two-dimensional Gaussian to the central3′′.For all epochs,wefind an error in the totalflux density of10mJy.We determined the size byfitting in the(u,v)plane with the totalflux densityfixed and with a minimum cutoffin(u,v)distance.For values of the totalflux density that range from−1σto+1σand for a minimum(u,v)distance from20to120 kλ,wefind that the major axis size varies systematically from580to693mas.The minor axis is very poorly constrained.We estimate the size from the mean of these results as 640±40mas.We consider this to be a more reasonable estimate of the error in the size of Sgr A*than previously given.This size is consistent at<1σwith case B and∼1.5σwith case A,favoring slightly detection of the intrinsic size.Although all minor axis data are adequatelyfit withαminor=2,we can check the consistency of our results by estimating intrinsic sizes for this axis in the same way. The minor axis sizes show the same trend as the major axis sizes:smaller than the L98scattering law at long wavelengths and larger than the L98scattering law at short wavelengths(Fig.3).Using the solution forαminor=2andλ≥2.0cm,we estimate intrinsic sizes of1.1±0.3mas at1.35cm and0.26±0.06mas at0.69cm.These are comparable to the sizes determined for the major axis.For the case of unconstrained power-law indexfit to all data,wefindαminor=1.85+0.06,marginally consistent with no−0.06intrinsic source.Changes In the Source Size with TimeAt0.69cm,the only measurement deviating significantly from the mean result is inµas while the mean result the major axis for BB130B.The BB130B result is770+30−18µas.We note that the greatest deviation in is712+4−3µas giving a difference of58+30−19the0.69cm position angle also occurs for BB130B,although the difference is significant only at the2σlevel.Any such deviation would indicate a non-symmetric expansion or a non-symmetric intrinsic source size.We can estimate the change in the size of the intrinsic source between BB130B and the mean size by subtracting in quadrature the case B scattering size from each.As stated above,the mean result implies an intrinsic size ofmas.Thus, 0.24±0.01mas.The intrinsic size implied by the BB130B result is0.38+0.06−0.04the growth in major axis size is0.14+0.06mas in the N-S direction.We cannot associate−0.04this change in structure with aflux density change.This maximum in the size comes ∼10days before detection of an outburst at0.69cm with the VLA(20).The following epoch,BB130C,occurs only two days before this outburst but shows no deviation from the mean size,although the size is particularly poorly determined in this case.The Interstellar Scattering ScreenThe image of a scattered source is created by turbulent plasma along the line of sight. The minimum time scale for the scattered image to change is the refractive time scale, the time in which the relative motions of the observer,turbulent plasma and background source lead to the background source being viewed through a completely different region of the interstellar plasma.The refractive time scale for Sgr A*is∼0.5λ2y cm−2given a relative velocity of100km s−1(13).At our longest wavelength for VLBA observations,6cm,then the time scale is20y.At our shortest wavelength of7mm,the time scale for refractive changes is3months.Our observations are distributed over a much larger time frame than three months,implying that the mean result may be affected by refractive changes.Two subsets of the archival data have much smaller span,however.The BS055experi-ments cover6.0to0.69cm in1week and the BL070and BB113experiments cover6.0cm to0.67cm in3months.These data sets include all of the2.0cm and longer wavelength data.If we compare the0.7cm size,we see that it is larger in these quasi-simultaneous experiments than in the mean of all experiments and also larger than the expectation ofthe new scattering law.Wefind0.69cm major axis sizes of728+16−11µas and713+12−9µasfor BS055C and BL070B,respectively,both larger than the mean size of712+4−3µas(STa-ble2).We conclude that if refractive effects are altering the short wavelength results, then their effect is to reduce the deviation from the scattering law,not enhance it.DiscussionOur results allow us to probe the mechanisms responsible for accretion,outflow and emission in the vicinity of the black hole.We can compare the measured7mm intrinsic major axis size of24R s and its dependence on wavelength with expected values(Fig.4). The intrinsic size of the major axis decreases with wavelength and is best-fit with a power-law as a function of wavelength with indexαintrinsic=1.6±0.2.Wefind for the minor axis a similar valueαintrinsic=2.1±0.5.Assuming that the source is circularly symmetric and using the meanflux density of1.0Jy at7mm(20),we compute a brightness temperature T b=1.2×1010× λThe wavelength-dependent size of Sgr A*now unambiguously shows that the source is stratified due to optical depth effects.We rule out models in which the emission originates from one or two zones with simple mono-energetic electron distributions(27). These models predict a size which is constant with wavelength and is larger than our measured size.The results are well-fit by a multi-zone or inhomogeneous model,in which the size is equal to the radius at which the optical depth is equal to unity(28).In a jet model, declining magneticfield strength,electron density and electron energy density contribute to a size that becomes smaller with wavelength.A detailed jet model for Sgr A*predicts an intrinsic size of0.25mas at0.69cm and0.6mas at1.3cm(Fig.4)(29).Exact values and wavelength dependence are a function of a number of parameters including the relative contributions of the extended jet and the compact nozzle component of the jet. The jet model also predicts that the source should be elongated with an axial ratio of4:1. The apparent measured symmetry in the deconvolved sizes in each axis,however,does not imply that the intrinsic source is symmetric.For example,an elongated intrinsic source that is oriented at45degrees to the scattering axis will produce equal deconvolved sizes in each axis.Modeling of the closure amplitudes with a complete source and scattering model is necessary to determine the elongation for the most general case.The thermal,high accretion rate models such as Bondi-Hoyle accretion(30)and ad-vection dominated accretionflows(31)require T e∼109K,which overpredicts the size in each axis by a factor of3.This disagreement confirms the elimination of these models on the basis of the polarization properties of Sgr A*(9).On the other hand,the radiatively inefficient accretionflow(RIAF)model(32)has a lower accretion rate and higher T e,com-patible with the polarization and with this measurement.The RIAF model also predicts an inhomogeneous electron distribution consistent with a size that reduces with decreasingwavelength.Both the RIAF model and the jet model are similar in the electron energy distribution and magneticfield distribution required to produce the observedflux density within the observed size.These models differ principally in the relative contribution of thermal electrons to emission in the submillimeter region of the spectrum.Extrapolating our size-wavelength relation to longer wavelengths,we estimate a size at2cm of130R s with a characteristic light travel time of85minutes.This is comparable to the shortest time scale for radio variability detected,2hours,during which the2.0cm radioflux density changed by20%(8).The smooth nature of the spectrum from90cm to7mm,suggests that our size-wavelength relation holds over that entire range(33).Our relation implies a size<2R s at1.3mm,comparable to the size of the event horizon.The decrease of the source size with wavelength cannot continue much farther due to thefinite size of the central object itself.In the millimeter and submillimeter, however,the spectral index rises(34),indicating that there may be a break in the size-wavelength relation.Ultimately,the size of the event horizon can be viewed as setting a limit on the wavelength of the peak emission.The strong break in the spectrum between the submillimeter and the NIR may correspond to the wavelength at which the source size becomes comparable to the event horizon.Even with a weaker dependence of size on wavelength,the light travel time scale at millimeter wavelengths is a few minutes, comparable to the shortest time scale observed at X-ray and NIR wavelengths.This coincidence suggests that the brightflares observed in at higher energies(7,5,6)are related to the submillimeter part of the spectrum and come from the vicinity of the black hole.The proximity of the millimeter emission indicates that emission at this and shorter wavelengths will be subject to strong light bending effects,providing a unique probe of strong-field general relativity(10,35).The size-wavelength relation also implies that the black hole mass must be containedwithin only a few Schwarzschild radii.Radio proper motion measurements require that Sgr A*must contain a significant fraction if not all of the compact dark mass found in the Galactic Center(36,37,38).Using conservatively only our7mm size and the lower limit of the Sgr A*mass of4×105M⊙,wefind that the mass density in Sgr A*has to be strictly aboveρ•>1.4×104M⊙AU−3.The dynamical lifetime of a cluster of objects with that density would be less than1000years,making Sgr A*the most convincing existing case for a massive black hole(39).References and Notes1.F.Melia,H.Falcke,Annual Rev.Astron.Astrophys.39,309(2001).2.A.M.Ghez,et al.,Astrophys.J.Lett.586,L127(2003).3.R.Sch¨o del,et al.,Astrophys.J.596,1015(2003).4.H.Falcke,et al.,Astrophys.J.499,731(1998).5.R.Genzel,et al.,Nature425,934(2003).6.A.M.Ghez,et al.,Astrophys.J.Lett.601,L159(2004).7.F.K.Baganoff,et al.,Nature413,45(2001).8.G.C.Bower,H.Falcke,R.J.Sault,D.C.Backer,Astrophys.J.571,843(2002).9.G.C.Bower,M.C.H.Wright,H.Falcke,D.C.Backer,Astrophys.J.588,331(2003).10.H.Falcke,F.Melia,E.Agol,Astrophys.J.Lett.528,L13(2000).11.W.Goss,R.Brown,K.Lo(2003).Astron.Nachr.,Vol.324,No.S1(2003),SpecialSupplement”The central300parsecs of the Milky Way”,Eds.A.Cotera,H.Falcke, T.R.Geballe,S.Markoff.zio,J.M.Cordes,Astrophys.J.505,715(1998).13.R.Narayan,J.Goodman,Mon.Not.R.Astron.Soc.238,963(1989).14.P.N.Wilkinson,R.Narayan,R.E.Spencer,Mon.Not.R.Astron.Soc.269,67(1994).15.K.M.Desai,A.L.Fey,Astrophys.J.Supp.133,395(2001).16.A.S.Trotter,J.M.Moran,L.F.Rodriguez,Astrophys.J.493,666(1998).17.K.Y.Lo,Z.Q.Shen,J.H.Zhao,P.T.P.Ho,Astrophys.J.Lett.508,L61(1998).18.We assume for Sgr A*a black hole mass of4×106M⊙and a distance of8.0kpc(2).The latter implies that0.1mas=0.8AU=1.1×1013cm.Together,these quantities imply a Schwarzschild radius R s=2GM/c2=1.2×1012cm=0.08AU=0.01mas.19.S.S.Doeleman,et al.,Astron.J.121,2610(2001).20.R.Herrnstein,J.-H.Zhao,G.C.Bower,W.M.Goss(2004).Astron.J.,in press.21.G.C.Bower,D.C.Backer,R.A.Sramek,Astrophys.J.558,127(2001).22.E.W.Greisen,Information Handling in Astronomy-Historical Vistas(2003),pp.109–+.23.A.E.E.Rogers,S.S.Doeleman,J.M.Moran,Astron.J.109,1391(1995).。

滤波器的英文介绍

滤波器的英文介绍

The basic principle of spread spectrum communicationSo-called spread spectrum communication, but simply indicates as follows: The "wide frequency communication is one intelligence transmission mode, its signal cabin holds the bandwidth far is bigger than passes on the information essentially the minimum bandwidth ; The frequency band expansion is completes through an independent code sequence, implements with the code and the modulation method, with passes on the information data to have nothing to do with; Uses the similar code in the receiving end to carry on the correlation synchronization receive, demodulation and recover passes on information data ".The wide frequency communication essential feature, is transmits the minimum bandwidth which the signal cabin takes (W) far to be bigger than (B) which primary information itself actual needs, its ratio is called the processing gain (Gp): In brief, we use the spread spectrum the wide band signal to transmit the information, is for enhance the communication the antijamming ability, namely under strong interferes the condition guaranteed the reliable security communicates. This is the spread spectrum communication basic thought and the theory basis.First, the main merit of wide frequency communications system* It’s easy to duplicate the frequency of use, raise the wireless frequency spectrum use factor* Strong anti-jamming, the error rate is low. Wide frequency communication when spatial transmission holds the bandwidth relative is wider, but the receiving end uses the correlation detection the means to solve expands, makes the useful wide band information signal to recover the narrow band signal, but non- needs the signal to expand the wide band signal, then extracts the useful signal through the narrow band filtering technology. This auspicious, regarding each kind of unwanted signal, because it in receives the end the non- relevance, after demodulation in the narrow band signal only has the very weak ingredient, the signal to noise ratio very high, therefore the anti-jamming is strong.* Good Privacy, is very small to each kind of narrow band communications system disturbance. Because the wide frequency signal expanded in the relative wider frequency band, in unit frequency band power very small, the signal is neglected in the noise, generally not easily was detected, but wants further to examine the signal the parameter (for example pseudo-random code sequence) difficultly, therefore said its privacy is good.* May implement a code minute site. The wide frequency communication enhanced the resistance to interference, the price takes the band width. But if many usersaltogether use this wideband, then may enhance the frequency band the use factor. Because has the wide frequency code sequence in the wide frequency communication the wide frequency modulation, fully uses between each kind of different code wide frequency code sequence the fine autocorrelation identity and the mutual correlation identity, carries on the solution in the receiving end using the correlation detection technology to expand, then in allocate may differentiate the different user for the different user code situation in the signal, extracts the useful signal. Like this many pair of users may simultaneously converse on the telephone in this frequency band but mutually does not disturb.* Anti- multi- diameters disturbance. In the wireless communication, since long ago, the multi- diameters disturbance throughout is one of questions which solves with difficulty. Uses the wide frequency code in the wide frequency communication the autocorrelation identity, extracts and separates the strongest useful signal in the receiving end from the multi- diameters signal, or the identical code sequence profile which comes many paths adds together the synthesis, all may the anti- multi- diameters disturbance function.Be different according to the spread spectrum mode, the existing wide frequency communications system may divide into following several kinds:* Direct sequence spread spectrum. The direct sequence spread spectrum (Direct Sequence Spread Spectrum) the work mode, is called straight expands (DSSS) the mode. The so-called direct sequence spread spectrum, is directly use the high rate wide frequency code sequence to expand the signal in the start the frequency spectrum. But in the receiving end, carries on the solution with the same wide frequency code sequence to expand, returns to original state the primitive information the stretch wide frequency signal.* Frequency-hopping (Frequency Hopping). Moreover one expansion signal frequency spectrum mode is called the frequency-hopping (FH - Frequency Hopping). The so-called frequency-hopping, compared with the accurate meaning is: Carries on the selection with the certain code sequence the multi- frequencies frequency-shift keying. In other words, carries on the frequency-shift keying modulation with the wide frequency code sequence, causes the carrier frequency unceasingly to jump, therefore is called the frequency-hopping.* Jumps when (Time Hopping). Is similar, jumps when (TH – Time Hopping) with the frequency-hopping is causes the transmitting message to jump in the time axis. First divides into the time axis many o'clock pieces. In does an in which time piece transmitting message carry on the check by the wide frequency code sequence. May jumps when the understanding be: Carries on selection with the certain code sequence many when pieces when moves the key modulation. Because used has very been narrow the very many time piece to transmit the signal, relatively mentioned, thesignal frequency spectrum also stretched.* Wide band linear frequency modulation (Chirp Modulation). The wide band linear frequency modulation work mode, is called the Chirp mode. If emanates radio frequency pulse signal in a cycle, its carrier spectrum frequency do change, then is called the linear frequency modulation.Second, direct sequence spread spectrum systemCompares with the general simulation or the digital communication system, the direct sequence spread spectrum in the information recognition and the demodulation, the radio frequency on frequency conversion and under the frequency conversion situation basic is same. Straight expands the communications system the main characteristic to lie in straight expands the signal the production, namely the wide frequency modulation and straight expands the signal the receive, namely the related solution expands.The wide frequency modulation is carries on the modulation with high rate PN code pulse sequence thus the expansion signal frequency spectrum. Usually uses the modulation mode is BPSK, the input signal and the PN code modulates in the balanced modulator outputs the stretch the wide frequency signal.Straight expands the system to carry on the modulation in the start with the PN code to expand the signal frequency spectrum. Is receiving the end generally to use the correlation detection or the matched filtering method solves expands. The so-called correlation detection, a simple metaphor is compares with the picture looks for the person. If wants to search the person in group of people which some is not acquainted with one another, the simplest valid method is in the hand has a Zhang person's picture, then with the picture one by one contrast, gets down like this, naturally can find some person. Same principle, when you want to examine the useful signal which needs, the valid method is in local produces a same signal, then with it with the signal contrast which receives, as desired similarity. In other words, is with the same signal which local produces with the signal which receives carries on the correlation operation, correlation function is biggest on the useful signal which most possibly is wants.The connected demodulation to be no doubt very good in the performance, but it needs to have the local PN code in the receiving end. This point sometimes brings many is not convenient. For example, the solution local signal and the receive signal synchronized question very is troublesome, but also cannot achieve real-time examines the useful signal. Because the matched filtering and the correlationdetection function in essentially is same, we may use the matched filter to demodulate t the straight -expand signal.The so-called matched filter, is a filter which matches with the signal, it can examines in the many kinds of signals or the disturbance with it match signal. This similarly is one kind of "looks for person" with the photograph the method. As for the video frequency rectangle pulse sequence that, the passive matched filter is on the tap delay line adds on the adder-accumulator.Third, frequency-hopping systemWe usually contact the wireless communications system all is the carrier frequency fixed communications system, like mobile phone and so on, therefore also is called as decides the frequency communication. This kind decides the frequency communications system, once will receive the disturbance on to cause the communication drop in quality, will be serious when will even cause the communication interrupt.Moreover in the enemy I in the duplex communication resistance, the enemy side attempt detects our communication frequency, in order to transmits to the interception the information content, or detected our telegraph is at position. Decides the frequency communications system to be easy to expose the goal also easy to intercept, by now, used the frequency-hopping communications quite to be covert with difficulty is also intercepted.Therefore, the frequency-hopping communications has the antijamming, the anti- interception ability, and can do frequency spectrum resource sharing. Therefore the frequency-hopping communications has displayed the huge superiority in the current modernized electronic warfare. Moreover, the frequency-hopping communications also applies in the civil communication by between the anti- decline, the anti- multi- diameters, the anti- network disturbs and raises the frequency spectrum use factor.In order to does not let the enemy side know we communicate the use frequency, needs frequently to change the carrier frequency, namely carries on the jump to the carrier frequency, in the frequency-hopping communications the carrier frequency change rule, is called the frequency-hopping design.The frequency-hopping signal receive, its process with decides the frequency to be similar. In order to after guarantee the mixing obtains the intermediate frequency signal, the requirement frequency synthesizer output frequency must outdo an intermediate frequency compared to the external signal. Because the external signal-carrier frequency is the jump, then requires the frequency which the local frequency synthesizer outputs also along with the external signal jump rule to jump, like this can obtain a fixed center quite signal through the mixing.The frequency-hopping is the frequency-hopping system key component, but frequency-hopping synchronization is the frequency-hopping system core technology. Frequency-hopping system synchronization including following several contents:* Receives the frequency-hopping design which the end and the start produces to be same, namely has the same frequency-hopping rule.* Receives, the start jump frequency should guarantee produces the fixed intermediate frequency signal in the receiving end, namely the jump carrier frequency with receives this locality which the end produces to jump the frequency to differ an intermediate frequency.* Frequency jump beginning and end time in time synchronization, namely synchronized jump, or phase coincidence.* When transmission numerical information, but also should achieve the frame synchronization and position synchronization.Fourth, PN codeThe PN code also calls the pseudo-random sequence. It has the approximate random sequence (noise) nature, but also can (cycle) produce according to the certain rule with the copying sequence. Because the random sequence is only can produce but cannot copy, therefore called it is the "pseudo" random sequence. The commonly used pseudo-random sequence has the m sequence, the M sequence and R –the S sequence.The msequencer from the belt feedback m level shift register framing,feeds back from certain levels after mold two Canada to the first level. It produces the sequence greatest length (cycle) is the 2n – 1 bit, altogether has 2m to plant the different state, one kind is entire "0" state. Only has when the feedback logic satisfies some kind of condition, the shift register outputs the sequence length is the 2n - 1 bit, achieves the greatest length. Otherwise produces the sequence could not achieve 2n - 1 bit of such is long. Therefore also is called the m sequence the greatest length linearity shift register sequence. Also is called the biggest shift register sequence.If in the feedback logic operation includes the multiplication operation or other logic operations, then is called as the nonlinear feedback logic. The sequencer frame which by the nonlinear feedback logic and the shift register can have the greatest length sequence, is called the greatest length non-linearity shift register sequence, or is called the M sequence, the M sequence greatest length is 2n.。

Headphone with Active Noise Control using Analog Adaptive Filters

Headphone with Active Noise Control using Analog Adaptive Filters

Headphone with Active Noise Control using AnalogAdaptive FiltersAlex Jos´e Veloso a,V´ıtor Heloiz Nascimento ba,b Electronic Systems Eng.Dept,Escola Polit´e cnica,Universidade de S˜a o Paulo, Av.Prof Luciano Gualberto,trav.3,n158,05508-90,S˜a o Paulo,SP,Brazila alex.veloso@p.br,b vitor@p.brAbstract Traditional methods of acoustic noise control are based on materials that absorb sound waves.These methods employ relatively bulky materials,and present some deficiencies that become more important at low frequencies.An active noise control (ANC)system may be used to efficiently reduce noise with frequencies below500Hz. Most ANC systems for headphones are based onfixed analog controllers with feedback, that usually reduce the noise up to15dB for frequencies below500Hz,independently of the noise spectrum.Changes in the environment,in the positioning of the phone, in the ageing of the components,and different users,modify the transfer function of the secondary path,reducing the system performance.Given this limitation,the use of feedback digital adaptivefilters has been investigated.The main factor limiting the performance of feedback digital controllers applied to headphones is the group delay of the secondary path.The group delay of an analog system is smaller than that introduced by a digital system(due to the anti-aliasing,reconstructionfilters,analog-to-digital and digital-to-analog converters used for digital processing).In order to reduce the contribution of the group delay due to the electrical system and to improve the correlation between the reference signal and the noise,we propose the use of a headphone with active noise control using analog adaptivefilters with a FIR-like structure,using instead of a delay line a cascade of gammafilters.1.INTRODUCTIONActive noise control follows the principle of the destructive wave interference,reducing an unwanted acoustic noise generated by a primary source through an anti-noise produced by a secondary source.Figures1and2show a headphone with feedback ANC and its electrical block diagram[1]using digital adaptivefilters,where d(n)is the primarynoise,e(n)is the residual noise measured by the error sensor,y (n)is the anti-noiseproduced by adaptivefilter W(z),x(n)is the estimate of the primary noise obtained asx(n)≡ˆd(n)=e(n)+K−1k=0ˆs k y(n−k),(1)whereˆs k is an approximation to the impulse response of the secondary path transfer function S(z).The secondary path is composed by the transfer functions of the error microphone,the pre-amplifier,the anti-aliasingfilter,the analog-to-digital converter (ADC),the digital-to-analog converter(DAC),the reconstructionfilter,the power am-plifier,the loudspeaker,and the acoustic path from loudspeaker to error microphone. The feedback ANC system may be viewed as a feedforward ANC system able to create its own reference signal.Figure1:Headphone with ANC system Figure2:Feedback ANC systemIn order that y (n)exactly cancels d(n),the adaptivefilter W(z)would need to model the inverse transfer function of S(z),that isW(z)=1S(z).(2)Real-time ANC requires that the processing time of the samples be less than than the sampling period,and that W(z)be causal.Unfortunately,in general S(z)is not minimum-phase,so its inverse is non-causal,and hence unrealizable.If the noise were periodical(such as a pure tone),this would not be a problem,but for broadband or white noise W(z)would be unable to produce an exact anti-noise to completely cancel the primary noise.In order to have a causal W(z)close to the ideal solution,the group delay of S(z)must be decreased as much as possible.2.SECONDARY PATHIn this section we present the characteristic delays introduced by electrical,electroa-coustic,and acoustic transfer functions for headphones with ANC.The ADC,DAC,and two low-passfilters(LPF)of the digital secondary path generate an eletrical delayτd given approximately byτd=1f s 1+3.2.n8,(3)where f s is the sampling frequency,and n is the number of poles of each LPF[2].The delay of the acoustic path is given byτa=rc,(4)where r is the distance between the loudspeaker and microphone,and c is the speed of sound.The condenser microphone we used has a negligible delay.The total secondary path delay can be express asτT=τd+τa+τl,(5) whereτl is the delay,due to loudspeaker.If instead of the digital system,an analog system were applied,the group delay of the secondary path would be decreased toτT=τa+τl,(6) due to not use of the LPF´s and converters.It should be emphasised thatτd could be reduced in two ways.Thefirst would be to reduce the number of poles,but there is a lower limit to thefilter order that still avoids aliasing.Another possibility would be to increase the sampling frequency,but at a high sampling rate the adaptive control algorithm would have some problems that we discuss in the next section.3.ADAPTIVE CONTROLIn order that the secondary noise source cancels the primary noise,we need both a reference signal correlated with the primary noise source and afilter able to transform the reference signal into an anti-noise.Changes in the transfer functions due to changes in the environment,in the positioning of the error microphone,the ageing of the compo-nents,and different users,degrade the performance offixed(non-adaptive)controllers. Adaptivefilters can be realizated using FIR or IIR structures.Unlike IIR adaptive filters,FIR adaptivefilters have the advantage of having an unimodal error surface. FIR adaptivefilters are also simpler to implement,and their convergence is usually faster.However,for systems with long impulse responses,IIR structures may lead to better modeling.In general,an IIRfilter will require less coefficients for a given level of performance than an FIRfilter.In the case of ANC systems with a single microphone,the secondary path has a relatively short impulse response,so the simpler FIR structure should be the best choice.The least-mean-square(LMS)algorithm for a FIR digital adaptivefilter W(z)of order M is[3]given byw m(n+1)=w m(n)+µx (n)e(n),(7) where w m(n)is a coefficient of the vector W(n),x m(n)is afiltered version of x m(n) express byx m(n)=K−1k=0ˆs k x m(n−k),(8)m=0,1,···,M−1,andµis the digital adaptive gain whose upper bound is given byµ<231Mσ2x,(9)whereσ2x is the variance of x (n).The performance of an ANC system for non-periodical noise depends on group delay introduced by the secondary path,which in principle favours the use of the analog system.The group delay of a digital system could be diminished if a high sampling frequency were used.The advantage of increasing the sampling frequency is the possibility of reducing the order of the LPF,and/or of keeping the group delay low in the frequency range of interest(up to approximately500Hz)by pushing up the LPF’s cut-offfrequency. However,a higher sampling rate means less time to compute every new output,and also requires the use of more parameters for the digital adaptivefilter.Adaptivefilters using the LMS algorithm have a misadjustment[3]given byσ2misad=σ2o 1+µMσ2x 2 .(10)We can note in(10)that a high value of M results in higher misadjustment.Increasing M would also require a reduction in the speed convergence,since the condition for stable performance of thefilter must be as(9).In order to implement analog adaptivefilters with a FIR structure without the need of analog delay lines(all-zero structure),cascades of all-pass or low-passfilters can be used.With all-passfilters(such as Laguerre or Kautzfilters),the estimation problem is orthogonal,and better-conditioned[4],however,for the sake of simplicity we used a cascade of low-pass(gamma)filters[5].The use of a cascade offilters instead of a delay line,may also lead to a smaller number of parameters to estimate,if the(fixed)low-pass filter parameters are properly chosen.The block diagram of the digital and analog adaptivefilters used to minimize the square error between y (n)and d(n)are shown infigures3and4.Analog adaptivefilters implemented via gammafilters have a structure similar to that of a transversalfilter,with the delay line replaced by a cascade of low-passfilters withtransfer functionH(s)=a0s+w0.(11)Figure3:Digital adaptivefilterFigure4:Analog adaptivefilterThefiltered-X LMS algorithm for an analog adaptivefilter W(s)of order M is[6,7] given byd w m(t)d t=ρx m(t)e(t)dt,(12) where w m(t)is the m-th coefficient of the vector W(t)and x m(t)is afiltered version of x m(t),expressed by the convolution operation belowx m(t)=ˆs(t)∗x m(t)= t0ˆs(ϕ)x m(t−ϕ),(13)Unlike the case of digital adaptivefilters,the analog adaptive gainρcan be arbitrarily large without affecting the stability of the algorithm,but a faster convergence increases the misadjustment[8].4.RESULTSThe transfer function of the digital secondary path was estimated through the off-line method[9],with a pseudo-noise as input(a maximum-length sequence)[10].A sucessive-approximation analog/digital converter was used,due to its group delay that is,in general,significantly lower than the group delay introduced by sigma-delta converters. In our experiments,we used f s=8000Hz,n=7,r=5mm,and c=343m/s,that result inτd=0.781ms,andτa=0.015ms.The loudspeaker has a significative delay, especially at low frequencies;for100Hz the delay wasτl=2.055ms,and for500Hz,τl=0.957ms.The total digital secondary path delay wasτT=2.851ms for100Hz andτT=1.753ms for500Hz.In order to estimate the analog secondary path a sine sweep(50to4000kHz)was used.With the frequency response,the Matlab function invfreqs was used tofind an approximate transfer function S(s).For100Hz the total delay wasτT=2.070ms,and for500HzτT=0.972ms.Figure5shows the magnitude and phase of the digital(solid line)and analog(dashed line)secondary paths.We analyzed the performance of the digital and analog systems through computer si-mulations.A signal composed by sinusoids with frequencies of200Hz,300Hz,and40005001000150020002500300035004000−70−60−50−40−30−20Frequency (Hz)M a g n i t u d e (d B )Secondary pathFrequency (Hz)P h a s e (d e g r e e s )Figure 5:Digital secondary path (solid line)and analog secondary path (dashed line)Hz,added to a white noise with zero mean and unity variance band-limited between 120Hz and 600Hz,was used as primary noise source.The digital adaptive filter W (z )used to model the inverse of S (z )was an FIR filter with 128-taps and µ=0.1.For the analog system,W (s )was implemented with a 10-taps gamma filter cascade and ρ=104(these values of parameters µand ρwere chosen by trial and error,and achieved the best results for each filter).Since the variance of the filtered version of x (n )is low,the parameters µand ρresult in high values.The gamma filter was implemented with the transfer functionH (s )=2512s +2512.Figures 6and 7show the noise at the error microphone without (solid line)and with (dashed line)the ANC system.For the digital ANC system,the primary noise was reduced by aproximadely 10dB,while the feedback ANC system implemented with the analog adaptive filter achieved a reduction of about 20dB.5.CONCLUSIONWe proposed a method to improve the performance of ANC headphones,by using an analog adaptive filter.This approach allows the use of adaptive filters with a reduced total delay in the ANC system’s secondary path,compared with digital filters,thus keeping the advantages of both analog (low group delay)and digital (adaptiveness)ANC systems.We have tested the performance of our analog adaptive filter through simulations,using transfer functions measured in a comercial headphone.Our simula-tions show the potential performance improvement that can be achieved by using analog adaptive filters.In future works we intend to implement our system,and compare its results with digitalTime (s)N o i s e x E r r o r Figure 6:Feedback ANC system with digital adaptive filter without (solid line)and with (dashed line)using the ANC systemFigure 7:Feedback ANC system with analog adaptive filter without (solid line)and with (dashed line)using the ANC systemadaptive and fixed analog headphone ANC systems.We also intend to investigate which are the best choices for the gamma filter parameters,and experiment with orthogonal filters,such as Laguerre and Kautz filters [4].ACKNOWLEDGEMENTSThis work was supported in part by the S˜a o Paulo State Research Council (FAPESP).REFERENCES[1]T.G.Tsuei,A.Srinivasa,S.M.Kuo,“An Adaptive Feedback Active Noise ControlSystem”,Proceedings of the IEEE,249-254,2000[2]P.A.Nelson,S.J.Elliott,Active Control of Sound,Academic Press,San Diego,1992[3]A.H.Sayed,Fundamentals of Adaptive Filtering,Wiley-Interscience,NY,2003[4]B.Ninness,F.Gustafsson,“A Unifying Construction of Orthonormal Bases for SystemIdentification”,tac,515-521,1997[5]J.Juan,J.G.Harris,J.C.Principe,“Analog Hardware Implementation of AdaptiveFilter Structures”,International Conference on Neural Networks,916-921,1997[6]P.Ioannou,J.Sun,Robust Adaptive Control,PTR Prentice-Hall,NJ,1996[7]V.H.Nascimento,Y.S.Provase,“Redu¸c˜a o de Distor¸c˜a o Harmˆo nica em Amplificadorespor meio de Filtros Adaptativos Anal´o gicos”,XX Simp´o sio Brasileiro de Telecomu-nica¸c˜o es,2003(in Portuguese)[8]S.Karni,G.Zeng,“The Analysis of the Continuous-Time LMS Algorithm”IEEE Tran-sactions on acoustic,speech,and signal processing,595-597,1989[9]S.M.Kuo,D.R.Morgan,Active Noise Control Systems:Algorithms and DSP Imple-mentations,John Wiley&Sons,Inc.,NY,1996[10]P.M.S.Burt,“Measuring Acoustic Responses with Maximum-Length Sequences”,Pro-ceedings.Piscataway:IEEE,1998[11]S.M.Kuo,D.R.Morgan,“Active Noise Control:A tutorial review”,Proceedings of theIEEE,943-973,1999[12]C.H.Hansen,S.D.Snyder,Active control of noise and vibration,E&FN Spon,London,1997[13]W.S.Gan,S.M.Kuo,“Integrated Active Noise Control Communication Headsets”,Proc.IEEE International Symposium on Circuits and Systems,353-356,2003[14]W.S.Gan,S.M.Kuo,“An Integrated Audio and Active Noise Control Headsets”,IEEETransactions on Consumer Eletronics,242-247,2002[15]S.M.Kuo,X.Kong,W.S.Gan,“Applications of Adaptive Feedback Active Noise ControlSystem”,IEEE Transactions on Control Systems Technology,216-220,2003[16]L.E.Kinsler,A.R.Freg.,Fundamentals of Acoustics,John Wiley&Sons,Inc.,NY,2000[17]S.J.Elliott,T.J.Sutton,“Performance of Feedforward and Feedback Systems for ActiveControl”,IEEE Transactions on Speech,Audio Processing,214-223,1996[18]J.J.Shynk,“Adaptive IIR Filtering”,IEEE ASSP Magazine,4-21,1989[19]M.Pawelczyk,“Analogue active noise control”,Applied Acoustics,1193-1213,2002。

一种改进的光载波抑制产生光毫米波的方法

一种改进的光载波抑制产生光毫米波的方法

一种改进的光载波抑制产生光毫米波的方法陈罗湘;黄诚;陈林【摘要】为了延长光毫米波的传输距离,提出了一种改进的光载波抑制产生光毫米波的方法.在中心站采用马赫-曾德尔调制器将射频信号调制到光载波上产生光载波抑制调制光信号,再将产生光信号的2个边带分离,将2.5Gbit/s数据信号调制到其中1个边带上,再与未调信号耦合后产生光毫米波并通过光纤传送至基站.在基站中通过光电转换器产生电毫米波.从理论上分析了这种光毫米波的传输特性并通过实验验证了光毫米波在光纤中可以传输40km.仿真和实验结果表明,这种方式产生的光毫米波具有很好的抗色散能力,延长了传输距离.【期刊名称】《激光技术》【年(卷),期】2008(032)006【总页数】4页(P659-662)【关键词】光通信;光纤无线通信系统;光毫米波产生;光载波抑制【作者】陈罗湘;黄诚;陈林【作者单位】湘潭职业技术学院,信息工程系,湘潭,411102;湖南大学,计算机与通信学院,长沙,410082;湖南大学,计算机与通信学院,长沙,410082;湖南大学,计算机与通信学院,长沙,410082【正文语种】中文【中图分类】TN929.11引言光纤无线通信系统(radio over fiber,ROF)将成为未来超宽带无线接入的最理想的通信方式,人们已对ROF研究了多年[1-13]。

光毫米波产生方法是降低ROF系统代价的最关键的技术之一。

迄今为止,已提出的光毫米波的产生的方法有3种:直接强度调制、外部强度调制和远程外差技术[1-13]。

基于外部调制器的光毫米波产生方案具有较高的可靠性,可降低代价,因而最有可能成为ROF系统中产生光毫米波的首选技术[5]。

采用外部调制器产生光毫米波的方法有3种:单边带调制(single sideband,SSB),双边带调制(double sideband,DSB)以及光载波抑制(optical carrier supression,OCS)。

关于滤波器专业英语作文

关于滤波器专业英语作文

关于滤波器专业英语作文Introduction。

Filters are electronic circuits that selectively allow certain frequencies to pass through while attenuating others. They are used in a wide range of applications, from audio and video processing to power supplies and communication systems. In this essay, we will explore the different types of filters, their characteristics, andtheir applications.Types of Filters。

Filters can be classified into two main categories: passive filters and active filters. Passive filters use only passive components such as resistors, capacitors, and inductors to filter out unwanted frequencies. Active filters, on the other hand, use active components such as transistors, operational amplifiers, and digital signal processors to amplify, attenuate, or shift frequencies.Passive Filters。

Passive filters are the simplest type of filters and are widely used in audio and video processing. They can be further classified into three types: low-pass filters, high-pass filters, and band-pass filters.Low-pass filters allow low frequencies to pass through while attenuating high frequencies. They are commonly used in audio systems to filter out noise and unwanted signals. High-pass filters, on the other hand, allow high frequencies to pass through while attenuating low frequencies. They are used in applications such as radio communication and audio equalization.Band-pass filters allow a range of frequencies to pass through while attenuating frequencies outside the range. They are used in applications such as radio frequency (RF) communication and audio processing.Active Filters。

Silicon Labs EFR32MG 2.4 GHz 19.5 dBm 无线模组板参考手册说明书

Silicon Labs EFR32MG 2.4 GHz 19.5 dBm 无线模组板参考手册说明书

EFR32MG 2.4 GHz 19.5 dBm Radio BoardBRD4151A Reference Manualance, low energy wireless solution integrated into a small formfactor package.By combining a high performance 2.4 GHz RF transceiver with an energy efficient 32-bitMCU, the family provides designers the ultimate in flexibility with a family of pin-compati-ble devices that scale from 128/256 kB of flash and 16/32 kB of RAM. The ultra-lowpower operating modes and fast wake-up times of the Silicon Labs energy friendly 32-bit MCUs, combined with the low transmit and receive power consumption of the 2.4GHz radio, result in a solution optimized for battery powered applications.To develop and/or evaluate the EFR32 Mighty Gecko, the EFR32MG Radio Board canbe connected to the Wireless Starter Kit Mainboard to get access to display, buttons andadditional features from Expansion Boards.Introduction 1. IntroductionThe EFR32 Mighty Gecko Radio Boards provide a development platform (together with the Wireless Starter Kit Mainboard) for the Silicon Labs EFR32 Mighty Gecko Wireless System on Chips and serve as reference designs for the matching network of the RF inter-face.The BRD4151A Radio Board is designed to operate in the 2400-2483.5 MHz band with the RF matching network optimized to operate with 19.5 dBm output power.To develop and/or evaluate the EFR32 Mighty Gecko, the BRD4151A Radio Board can be connected to the Wireless Starter Kit Main-board to get access to display, buttons and additional features from Expansion Boards and also to evaluate the performance of the RF interface.2. Radio Board Connector2.1 IntroductionThe board-to-board connector scheme allows access to all EFR32MG1 GPIO pins as well as the RESETn signal. For more information on the functions of the available pin functions, see the EFR32MG1 data sheet.2.2 Radio Board Connector Pin AssociationsThe figure below shows the pin mapping on the connector to the radio pins and their function on the Wireless Starter Kit Mainboard.GND F9 / PA3 / VCOM.#RTS_#CS 3v3UIF_BUTTON1 / PF7 / P36P200Upper RowNC / P38NC / P40NC / P42NC / P44DEBUG.TMS_SWDIO / PF1 / F0DISP_ENABLE / PD15 / F14UIF_BUTTON0 / PF6 / F12DISP_EXTCOMIN / PD13 / F10VCOM.#CTS_SCLK / PA2 / F8#RESET / F4DEBUG.TDO_SWO / PF2 / F2DISP_SI / PC6 / F16VCOM.TX_MOSI / PA0 / F6PTI.DATA / PB12 / F20DISP_EXTCOMIN / PD13 / F18USB_VBUS5VBoard ID SCLGND Board ID SDAUSB_VREG F7 / PA1 / VCOM.RX_MISO F5 / PA5 / VCOM_ENABLE F3 / PF3 / DEBUG.TDI F1 / PF0 / DEBUG.TCK_SWCLK P45 / NC P43 / NCP41 / NCP39 / NCP37 / High / SENSOR_ENABLEF11 / PF5 / UIF_LED1F13 / PF7 / UIF_BUTTON1F15 / PC8 / DISP_SCLK F17 / PD14 / DISP_SCS F19 / PB13 / PTI.SYNC F21 / PB11 / PTI.CLK GNDVMCU_INVCOM.#CTS_SCLK / PA2 / P0P201Lower RowVCOM.#RTS_#CS / PA3 / P2PD10 / P4PD11 / P6GND VRF_INP35 / PD15 / DISP_ENABLE P7 / PC9P5 / PC8 / DISP_SCLK P3 / PC7P1 / PC6 / DISP_SI P33 / PD14 / DISP_SCSP31 / PD13 / DISP_EXTCOMIN P29 / NCP27 / NC P25 / NC P23 / NC P21 / NC P19 / NC P17 / NC P15 / NC P13 / PC11P11 / PA1 / VCOM.RX_MISO P9 / PA0 / VCOM.TX_MOSI UIF_BUTTON0 / PF6 / P34UIF_LED1 / PF5 / P32UIF_LED0 / PF4 / P30DEBUG.TDO_SWO / PF2 / P28DEBUG.TMS_SWDIO / PF1 / P26DEBUG.TCK_SWCLK / PF0 / P24PTI.SYNC / PB13 / P22PTI.DATA / PB12 / P20PTI.CLK / PB11 / P18VCOM_ENABLE / PA5 / P16PA4 / P14PC10 / P12DEBUG.TDI / PF3 / P10PD12 / P8Figure 2.1. BRD4151A Radio Board Connector Pin MappingRadio Board Connector3. Radio Board Block Summary3.1 IntroductionThis section gives a short introduction to the blocks of the BRD4151A Radio Board.3.2 Radio Board Block DiagramThe block diagram of the EFR32MG Radio Board is shown in the figure below.Figure 3.1. BRD4151A Block Diagram3.3 Radio Board Block Description3.3.1 Wireless MCUThe BRD4151A EFR32 Mighty Gecko Radio Board incorporates an EFR32MG1P232F256GM48 Wireless System on Chip featuring 32-bit Cortex-M4 with FPU core, 256 kB of flash memory and 32 kB of RAM and a 2.4 GHz band transceiver with output power up to 19.5 dBm. For additional information on the EFR32MG1P232F256GM48, refer to the EFR32MG1 Data Sheet.3.3.2 LF Crystal Oscillator (LFXO)The BRD4151A Radio Board has a 32.768 kHz crystal mounted.3.3.3 HF Crystal Oscillator (HFXO)The BRD4151A Radio Board has a 38.4 MHz crystal mounted.3.3.4 Matching Network for 2.4 GHzThe BRD4151A Radio Board incorporates a 2.4 GHz matching network which connects the 2.4 GHz TRX pin of the EFR32MG1 to the one on-board printed Inverted-F antenna. The component values were optimized for the 2.4 GHz band RF performace and current con-sumption with 19.5 dBm output power.For detailed description of the matching network, see Chapter 4.2.1 Description of the 2.4 GHz RF Matching.| Smart. Connected. Energy-friendly.Rev. 1.7 | 33.3.5 Inverted-F AntennaThe BRD4151A Radio Board includes a printed Inverted-F antenna (IFA) tuned to have close to 50 Ohm impedance at the 2.4 GHz band.For detailed description of the antenna see Chapter 4.5 Inverted-F Antenna.3.3.6 UFL ConnectorTo be able to perform conducted measurements, Silicon Labs added an UFL connector to the Radio Board. The connector allows an external 50 Ohm cable or antenna to be connected during design verification or testing.Note: By default the output of the matching network is connected to the printed Inverted-F antenna by a series component. It can be connected to the UFL connector as well through a series 0 Ohm resistor which is not mounted by default. For conducted measurements through the UFL connector the series component to the antenna should be removed and the 0 Ohm resistor should be mounted (see Chapter 4.2 Schematic of the RF Matching Network for further details).3.3.7 Radio Board ConnectorsTwo dual-row, 0.05” pitch polarized connectors make up the EFR32MG Radio Board interface to the Wireless Starter Kit Mainboard. For more information on the pin mapping between the EFR32MG1P232F256GM48 and the Radio Board Connector, refer to Chapter 2.2 Radio Board Connector Pin Associations.4. RF Section4.1 IntroductionThis section gives a short introduction to the RF section of the BRD4151A.4.2 Schematic of the RF Matching NetworkThe schematic of the RF section of the BRD4151A Radio Board is shown in the following figure.U1BPath Inverted-F Antenna2.4 GHz Matching Figure 4.1. Schematic of the RF Section of the BRD4151A4.2.1 Description of the 2.4 GHz RF MatchingThe 2.4 GHz matching connects the 2G4RF_IOP pin to the on-board printed Inverted-F Antenna. The 2G4RF_ION pin is connected to ground. For higher output powers (13 dBm and above) beside the impedance matching circuitry it is recommended to use additional harmonic filtering as well at the RF output. The targeted output power of the BRD4151A board is 19.5 dBm. As a result, the RF output of the IC is connected to the antenna through a four-element impedance matching and harmonic filter circuitry.For conducted measurements the output of the matching network can also be connected to the UFL connector by relocating the series R1 resistor (0 Ohm) to the R2 resistor position between the output of the matching and the UFL connector.4.3 RF Section Power SupplyOn the BRD4151A Radio Board the supply pin of the RF Analog Power (RFVDD) is connected directly ot the output of the on-chip DC-DC converter while the supply for the 2.4 GHz PA (PAVDD) is provided directly by the mainboard. This way, by default, the DC-DC converter provides 1.8 V for the RF analog section, the mainboard provides 3.3 V for the PA (for details, see the schematic of the BRD4151A).4.4 Bill of Materials for the 2.4 GHz MatchingThe Bill of Materials of the 2.4 GHz matching network of the BRD4151A Radio Board is shown in the following table.Table 4.1. Bill of Materials for the BRD4151A 2.4 GHz 19.5 dBm RF Matching Network | Smart. Connected. Energy-friendly.Rev. 1.7 | 54.5 Inverted-F AntennaThe BRD4151A Radio Board includes an on-board printed Inverted-F Antenna tuned for the 2.4 GHz band. Due to the design restric-tions of the Radio Board the input of the antenna and the output of the matching network can't be placed directly next to each other. Therefore, a 50 Ohm transmission line was necessary to connect them. The resulting impedance and reflection measured at the output of the matcing network are shown in the following figure. As it can be observed the impedance is close to 50 Ohm (the reflection is better than -10 dB) for the entire 2.4 GHz band.Figure 4.2. Impedance and Reflection of the Inverted-F Antenna of the BRD4151A| Smart. Connected. Energy-friendly.Rev. 1.7 | 65. Mechanical DetailsThe BRD4151A EFR32 Mighty Gecko Radio Board is illustrated in the figures below.45 mmFigure 5.1. BRD4151A Top View5 mm ConnectorConnectorFigure 5.2. BRD4151A Bottom ViewMechanical DetailsRev. 1.7 | 7EMC Compliance 6. EMC Compliance6.1 IntroductionCompliance of the fundamental and harmonic levels is tested against the following standards:• 2.4 GHz:•ETSI EN 300-328•FCC 15.2476.2 EMC Regulations for 2.4 GHz6.2.1 ETSI EN 300-328 Emission Limits for the 2400-2483.5 MHz BandBased on ETSI EN 300-328 the allowed maximum fundamental power for the 2400-2483.5 MHz band is 20 dBm EIRP. For the unwan-ted emissions in the 1 GHz to 12.75 GHz domain the specified limit is -30 dBm EIRP.6.2.2 FCC15.247 Emission Limits for the 2400-2483.5 MHz BandFCC 15.247 allows conducted output power up to 1 Watt (30 dBm) in the 2400-2483.5 MHz band. For spurious emmissions the limit is -20 dBc based on either conducted or radiated measurement, if the emission is not in a restricted band. The restricted bands are speci-fied in FCC 15.205. In these bands the spurious emission levels must meet the levels set out in FCC 15.209. In the range from 960 MHz to the frequency of the 5th harmonic it is defined as 0.5 mV/m at 3 m distance (equals to -41.2 dBm in EIRP).Additionally, for spurious frequencies above 1 GHz, FCC 15.35 allows duty-cycle relaxation to the regulatory limits. For the EmberZNet PRO the relaxation is 3.6 dB. Therefore, the -41.2 dBm limit can be modified to -37.6 dBm.If operating in the 2400-2483.5 MHz band the 2nd, 3rd and 5th harmonics can fall into restricted bands. As a result, for those the -37.6 dBm limit should be applied. For the 4th harmonic the -20 dBc limit should be applied.6.2.3 Applied Emission Limits for the 2.4 GHz BandThe above ETSI limits are applied both for conducted and radiated measurements.The FCC restricted band limits are radiated limits only. Besides that, Silicon Labs applies those to the conducted spectrum i.e., it is assumed that, in case of a custom board, an antenna is used which has 0 dB gain at the fundamental and the harmonic frequencies. In that theoretical case, based on the conducted measurement, the compliance with the radiated limits can be estimated.The overall applied limits are shown in the table below.Table 6.1. Applied Limits for Spurious Emissions for the 2.4 GHz Band | Smart. Connected. Energy-friendly.Rev. 1.7 | 87. RF Performance7.1 Conducted Power MeasurementsDuring measurements, the EFR32MG Radio Board was attached to a Wireless Starter Kit Mainboard which was supplied by USB. The voltage supply for the Radio Board was 3.3 V.7.1.1 Conducted Measurements in the 2.4 GHz bandThe BRD4151A board was connected directly to a Spectrum Analyzer through its UFL connector (the R1 resistor (0 Ohm) was removed and a 0 Ohm resistor was soldered to the R2 resistor position). During measurements, the voltage supply for the board was 3.3 V provi-ded by the mainboard. The supply for the radio (RFVDD) was 1.8 V provided by the on-chip DC-DC converter, the supply for the power amplifier (PAVDD) was 3.3 V (for details, see the schematic of the BRD4151A). The transceiver was operated in continuous carrier transmission mode. The output power of the radio was set to the maximum level.The typical output spectrum is shown in the following figure.Figure 7.1. Typical Output Spectrum of the BRD4151AAs it can be observed, the fundamental is slightly lower than 19.5 dBm and the strongest unwanted emission is the double-frequency harmonic and it is under the -37.6 dBm applied limit.Note: The conducted measurement is performed by connecting the on-board UFL connector to a Spectrum Analyzer through an SMA Conversion Adapter (P/N: HRMJ-U.FLP(40)). This connection itself introduces approximately a 0.3 dB insertion loss.RF PerformanceRev. 1.7 | 97.2 Radiated Power MeasurementsDuring measurements, the EFR32MG Radio Board was attached to a Wireless Starter Kit Mainboard which was supplied by USB. The voltage supply for the Radio Board was 3.3 V. The radiated power was measured in an antenna chamber by rotating the DUT 360degrees with horizontal and vertical reference antenna polarizations in the XY , XZ and YZ cuts. The measurement axes are shown inthe figure below.Figure 7.2. DUT: Radio Board with the Wireless Starter Kit Mainboard (Illustration)Note: The radiated measurement results presented in this document were recorded in an unlicensed antenna chamber. Also the radi-ated power levels may change depending on the actual application (PCB size, used antenna, and so on). Therefore, the absolute levels and margins of the final application are recommended to be verified in a licensed EMC testhouse.7.2.1 Radiated Measurements in the 2.4 GHz bandFor the transmitter antenna, the on-board printed Inverted-F antenna of the BRD4151A board was used (the R1 resistor (0 Ohm) was mounted). During the measurements the board was attached to a Wireless Starter Kit Mainboard (BRD4001 (Rev. A02) ) which was supplied through USB. During measurements, the voltage supply for the board was 3.3 V provided by the mainboard. The supply for the radio (RFVDD) was 1.8 V provided by the on-chip DC-DC converter, the supply for the power amplifier (PAVDD) was 3.3 V (for details, see the schematic of the BRD4151A). The transceiver was operated in continuous carrier transmission mode. The output power of the radio was set to the maximum level.The results are shown in the table below.Table 7.1. Maximums of the Measured Radiated Powers of BRD4151AAs it can be observed, thanks to the high gain of the Inverted-F antenna, the level of the fundamental is higher than 19.5 dBm. The strongest harmonic is the double-frequency one but its level is under -45 dBm.RF PerformanceEMC Compliance Recommendations 8. EMC Compliance Recommendations8.1 Recommendations for 2.4 GHz ETSI EN 300-328 complianceAs it was shown in the previous chapter, the radiated power of the fundamental of the BRD4151A EFR32 Mighty Gecko Radio Board complies with the 20 dBm limit of the ETSI EN 300-328 in case of the conducted measurement but due to the high antenna gain the radiated power is higher than the limit by 2 dB. In order to comply, the output power should be reduced (with different antennas, de-pending on the gain of the used antenna, the necessary reduction can be different). The harmonic emissions are under the -30 dBm limit. Although the BRD4151A Radio Board has an option for mounting a shielding can, that is not required for the compliance.8.2 Recommendations for 2.4 GHz FCC 15.247 complianceAs it was shown in the previous chapter, the radiated power of the fundamental of the BRD4151A EFR32 Mighty Gecko Radio Board complies with the 30 dBm limit of the FCC 15.247. The harmonic emissions are under the -37.6 dBm applied limit both in case of the conducted and the radiated measurements. Although the BRD4151A Radio Board has an option for mounting a shielding can, that is not required for the compliance.Board Revisions 9. Board RevisionsTable 9.1. BRD4151A Radio Board RevisionsNote: The silkscreen marking on the board (e.g., PCBxxxx A00) denotes the revision of the PCB. The revision of the actual Radio Board can be read from the on-board EEPROM.Errata 10. ErrataTable 10.1. BRD4151A Radio Board ErrataDocument Revision History 11. Document Revision HistoryRevision 1.72016-11-20Minor editorial updates.Revision 1.62016-10-31Corrected error in radio board connector pinout diagram.Revision 1.52016-05-24Updating Board Revisions content. Fixing Errata description.Revision 1.42016-05-05Adding Introduction chapter; moving SoC Description chapter (short ver.) to Block Description chapter. Minor improvements.Revision 1.32016-02-11Addign RF Section Power Supply chapter. Minor improvements.Revision 1.22016-01-28Fixing image render problem.Revision 1.12015-25-25Updating Inverted-F Antenna Chapter and radiated measurement results based on board revision B02.Revision 1.02015-11-27Initial release.Table of Contents1. Introduction (1)2. Radio Board Connector (2)2.1 Introduction (2)2.2 Radio Board Connector Pin Associations (2)3. Radio Board Block Summary (3)3.1 Introduction (3)3.2 Radio Board Block Diagram (3)3.3 Radio Board Block Description (3)3.3.1 Wireless MCU (3)3.3.2 LF Crystal Oscillator (LFXO) (3)3.3.3 HF Crystal Oscillator (HFXO) (3)3.3.4 Matching Network for 2.4 GHz (3)3.3.5 Inverted-F Antenna (4)3.3.6 UFL Connector (4)3.3.7 Radio Board Connectors (4)4. RF Section (5)4.1 Introduction (5)4.2 Schematic of the RF Matching Network (5)4.2.1 Description of the 2.4 GHz RF Matching (5)4.3 RF Section Power Supply (5)4.4 Bill of Materials for the 2.4 GHz Matching (5)4.5 Inverted-F Antenna (6)5. Mechanical Details (7)6. EMC Compliance (8)6.1 Introduction (8)6.2 EMC Regulations for 2.4 GHz (8)6.2.1 ETSI EN 300-328 Emission Limits for the 2400-2483.5 MHz Band (8)6.2.2 FCC15.247 Emission Limits for the 2400-2483.5 MHz Band (8)6.2.3 Applied Emission Limits for the 2.4 GHz Band (8)7. RF Performance (9)7.1 Conducted Power Measurements (9)7.1.1 Conducted Measurements in the 2.4 GHz band (9)7.2 Radiated Power Measurements (10)7.2.1 Radiated Measurements in the 2.4 GHz band (10)8. EMC Compliance Recommendations (11)8.1 Recommendations for 2.4 GHz ETSI EN 300-328 compliance (11)8.2 Recommendations for 2.4 GHz FCC 15.247 compliance (11)9. Board Revisions (12)10. Errata (13)11. Document Revision History (14)Table of Contents (15)Silicon Laboratories Inc.400 West Cesar Chavez Austin, TX 78701USASimplicity StudioOne-click access to MCU and wireless tools, documentation, software, source code libraries & more. Available for Windows, Mac and Linux!IoT Portfolio /IoTSW/HW/simplicityQuality/qualitySupport and CommunityDisclaimerSilicon Labs intends to provide customers with the latest, accurate, and in-depth documentation of all peripherals and modules available for system and software implementers using or intending to use the Silicon Labs products. Characterization data, available modules and peripherals, memory sizes and memory addresses refer to each specific device, and "Typical" parameters provided can and do vary in different applications. Application examples described herein are for illustrative purposes only. Silicon Labs reserves the right to make changes without further notice and limitation to product information, specifications, and descriptions herein, and does not give warranties as to the accuracy or completeness of the included information. Silicon Labs shall have no liability for the consequences of use of the information supplied herein. This document does not imply or express copyright licenses granted hereunder to design or fabricate any integrated circuits. The products are not designed or authorized to be used within any Life Support System without the specific written consent of Silicon Labs. A "Life Support System" is any product or system intended to support or sustain life and/or health, which, if it fails, can be reasonably expected to result in significant personal injury or death. Silicon Labs products are not designed or authorized for military applications. Silicon Labs products shall under no circumstances be used in weapons of mass destruction including (but not limited to) nuclear, biological or chemical weapons, or missiles capable of delivering such weapons.Trademark InformationSilicon Laboratories Inc.® , Silicon Laboratories®, Silicon Labs®, SiLabs® and the Silicon Labs logo®, Bluegiga®, Bluegiga Logo®, Clockbuilder®, CMEMS®, DSPLL®, EFM®, EFM32®, EFR, Ember®, Energy Micro, Energy Micro logo and combinations thereof, "the world’s most energy friendly microcontrollers", Ember®, EZLink®, EZRadio®, EZRadioPRO®, Gecko®, ISOmodem®, Precision32®, ProSLIC®, Simplicity Studio®, SiPHY®, Telegesis, the Telegesis Logo®, USBXpress® and others are trademarks or registered trademarks of Silicon Labs. ARM, CORTEX, Cortex-M3 and THUMB are trademarks or registered trademarks of ARM Holdings. Keil is a registered trademark of ARM Limited. All other products or brand names mentioned herein are trademarks of their respective holders.。

类专业英语电子教案第十三课件

类专业英语电子教案第十三课件
Mobile phones contain a large amount of circuitry, each of which is carefully designed to optimize its performance. The cell phone comprises analogue electronics as well as digital circuits ranging from processors to display and keypad electronics. A mobile phone typically consists of a single board only, but within this there are a number of distinct functional areas: radio frequency - receiver and transmitter; digital signal processing; analogue / digital conversion; control processor; SIM or USIM card; power control and battery.
equalisation [ i:kwili‘zeiʃən ] n. 均衡;补偿 handset [ 'hændset ] n. 电话听筒, 手持机 infrastructure [ 'infrə'strʌktʃə ] n. 基础结构;基础设施 keypad [ 'ki:pæd ] n. 小键盘 lithium [ 'liθiəm ] n. 锂 migrate [ mai'greit, 'maigreit ] v. 移动, 迁移 monitor [ 'mɔnitə ] v. 监视 n. 显示器 nickel [ 'nikl ] n. 镍 v. 镀镍于 protocol [ 'prəutəkɔl ] n. 草案;协议 ringtone [ riŋtəun ] n. 手机铃音 stack [ stæk ] n. 堆叠;堆栈 animation [ æni‘meiʃən ] n.卡通制作 subscriber [ sʌbs'kraibə ] n. 电话用户;签署者;预订者;订阅者

Coyote 2.5高级测试设备租赁手册说明书

Coyote 2.5高级测试设备租赁手册说明书

1981Coyote manual version 2.5Page 1ABOUT YOUR COYOTE............................................................................... 2COYOTE SIDE PANEL................................................................................. 2 COYOTE TOP PANEL.................................................................................. 2POWER ON............................................................................................. 2COYOTE BATTERY INFORMATION................................................................... 3MEASURING SIGNAL STRENGTH................................................................... 4MENU DISPLAY........................................................................................ 4INFORMATION SCREEN.............................................................................. 4GENERAL OPTIONS................................................................................... 5 SET TIME (RTC)....................................................................................... 5CARD OPTIONS........................................................................................ 5 FORMAT CARD........................................................................................ 6 SINGLE CHANNEL..................................................................................... 6 CHANGE MEASUREMENT FREQUENCY............................................................ 7GRAPHIC DISPLAY SCREEN......................................................................... 7 NUMERIC DISPLAY SCREEN......................................................................... 7FILTER SELECTION SCREEN......................................................................... 8RATE SELECTION SCREEN........................................................................... 8ACQUIRING GPS....................................................................................... 9GPS LOCKED........................................................................................... 9GPS OFF................................................................................................ 9SCAN OPTIONS........................................................................................ 10CHANNEL SCAN LIST................................................................................. 10LIST SCAN.............................................................................................. 10COYOTE MECHANICAL DRAWING.................................................................. 11COYOTE ACCESSORIES (12)Coyote Data Logger Software Introduction ................................................................................... 13 Installation .................................................................................... 13 Application Overview ....................................................................... 13 Quick Start .................................................................................... 13 Speed Button Panel ......................................................................... 14 RTC/GPS Status Bar .......................................................................... 14 Receiver Status Bar ......................................................................... 15 Bottom Status Bar ........................................................................... 15 Logging Warning .. (15)Coyote Channel Table Utility (16)BVS Chameleon CW User Manual (19)Coyote Dead Reckoning Optional Software (25)Optional Omni-Directional Antennae (30)Glossary of Acronyms (33)General Safety InstructionsCoyote is a trademarks of Berkeley Varitronics Systems.SolidState Data Drive is a registered trademark ® of ADTRON Corporation.IBM PC is a registered trademark ® of IBM Corporation.MS-DOS is a registered trademark ® of Microsoft Corporation.Microsoft EXCEL is a registered trademark ® of Microsoft Corporation.Page 2The Coyote™ is a high performance, modular receiversystem providing precision, two-channel signal strengthmeasurements using two independent receivers. It isinternally powered (or may be externally) and logs mea-surements and displays graphically a wide assortment ofbuilt-in realtime macro measurements.COYOTE OPERATIONLCD SCREEN - displays realtime dataPOWER SWITCH - turns On/OFF CoyoteLOW BATTERY LED - indicates when to recharge unitGPS ON/OFF SWITCH - disables GPS for indoor useDIRECTION ARROWS - navigates through menusNUMERIC KEYPAD - allows for direct frequency & data entryOperation of the Coyote is simple. Use the push buttonarrows to navigate through the menus. Coyote is designedto be used with or without a PC laptop in the field. A USBconnection allows for realtime data analysis and record-ing from the Coyote to any USB equipped PC. Becausedata can be stored internally, removed and then insertedinto any standard compact flash PCMCIA adaptor, COYOTEallows for flexibility in walk-about, drive studies and postprocessing sessions.COYOTE BACK PANELRECEIVER 1 MODULE HOST (REMOVABLE)RECEIVER 2 MODULE HOST (REMOVABLE)GPS RECEIVER MODULE HOST (REMOVABLE)COMPACT FLASH CARD SLOT (REMOVABLE)USB PORT (TYPE B)POWER INPUT (12V)RS-232 SERIAL PORT ( RJ-11CONNECTOR)F rom this side, the user has access to all of Coyote’smodular features (except for Li-Ion battery located atopposite end of Coyote ). Any Coyoteshown here.Page 4Page 5keys to scroll up and down through settings.Page 6NOTE: BVS does not recommend the use of any CF card for data logging during drive-studies. Use a laptop with the recommended minimum specifications.Page 7Page 8Coyote FilteringChameleon CW averages 16 samples to produce one sample in the output file. This is selectable in all versions greater than 1.10. You can set the report rate to 16 times the value that you need on the Coyote for earlier versions of Chameleon CW.The filters in Coyote are low pass digital filters and are conceptually the same as exponential filtering. This means that both filters essentially average the data in a moving time window. On Coyote, you set the time window with the “Filter” setting in the single measurement options. Note that regardless of the time constant of the filter, there is still the same number of samples per second out of the filter as there was going into the filter. The “Rate” selection in the Coyote allows you to reduce the number of reported samples after the filter.What is confusing about this is that these two processes are often combined in other products. In this case, N samples are taken, averaged and a single data point is produced. This is equivalent to filtering with a window of length N followed by a rate reduction of 1/N. This combination provides less flexibility, but it is certainly easier to make. If you wish to do this type of filtering on the Coyote, set the filter time constant and the sampling period the same, i.e. 1 second filter and 1 report per second or 1/2 second filter and 2 reports per second.Page 9COYOTE MECHANICAL DRAWINGCoyoteAccessories for your GPS Module Receiver Module User specified frequencyRS-232 Serial Cable DB-9 Female to RJ-11Compact Flash Card USB Cable A to B connectorSame day factory calibration SMA connectorCoyote Power1963.751980.00orChannel Table 1, 07/0301, 99, C 1223331111Coyote ChameleonSoftware User ManualI n t r o d u c t i o nThe Chameleon application software is the data conversion and filtering tool for BVS Coyote CW Receiver(s). Chameleon was designed to greatly simplify the conversion of binary receiver data to an ASCII delimited format. Most post-processing applications can then import this converted data. The following sections of this document outline how to convert data using the Chameleon software.I n s t a l l a t i o nInstallation of Chameleon is straightforward. Insert the media provided with the Coyote purchased into the PC and find the Chameleon folder. Run the setup application.Next, follow the steps outlined for this application. After the installation has been completed, an icon will be placed in the chosen folder.Figure 1 – Main ScreenRunning the ApplicationAfter starting the application, the main screen will appear. The steps to conversion are outlined in the following sections.M a i n S c r e e nThe main screen is the gateway to all of the conversion options available. Once the requested options have been selected, begin the data conversion process by pressing the “CONVERT” button. Cancel at any time by pressing the “CANCEL” button.The options chosen are saved in a configuration file on the hard drive. When starting the Chameleon application, the last configuration used will be loaded.There are three separate areas from where options are to be chosen. These are “File Options”, “Field Options”, and “Filter Options”. All of these areas are discussed below.F i l e O p t i o n sPressing the “File Options” button will display a dialog box as shown in Figure 2.Figure 2 - File OptionsThe configuration files are stored in ASCII form. DO NOT modify these configurations manually! Any manual change to the configuration files may result in the loss of configuration information.I n p u t/O u t p u t F i l e sChameleon can convert a single file or can convert multiple files. The choice is made by selecting the radio button next to either “Single File” or “Multiple Files”.Single FileChoose the data file that is to be converted as the input.Multiple FilesFor multiple files, select the input folder. Then select the file extension (default is *.klf). After the folder has been selected, the number of files that have been found will show up in the “Count” box.Choose the output folder for the output files. The resulting output will be placed in this folder.Output File SplitThere is a check box option to split the output file. If this choice is selected, the output will be be split into multiple files if the record count is greater than thenumber selected.This option is useful where a post-processing application (such as Microsoft Excel TM) cannot except files with more than a certain number of rows.Field OptionsPressing the “Field Options” button from the main screen will display a dialog box to choose output fields. This box can be seen in Figure 3.O u t p u t F i e l d sSelect the fields that are to be placed in the output file. Simply highlight fields desired in the “Available Fields” box and then move them to the “Selected Fields” box by using the directional arrow buttons.The delimiting character may also be chosen. Field titles may be placed in the output file by checking the “Header” box. If the resulting output is to be used in iBwave, check the “Export to iBwave format” box.Units may also be chosen. The units may be English (e.g. MPH), or Metric (e.g. km/h).Figure 3 – Field OptionsFilter OptionsPressing the “Filter Options” button from the main screen will display a dialog box to choose different ways of reducing output data or averaging the output data. This box can be seen in Figure 4.Either one of the Coyote receivers' data may be converted and output. If it is desired, both receivers' data can be output.The averaging selection has three choices. No averaging may be chosen. Selecting this option, a record will be output for every RSSI value collected from the receiver(s).Selecting standard averaging, any RSSI values which appear in a single data packet from the Coyote (up to 16 values) will be averaged down to one value and output to a record.The final averaging type that can be selected is “40 Lambda”. This will average the data to suppress the effect of Rayleigh fading.For a greater understanding of Rayleigh fading, please see the documentation available on the BVS website at .Figure 4 – Filter OptionsThere are three filtering options. The first two options are “None” and “Best Server”. “None” will not alter the data in any way.“Best Server” will choose the frequency which has the strongest signal for every pass. This option is only available for “List” scanning or “Table” scanning modes. There will be no effect if selected with “Single Channel” mode data.The final option is to drop all data where there is not a GPS position lock. This will eliminate any data where the position cannot be guaranteed according to GPS. The amount of data dropped will follow the value selected for the “seconds”. For example, if 40 seconds is selected, Data will be dropped if a GPS outage lasts longer than 40 seconds.Coyote Dead Reckoning Optional SoftwareThe Coyote Dead Reckoning (DR) version allows you to choose which navigation data to use. The naviga-tion data can come either from the internal GPS on the Coyote or from an external Blaupunkt TravelPilot. The DX-V or EX-V versions are supported.IMPORTANT ALWAYS TURN THE COYOTE’S GPS POWER OFF WHEN USING THE TRAVELPILOT!The TravelPilot will feed data into the Coyote DR via an additional serial port (RS-232).The opening screen (F igure 1) will let you choose between the internal GPS and a port for the TravelPilot.If choosing the TravelPilot, The Coyote DR will continuously ask the TravelPilot for its version information. Once that is received, it will continuously ask for updated latitude and longitude information.When logging a file, the TravelPilot position information will be coupled with current date and time from the computer running the Coyote DR software.Figure 1 – Coyote Port Selection ScreenThe information from the TravelPilot will continuously be displayed in the lower right-hand portion of the main screen. The TravelPilot data will be successfully received if the version information is there and the update count continues to increase.Figure 2 – Coyote Main ScreenThe logged data can be used as GPS data in Chameleon, Sieve, and/or Forecaster for post-processing.There are 4 extra fields to choose from for a Coyote log file with dead reckoning. They will be on the bottom of the list and include:TP_Latitude – TravelPilot Latitude in decimal degreesTP_Longitude – TravelPilot Longitude in decimal degreesTP_Date – MM/DD/YY from the computer on which the Coyote DR is running.TP_Time – HH:MM:SS from the computer on which the Coyote DR is running.Figure 3 – Coyote GPS OffThis shows “GPS OFF”. This will either appear at the beginning of the session (before a measurement is started and/or GPS power is applied) or after the GPS power has been shut off and RTC packets are beingreceived (accompanied by a PING sound).Figure 4 – Dead Reckoning GPS OffThis shows “GPS OFF” for the dead reckoning software. This will either appear at the beginning of the session (before a measurement is started and/or GPS power is applied) or after the GPS power has been shut off and RTC packets are being received (accompanied by a PING sound). Note that the TravelPilotfields are no longer there. This means that Internal GPS was selected.Figure 5 – TravelPilot Data Not Being ReceivedThis shows TravelPilot has been selected. The GPS data fields will not display on the main screen. This picture shows TravelPilot data not being received.TravelPilot Serial Cable DiagramWhen connecting the TravelPilot to your computer, if a gender changer is needed, it cannot be pin-for-pin. The Rx and Tx lines need to be crossed (as shown in the diagram). Some manufacturers of gender changers will connect pins 1-9 to pins 1-9 directly. The lines need to be connected as in the diagram. Otherwise, youwill not be able to communicate to the TravelPilot.4.9/5 GHz Omni-Directional (5.5” long)Optional Omni-Directional Antennae 900 MHz Omni-Directional (9.5” long)50 Ohm < 2.0:1 avg.Maximum Input Power100 W Mechanical Specifications Vertical Vertical Beam Width15°Horizontal Beam Width360°0.5 lbs. (.22 kg)20 in. (508 mm)1.27 in. (3.22 cm)Radome Diameter.75 in. (1.90 cm)Radome MaterialGray Fiberglass >150 MPH Operating Temperature-40° C to to 85° C (-40° F to 185° F)Antenna ConnectorIntegral N-MaleWind Loading Data Wind Speed (MPH)Loading 3 lb.5 lb.Glossary of AcronymsAC Alternating CurrentA/D Analog to Digital converterAGC Automatic Gain ControlApplet a small ApplicationBER Bit Error RateBPSK Binary Phase Shift KeyingBW Band WidthCDMA Code Division Multiple Access (spread spectrum modulation)DC Direct CurrentD/A Digital to AnalogdB decibeldBm decibels referenced to 1 milliwattDOS Digital Operating SystemDSP Digital Signal ProcessingFIR Finite Impulse ResponseGHz GigaHertzGPS Global Positioning System (satellite based)GPS diff. GPS error correction signal which enhances GPS accuracyIF intermediate frequencyI and Q I n phase and QuadraturekHz kiloHertzLCD Liquid Crystal DisplayLO Local OscillatorMbits MegabitsMHz MegaHertzmodem modulator/demodulatorPC Personal ComputerPCS Personal Communications Service (1.8 to 2.1 GHz frequency band)PN Pseudo NoiseQPSK Quaternary Phase Shift Keying, 4-level PSKRF Radio FrequencyRSSI Receiver Signal Strength IndicatorUCT Universal Coordinated TimeVAC Volts Alternating CurrentVGA Video graphicIMPORTANT GENERAL SAFETY INSTRUCTIONSWhen using your telephone equipment, basic safety precautions should always be followed to reduce the risk of fire, electric shock and injury to persons, including the following:1)Read and understand all instructions.2)Follow all warnings and instructions marked on the product.3)Unplug this product from the wall outlet before cleaning. Do not use liquid cleaners or aerosol cleaners. Use a damp cloth for cleaning.4)Do not use this product near water, for example, near a bath tub, wash bowl, kitchen sink, or laundry tub, in a wet basement, or near a swimming pool.5)Do not place this product on an unstable cart, stand, or table. The product may fall, causing serious damage to the product.6)Slots and openings in the cabinet and the back or bottom are provided for ventilation, to protect it from overheating these openings must not be blocked or covered The openings should never be blocked by placing the product on the bed, sofa, rug or other similar surface. This product should never be placed near or over a radiator or heat register. This product should not be placed in a built-in installation unless proper ventilation is provided.7) This product should be operated only from the type of power source indicated on the appliance. If you are not sure of the type of power supply to your home, consult your dealer or local power company.8)Do not allow anything to rest on the power cord. Do not locate this product where the cord will be abused by persons walking on it.9)Do not overload wall outlets and extension cords as this can result in the risk of fire or electric shock.10)Never push objects of any kind into this product through cabinet slots as they may touch dangerous voltage points or short out parts that could result in a risk of fire or electric shock. Never spill liquid of any kind on the product.11) To reduce the risk of electric shock, do not disassemble this product, but take it to a qualified service faciI4 when some service or repair work is required. Opening or removing covers may expose you to dangerous voltages or other risks. Incorrect reassembly can cause electric shock when the appliance is subsequently used.12)Unplug this product from the wall outlet and refer servicing to qualified service personnel under the following conditions:A) When the power supply cord or plug is damaged or frayed. B) If liquid has been spilled into the product.C)If the product has been exposed to rain or water.D) If the product does not operate normally by following the operating instructions. Adjust only those controls, that are covered by the operating instructions because improper adjustment of other controls may result in damage and will often require extensive work by a qualified technician to restore the product to normal operation.E) If the product has been dropped or the cabinet has been damaged. F) If the product exhibits a distinct change in perfor-mance.13)Avoid using the product during an electrical storm. There may be a remote risk of electric shock from lightning.14)Do not use the telephone to report a gas leak in the vicinity of the leak.INSTALLATION INSTRUCTIONS1. Never install telephone wiring during a lightning storm.2. Never install telephone jacks in wet locations unless the jack is specifically designed for wet locations.3. Never touch uninsulated telephone wires or terminals unless the telephone line has been disconnected at the network inter-face.4. Use caution when installing or modifying telephone lines.INSTRUCTION FOR BATTERIESCAUTION: To Reduce the Risk of Fire or Injury to Persons, Read and Follow these Instructions:1. Use only the type and size of batteries mentioned in owner’s manual.2. Do not dispose of the batteries in a fire. The cells may explode. Check with local codes for possible special disposal instructions.3. Do not open or mutilate the batteries. Released electrolyte is corrosive and may cause damage to the eyes or skin. It may be toxic if swallowed.4. Exercise care in handling batteries in order not to short the battery with conducting materials such as rings, bracelets, and keys. The battery or conductor may overheat and cause burns.5. Do not attempt to recharge the batteries provided with or identified for use with this product. The batteries may leak corrosive electrolyte or explode.6. Do not attempt to rejuvenate the batteries provided with or identified for use with this product by heating them. Sudden release of the battery electrolyte may occur causing burns or irritation to eyes or skin.7. When replacing batteries, all batteries should be replaced at the same time. Mixing fresh and discharged batteries could increase internal cell pressure and rupture the discharged batteries. (Applies to products employing more than one sepa-rately replaceable primary battery.)8. When inserting batteries into this product, the proper polarity or direction must be observed. Reverse insertion of bat-teries can cause charging, and that may result in leakage or explosion. (Applies to product employing more than one separately replaceable primary battery.)9. Remove the batteries from this product if the product will not be used for a long period of time (several months or more) since during this time the battery could leak in the product.10. Discard “dead” batteries as soon as possible since “dead” batteries are more likely to leak in a product.11. Do not store this product, or the batteries provided with or identified for use with this product, in high-temperature areas. Batteries that are stored in a freezer or refrigerator for the purpose of extending shelf life should be protected from con-densation during storage and defrosting. Batteries should be stabilized at room temperature prior to use after cold storage.。

ADB使用指南说明书

ADB使用指南说明书

Table of ContentsAbout1 Chapter 1: Getting started with adb2 Remarks2 Examples2 Installation or Setup2 Introduction3 Chapter 2: Collecting adb commands log4 Remarks4 Examples4 in Windows4 Chapter 3: Connecting to device5 Examples5 Finding devices connected to your PC5 Chapter 4: Showing Logs on ADB6 Examples6 Displaying and filtering with Logcat6 Chapter 5: Starting an app in debug mode8 Examples8 How to wait for debugger before starting the app?8 Chapter 6: Transferring files using push and pull9 Syntax9 Parameters9 Remarks9 Examples9 Push a file to the SD card9 Pull a file from the SD card to the current working directory9 Credits10AboutYou can share this PDF with anyone you feel could benefit from it, downloaded the latest version from: adbIt is an unofficial and free adb ebook created for educational purposes. All the content is extracted from Stack Overflow Documentation, which is written by many hardworking individuals at Stack Overflow. It is neither affiliated with Stack Overflow nor official adb.The content is released under Creative Commons BY-SA, and the list of contributors to each chapter are provided in the credits section at the end of this book. Images may be copyright of their respective owners unless otherwise specified. All trademarks and registered trademarks are the property of their respective company owners.Use the content presented in this book at your own risk; it is not guaranteed to be correct nor accurate, please send your feedback and corrections to ********************Chapter 1: Getting started with adbRemarksThis section provides an overview of what adb is, and why a developer might want to use it.It should also mention any large subjects within adb, and link out to the related topics. Since the Documentation for adb is new, you may need to create initial versions of those related topics. ExamplesInstallation or SetupSpecific to Windows System and android Phone:Requirements:1.USB CableAndroid Device2.3.Android Driver SoftwareBasically after connecting USB cable PC detects the Android Device and it will automatically search for the required Drivers for that Android Device. If that drivers are not found then you have to install manually.Manual Installation:1.First install Android SDK in your PC(Windows)After installing in Android SDK tools Right click on the SDK Manager and select "Run as2.Administrator"In the SDK Manager select "Extras->Google USB Driver". Enable the checkbox and click3."Install 1 Package"When the Google USB driver is installed, plug in your device. Warning: The driver won't4.install automatically. We will do it manually in the next steps.5.Open the System Properties dialog (press Win+Break on the keyboard or locate "Computer"in Start Menu, right-click on it and select "Properties".6.Click on the "Device Manager" link.7.In the Device Manager locate your Android device. Then right-click on it and select "Update Driver Software".8.Select "Browse my computer for driver software".9.Select "Let me pick from a list of device drivers on my computer".10.Select "Show All Devices".11.Press the "Have Disk" button.12.Enter the path to the Google USB driver. Normally it is located in the followingdirectory:C:\Program Files (x86)\Android\android-sdk\extras\google\usb_driver13.Select "Android ADB Interface" from the list of device types.14.Confirm the installation of the driver by pressing "Yes".15.Confirm the installation again by pressing "Install".16.When the installation is done, press "Close".17.Introductionadb is a command line tool for communicating with an emulator instance or connected device. It allows for installing and debugging apps, transferring files, as well as a variety of other interactions with the connected emulator or device. The ADB system consists of a client, which sends commands from the host computer, a daemon, which runs on the connected device and executes commands received from the client, and a server, which runs on the host computer and manages communications between the client and daemon.Official Documentationhttps:///studio/command-line/adb.htmlRead Getting started with adb online: https:///adb/topic/2633/getting-started-with-adbChapter 2: Collecting adb commands logRemarksMake sure that your automation does not use adb kill-server command.Examplesin WindowsOpen a Command Prompt window and run the following commands:adb kill-serverset ADB_TRACE=socketsadb nodaemon server 2>&1 | for /f "usebackq tokens=7*" %a in (`findstr /c:"): '"`) do @echo %a %b >> %USERPROFILE%\Desktop\adb_host_log.txtNow you can run your Android automation. When done run adb kill-server in another Command Prompt window. Now the adb_host_log.txt file on your Desktop contains the log of all commands all adb clients have sent to the adb host.Read Collecting adb commands log online: https:///adb/topic/5631/collecting-adb-commands-logChapter 3: Connecting to deviceExamplesFinding devices connected to your PCEnable USB Debugging on your device and from command line type adb devices. If everything is OK, the response should be:List of devices attached1234567890 deviceWhere 1234567890 is the device's id.If multiple devices are connected, you should see all of them:List of devices attached1234567890 device2222222222 device...When connecting a device for the first time, you'll get a pop-up window on your device, asking you to approve the connection.Read Connecting to device online: https:///adb/topic/3174/connecting-to-deviceChapter 4: Showing Logs on ADBExamplesDisplaying and filtering with LogcatDisplaying all the logs from the default buffer on the Command Line can be accomplished by:adb logcatThis command will show you all the logs from the device's main buffer. Notice that if you use it for the first time, you'll get a lot of information, an enormous stream of data. So you may want to clear the logs first...Cleaning the logs:adb logcat -cThis will clean clear the logs, and start fresh.Displaying Alternate BuffersThere are two other buffers besides the main buffer that may be displayed as follows:adb logcat -b buffer_name,where buffer_name is one of the following:•radio - view the buffer that contains radio/telephony related messages.•events - view the buffer containing events-related messages.•main - view the main log buffer (default)Filtering Log OutputLogcat logs got so called levels:V — Verbose, D — Debug, I — Info, W — Warning, E — Error, F — Fatal, S — Silent Those levels are specified when application uses those Log function:Log.v(); // VerboseLog.d(); // DebugLog.i(); // InfoLog.w(); // WarningLog.e(); // Errorif your code Log call is:Log.i("MainActivtyTag", "Showing the very first fragment");in logcat you'll see this output:07-27 11:34:21.027 I MainActivtyTag 66 : Showing the very first fragmentSo, this is the log convention:<timestamp> <logLevel> <tag> <line> : <messge>For instance, if you want to show all the logs that have Fatal (F) level:adb logcat *:F* is a what called a wild card - stands for all package namesFiltering by application package nameSince package names are guaranteed to be unique , you can filter logcat by your package name, of course you can combine it with the Level filter:adb logcat <package name>:<log level>For exiting/interrupting process - press Ctrl + XRead Showing Logs on ADB online: https:///adb/topic/4252/showing-logs-on-adbChapter 5: Starting an app in debug mode ExamplesHow to wait for debugger before starting the app?Let's say your launch activity is called MainActivity, in your app com.example.myapp. In the manifest:<activityandroid:name=".MainActivity"><intent-filter><action android:name="android.intent.action.MAIN"/><category android:name="UNCHER"/></intent-filter></activity>Now let's say you want to launch the app, so that it waits for the debugger to connect before the app really starts.You can use adb shell to achieve that.In our case, simply run:adb shell am start -D -n com.example.myapp/com.example.myapp.MainActivityNow, all that's left is to attach your favorite debugger. For example, if you use Intellij or Android Studio go to Run->Attach debugger to Android process-> select your app package nameRead Starting an app in debug mode online: https:///adb/topic/4009/starting-an-app-in-debug-modeChapter 6: Transferring files using push and pullSyntax•adb push [-p] LOCAL REMOTE•adb pull [-a] [-p] REMOTE [LOCAL]ParametersRemarksIf LOCAL is omitted in the adb pull command, the filename from REMOTE is usedLOCAL can be a relative path or an absolute path, but REMOTE must be an absolute path ExamplesPush a file to the SD cardadb push file.txt /sdcard/Pull a file from the SD card to the current working directoryadb pull /sdcard/file.txtRead Transferring files using push and pull online:https:///adb/topic/5844/transferring-files-using-push-and-pullCredits。

College+English+Test+Band+4+News+Listening+Coursew

College+English+Test+Band+4+News+Listening+Coursew
题型分析
对历年的新闻听力真题进行深入分析 ,总结出题规律和特点,帮助学生了 解考试要求和难度。
解题技巧
提供针对不同题型的解题技巧,如主 旨大意题、细节理解题、推理判断题 等,帮助学生提高答题正确率。
Training on News Listening Simulation Questions
模拟试题
Space exploration
Discuss the exploration and development of outer space, including satellites, rockets, and space stations.
03
News Listening Training
Analysis of News Listening Questions
科技发展: Staying up-to-date with the latest scientific and technological advancements is essential for comprehending news related to science, technology, and innovation. This includes knowledge of major scientific discoveries, technological breakthroughs, and their applications.
Labor market
Discuss the supply and demand in the labor market, including salaries, benefits, and working conditions.
Social news vocabulary

CIC+Filter+Introduction

CIC+Filter+Introduction

CIC Filter IntroductionMatthew P.Donadiom.p.donadio@18July2000For Free Publication by Iowegian1IntroductionAs data converters become faster and faster,the application of narrow-band extraction from wideband sources,and narrow-band construction of wideband signals is becoming more important.These functions require two basic signal processing procedures:decimation and interpolation.And while digital hardware is becoming faster,there is still the need for efficient solutions.Techniques found in[CR83]work very well in practice,but large rate changes require very narrow bandfirge rate changes require fast multipliers and very longfilters.This can end up being the largest bottleneck in a DSP system.In[Hog81],an efficient way of perfoming decimation and interpolation was introduced. Hogenauer devised aflexible,multiplier-freefilter suitable for hardware implementation,that can also handle arbitrary and large rate changes.These are known as cascaded integrator-combfilters,or CICfilters for short.This paper sumarizes thefindings published in[Hog81].An overview can also be found in [Fre94].An extension of CICfilters has been published in[KJW97],and is briefly mentioned here.When in doubt,the reader should refer to these sources.2Building BlocksThe two basic building blocks of a CICfilter are an integrator and a comb.An integrator is simply a single-pole IIRfilter with a unity feedback coeficient:y[n]=y[n−1]+x[n](1) This system is also known as an accumulator.The transfer function for an integrator on thez-plane isH I(z)=11−z−1(2)1Using the equations from [OS89]for a single pole system,we can determine that|H I (e jω)|2=12(1−cos ω)ARG[H I (e jω)]=−tan −1 sin ω grd[H I (e jω)]= undefined ω=0−1ω=0(3)The power response is basically a low-pass filter with a −20dB per decade (−6dB per octave)rolloff,but with infinite gain at DC.This is due to the single pole at z =1;the output can grow without bound for a bounded input.In other words,a single integrator by itself is unstable.#?--z −1Figure 1:Basic IntegratorA comb filter running at the high sampling rate,f s ,for a rate change of R is an odd-symetric FIR filter described byy [n ]=x [n ]−x [n −RM ](4)In this equation,M is a design parameter and is called the differential delay .M can be any positive integer,but it is usally limited to 1or 2.The corresponding transfer at f sH C (z )=1−z −RM(5)Again,we can determine that|H C (e jω)|2=2(1−cos RMω)ARG[H C (e jω)]=−RMωgrd[H C (e jω)]=RM 2(6)When R =1and M =1,the power response is a high-pass function with 20dB per decade (6dB per octave)gain (after all,it is the inverse of an integrator).When RM =1,then the power response takes on the familiar raised cosine form with RM cycles from 0to 2π.When we build a CIC filter,we cascade,or chain output to input,N integrator sections together with N comb sections.This filter would be fine,but we can simplify it by combining it with the rate ing a technique for multirate analysis of LTI systems from [CR83],we can “push”the comb sections through the rate changer,and have them becomey [n ]=x [n ]−x [n −M ](7)2at the slower sampling rate f s R .We accomplish three things here.First,we have slowed down half of the filter and therefore increased efficiency.Second,we have reduced the number ofdelay elements needed in the comb sections.Third,and most important,the integrator and comb structure are now independent of the rate change.This means we can design a CIC filter with a programmable rate change and keep the same filtering structure.#-?--z −MFigure 2:Basic CombTo summarize,a CIC decimator would have N cascaded integrator stages clocked at f s ,followed by a rate change by a factor R ,followed by N cascaded comb stages runningat f s .A CIC interpolator would be N cascaded comb stages running at f s ,followed be a zero-stuffer,followed by N cascaded integrator stages running at f s ."!#--------?I I I C C C R Figure 3:Three Stage Decimating CIC Filter "!#--------6R C C C I I I Figure 4:Three Stage Interpolating CIC Filter3Frequency CharacteristicsThe transfer function for a CIC filter at f s is H (z )=H N I (z )H N C (z )=(1−z −RM )N (1−z −1)N = RM −1k =0z −k N (8)This equation shows that even though a CIC has integrators in it,which by themselves have an infinite impulse response,a CIC filter is equivalent to N FIR filters,each having a3rectangular impulse response.Since all of the coeficients of these FIRfilters are unity,and therefore symetric,a CICfilter also has a linear phase response and constant group delay.The magnitude response at the output of thefilter can be shown to be|H(f)|=sinπMfsinRN(9)By using the relation sin x≈x for small x and some algebra,we can approximate this function for large R as|H(f)|≈RM sinπMfπMfN for0≤f<1M(10)We can notice a few things about the response.One is that the output spectrum has nulls at multiples of f=1.In addition,the region around the null is where aliasing/imaging occurs.If we define f c to be the cutoffof the usable passband,then the aliasing/imaging regions are at(i−f c)≤f≤(i+f c)(11) for f≤1and i=1,2,···, R .If f c≤M,then the maximum of these will occur at the lower edge of thefirst band,1−f c.The system designer must take this into consideration, and adjust R,M,and N as needed.Another thing we can notice is that the passband attenuation is a function of the number of stages.As a result,while increasing the number of stages improves the imaging/alias rejection,it also increases the passband“droop.”We can also see that the DC gain of the filter is a function of the rate change.4Bit GrowthFor CIC decimators,the gain G at the output of thefinal comb section isG=(RM)N(12) Assuming two’s complement arithmetic,we can use this result to calculate the number of bits required for the last comb due to bit growth.If B in is the number of input bits,then the number of output bits,B out,isB out= N log2RM+B in (13)It also turn out that B out bits are needed for each integrator and comb stage.The input needs to be sign extended to B out bits,but LSB’s can either be truncated or rounded at later stages.The analysis of this is beyond the scope of this tutorial,but is fully described in[Hog81].For a CIC interpolator,the gain,G,at the i th stage isG i= 2i i=1,2,···,N22N−i(RM)i−N,i=N+1,···,2N(14)4As a result the register width,W i,at i th stage isW i= B in+log2G i (15) andW N=B in+N−1(16) if M=1.Rounding or truncation cannot be used in CIC interpolators,except for the result, becuase the small errors introduced by rounding or truncation can grow without bound in the integrator sections.It is now worth revisiting the unstable aspect of the integrator stages.It turns out that it is not a problem.For decimators,integrator overflow is not a problem as long as two’s complement math is used and we don’t expect an overall system gain>1.For interpolators, the comb stages and zero stuffing will prevent integrator overflow.5Implementation DetailsBecause of the passband droop,and therefore narrow usable passband,many CIC designs utilize an additional FIRfilter at the low sampling rate.Thisfilter will equalize the passband droop and perform a low rate change,usually by a factor of two to eight.In many CIC designs,the rate change R is programmable.Since the bit growth is a function of the rate change,thefilter must be designed to handle both the largest and smallest rate changes.The largest rate change will dictate the total bit width of the stages, and the smallest rate change will determine how many bits need to be kept in thefinal stage. In many designs,the output stage is followed by a shift register that selects the proper bits for transfer to thefinal output register.A system designer can use the equation for B out for a decimator and W2N for an interpolator to calculate proper shift values.For a CIC decimator1,the normalized gain at the output of the last comb is given byg=(RM)N2 N log2RM(17)This lies in the interval(1,1].Note that when R is a power of two,the gain is unity.This gain can be used to calculate a scale factor,s,to apply to thefinal shifted output.s=2 N log2RM(RM)N(18)which lies in the interval[1,2).By doing this,the CIC decimationfilter can have unity DC gain.6Sharpened CIC FiltersFilter sharpening can be used to improve the response of a CICfilter.This technique applies the samefilter several times to an input to improve both passband and stopband 1This paragraph is an generalization of equations found in the datasheet for the Harris/Intersil HSP50016.5charecteristics.If H(z)is a symetric FIRfilter,then a sharpened version,H S(z),can be expressed asH S(z)=H2(z)[3−2H(z)](19)The magnitude response of a sharpened CICfilter would then be|H(f)|=3sinπMfsinπf2N−2sinπMfsinπf3N(20)The interested reader is referred to[KJW97]for more details.Please note that it uses different parameters and implements a CICfilter a bit differently than[Hog81].7ConclusionSince their inception,CICfilters have become an important building block for DSP systems. They have found a particular niche in digital transmitters and receivers.They are currently used in highly integrated chips from Intersil,Graychip,Analog Devices,as well as other manufacturers and custom designs.This paper has attempted to summarize key points found in[Hog81]and provide some insight into designs.While many journal submissions are of limited value to an engineer,this paper was written for designers.As such,the reader should try to locate[Hog81]as the definitive reference for CICfilters.References[CR83]Ronald E.Crochiere and Lawrence R.Rabiner.Multirate Digital Signal Processing.Pretice-Hall Signal Processing Series.Prentice Hall,Englewood Cliffs,1983. [Fre94]Marvin E.Frerking.Digital Signal Processing in Communication Systems.Kluwer Academic Publishers,Boston,1994.[Hog81] E.B.Hogenauer.An economical class of digitalfilters for decimation and inter-polation.IEEE Transactions on Acoustics,Speech and Signal Processing,ASSP-29(2):155–162,1981.[KJW97]Alan Y.Kwentus,Zhongnong Jiang,and Alan N.Wilson,Jr.Application offilter sharpening to cascaded integrator-comb decimationfilters.IEEE Transactions onSignal Processing,45(2):457–467,1997.[OS89]Alan V.Oppenheim and Ronald W.Schafer.Discrete-Time Signal Processing.Pretice-Hall Signal Processing Series.Pretice-Hall,Englewood Cliffs,1989.6。

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Radio Ad Filtering with Machine LearningMichael HolmesJosh JonesAdam FeldmanApril 17, 2003 IntroductionOur group explored an application of machine learning techniques to the binary classification of audio samples. The learned binary classification is intended to indicate whether a given sample is taken from music or from another type of audio signal (such as commercials or DJ talking).The datasets used for learning and evaluating this classifier consist of pairs of audio samples and their correct classification. The audio samples were obtained by digitally recording two different Internet radio stations – WREK and 99x. This method enabled the convenient acquisition of a large quantity of suitable audio data in digital format. These recordings were then broken into fixed-length chunks (samples). Each sample was transformed using Mel Frequency Cepstral Coefficients (MFCCs) in order to generate feature vectors.Once the feature vectors were generated, a random subset of data was used as a training set. The system to be trained was an SVM classifier. Several different kernels were experimented with (see results). The classification output from the SVM was used to train an HMM. This allowed accuracy and confidence information, as well as other data (such as time series data), to be considered in classifying a sample.This problem is interesting because it would allow for several applications. For example, many radio stations turn up the volume of their advertisements, which can be an inconvenience, to say the least. Therefore, a successful classifier could be used to monitor the start of commercials, and adjust the volume accordingly. Another application would be to serve as a“station changer”, allowing a listener to automatically change stations to ensure that music is always being played. Thus, a listener could supply a list of favorite stations that could be scrolled through whenever a commercial begins. Further, constant sampling across all the stations could help ensure that a station is chosen on which a song has recently started (as opposed to constantly tuning to stations for the tail ends of songs).Although existing research has focused on attempting to classify audio samples in various ways, little work has focused directly on music/non-music discrimination. Our learning task specifically requires determining what is integral to music, versus other kinds of sounds. This is a subtle, but significant difference between our project and the reference work we located. Most previous work has been on speech/music or genre discrimination. As an example of how this difference can influence system design decisions, notice that features that may be very useful for making certain kinds of distinctions between audio samples may not be useful for making other kinds of distinctions. For example, it might require a much more complicated decision to distinguish pure music from speech, speech with music in the background, effects noises, etc. than to simply distinguish music from speech. We also consider this project to be interesting and worthy of exploration because, to this point, no finished product with the capabilities we envision has been produced. Whether this is because no one has succeeded in enabling a machine to discriminate between music and non-music (by employing machine learning techniques or through other means), or simply because the technology has not been appropriately adapted to be useful in these contexts, it is clear that there is still progress to be made in this area.Prior WorkThe decision to use an SVM to produce initial classification confidences was based on a search of relevant literature detailing existing work on audio sample classification. Although little of this work focused directly on music/non-music classification, we considered it likely that techniques successful in some audio classification tasks would enjoy similar success when applied to our task. In particular, previous work shows SVMs to be superior to boosting for classification of audio samples into 16 distinct categories [1]. A number of additional studies confirm the applicability of SVM learning to this domain area [2, 5]. One other technique, “Nearest Feature Line” (NFL) was also described in several works, and appears in some cases within the audio classification domain to be superior to SVMs in both training time and accuracy of the resulting classifier [3,4]. The general principle behind NFL learning is to classify new instances by calculating the distance in feature space between the feature vector to be classified and a line between each possible pairing of training points within each class. The classification chosen is then whichever class contains the pair of training points generating the line in feature space from which the new example has minimum distance. In this way, it is similar to other memory-based methods. However, due to our greater familiarity with SVMs, the availability of high-quality software implementing SVM learning, and the existence of research leading to good results using SVMs for audio classification, we decided upon the use of SVM learning for this project. A potential avenue for future work could involve the implementation of a system similar to the one designed in this project, instead using NFL. Then, a direct comparison would be possible in the context of this task, with the possibility of producing interesting results that contrast the capabilities of the two techniques. But, such a comparison is outside the scope of this work.Most previous audio classification work has not incorporated time series information in any principled way. Scheirer and Slaney [7] implemented a music/speech discriminator that used 2.4-second window averages to boost performance on a frame-by-frame basis. However, this technique is rather ad hoc. We decided to incorporate time series information in Bayesian style through an HMM. Further, because the SVM output can be looked at as a measure of confidence in the classification, we developed a modification to normal HMM inference that uses the degree of confidence in the classification at each time step. This is in some sense similar to the hybrid HMM/neural network approach of [8], but their system used neural networks to change the probability table of observations given state at each time step, while our approach was to use the SVM to measure our belief in the observation, keeping the state/observation probability table constant throughout.The majority of previous work in audio classification has involved some form of Mel Frequency Cepstral Coefficients (MFCCs), which are a form of frequency spectrum adapted to the human perception of pitch. Two papers [4, 9] specifically contrast MFCCs with other audio features, and find that MFCCs perform favorably. This, combined with the availability of open source MFCC extractors, was the basis of our decision to use MFCC feature vectors in this work.ExperimentsOur first step was to gather a large amount of audio data. This was obtained by recording approximately two hours worth of Internet radio broadcasts from two different music stations. These audio files were then separated and regrouped into music and non-music segments, yielding approximately 600,000 samples.Since MFCCs are a well-established audio feature, we were able to use existing utilities to extract feature vectors from our audio samples. Specifically, we used the Edinburgh Speech Tools utility called sig2fv [10]. This gave us feature vectors containing twelve MFCCs for each 0.01-second interval. We wrote a utility to convert the output of sig2fv into a format useable by SVM lite [6], a well-regarded, open source, C-based SVM package. The data was then separated into a training set (5% or approximately 30,000 of the samples) and a test set (everything not in the training set). Because training times were prohibitively long, we decided against the normal 33%/67% splitting of the data. Several SVM kernels were tested, including linear, quadratic, cubic, quartic, etc. The cubic was found have best performance, and so was chosen for use in the time series system.Our initial approach to the incorporation of time series information was to create by hand a simple HMM and use the SVM classification as the HMM’s observed symbol at each time step. The HMM design is shown in Figure 1. The symbol M means music, and N means non-music.232.0)(,768.0)(==N M ππO=M O=N S=M0.9875 0.757 S=N 0.01250.243Figure 1: The HMM with transition probabilities, initial state probabilities, and observation probabilities.The numbers for the HMM were derived empirically from the raw data and the SVM results, i.e. 999947=MS=P because 99.9947% of the time a music sample is MS(=.0)'|πbecause 76.8% of the samples are music, followed by a music sample, 768M)(=.0S=MOP because the SVM classified 98.75% of the test music samples M=9875.0)|(=correctly.In order to appropriately test the HMM, it was necessary to obtain labeled data points in their original sequence. Previously all samples of a given type were grouped together, so only the set had to be labeled. We therefore cut an approximately 40-minute segment of the audio recordings and parsed it into music/non-music chunks, extracted ordered features and classifications from the chunks, and reassembled them into the full mixed sequence that included transitions between music and non-music. The fastest way to generate the sequential test set was to draw from only one radio station and not filter out 5% of data points that had been part of the training set. Time did not permit the creation of a more sophisticated sequential test set; however, 5% of the data points could not have had a large impact on overall performance.HMM inference was performed using the standard Viterbi algorithm [11], taking the final state of the most probable path for each time step as the state output for that step. The state output at each step was compared to the known label, and accuracies were computed for music and non-music classification. As we observed that music and non-music occur in large contiguous blocks, we added a domain-specific enhancement to the HMM: transitions out of a state were only allowed when the probability of being in the other state was higher than a threshold (0.9 to go from non-music to music, 0.55 to go from music to non-music; the reason for the asymmetry is that spurious SVM outputs are more common on non-music than on music samples).Finally, noting that the SVM output was not just a binary decision but the actual margin of each data point, we developed a modification to the HMM that used the SVM margin as a confidence measure in the correctness of the observation (M or N). This confidence measure was transformed into a probability that the observation is correct as follows. Since most margins had absolute value less than 1.5, the first confidence measure was either the absolute value of the margin if it was less than 1, or truncated to 1 if larger. This confidence measure alone was not a good estimate of the probability that the SVM gave the correct output; for example, a confidence of 0.2 that the sample is music does not mean that the probability of being non-music is 0.8. The key observation here is that a confidence of 0 should result in a 0.5 probability that the observation is correct, and a confidence of 1 should give a probability of 1. This leads to the formula)1(5.0)(confidence n Observatio Correct P +⋅=.So, for example, if the SVM classifies a sample as M with a certain confidence, then )1(5.0)(confidence M P +⋅= and )(1)(M P N P −=. Normally in Viterbi inference, the probabilities of the most probable path ending in state M would be updated as:)|(*),(*)(max )(,1M n Observatio P M j T j S P M S P t NM j t ====+. This assumes the observation is correct. We incorporate the possibility that the observation is incorrect by updating as follows:[][].)|(),()(max ))(1()|(),()(max )()(,1N n Observatio P M S T S P M Obs P M n Observatio P M j T j S P M Obs P M S P j jt j t N M j t ⋅⋅⋅=−+⋅⋅=⋅====+In other words, the new state probability is a linear combination of the normal Viterbi updates for each possible observation, weighted in proportion to the estimated probability that each observation was the correct one. We used the same state change thresholding in this modifiedHMM as we did in the normal HMM and again calculated accuracies for each class using the data labels.ResultsSeveral different SVM kernels were tested, with the results shown in Figure 2. The model dimensionality refers to the kernel used, ranging from a linear model (1), to a squared model (2), then a cubed model (3) and finally a quartic model (4). As the figure shows, the cubed model (3) has the best overall results. Specifically, this model correctly classified 98.8% of the tested music samples and 24.3% of all non-music samples, for an overall accuracy of 84.1%, which is higher than any other model in each measurement category. This was the basis of our decision to use the cubic model in the time series experiments.Figure 2: Test set accuracy for polynomial kernels of various dimensionalities.The SVM seemed adept at correctly classifying music samples, but performed very poorly on the non-music samples. Fortunately, this performance was increased greatly by the introduction of time series information. The accuracy results from the normal and modified HMMs are summarized with the original SVM accuracy in Table 1.Music Accuracy Non-Music Accuracy Overall Accuracy Static SVM 98.8% 24.3% 84.1% Normal HMM 88.6% 86.0% 88.0%Modified HMM 94.6% 73.5% 89.8%Table 1: Accuracy of normal and modified HMMs over the sequential test set. Note that the overall accuracy is not simply the average of the other two accuracies because there are more music than non-music data points.As can be seen, both types of HMM resulted in a relatively small decrease in the static music accuracy of 98.8%, while yielding very large increases over the 24.3% static non-music accuracy. Both HMMs outperformed the static SVM in overall accuracy, and the modified HMM performed 1.8% better than the normal HMM.ConclusionsAs the results of the experiment indicate, our method of classifying between music and non-music audio samples is successful. We have implemented a combination of SVM and HMM in order to provide the best possible chances of correctly identifying each sample. In this way, we have achieved an overall accuracy of 89.8%.Initially, MFCC is used to create a feature vectors for each audio sample. These feature vectors are then used to train a cubic SVM model. The SVM outputs a music or non-music margin, which we use for one-shot static sample classification, for normal HMM classification ina time series, and as a proto-confidence level in a modified HMM that uses a new version of Viterbi inference that takes into account uncertainty in the observation.We found that using the SVM alone produced very good results for classifying music samples (over 98%). However, the results of classifying non-music samples are significantly worse, at less than 25%. This averages to create an overall accuracy of approximately 84%. While this result is good on some level, the introduction of an HMM into the system yielded much better classifications. Specifically, the normal HMM had 88% accuracy, while our modified HMM technique yielded 90% accuracy. This is an especially interesting point, because the modified HMM technique is not at all domain specific, and the combination of a confidence-outputting classifier with the modified HMM technique could be a means of improving HMM performance in general.In the end, these results show promise, and indicate that these methods can be successful at classifying audio samples as music or non-music. Further, these results can possibly be improved upon through the addition of yet other techniques and/or making refinements to the techniques we employed. See below for possible future work and extensions to the project.Future WorkThere are several possible extensions to this project, some involving practical application of the classifier generated as the result of these experiments, and others focusing on improving the classifier itself by achieving a better understanding of applicable theories and methods within the context of binary audio classification. Ultimately, one goal would be to create a system for Internet radio listeners. This system would allow the user to define a series of favorite radio stations. Each radio station would be monitored for content – music vs. non-music. In this way,whenever a station is not playing music, it would be switched, in favor of a station that is playing music. Additionally, stations on which a song has recently started will be given priority over stations on which a song has been playing for longer (and therefore is nearer the end).Further, this concept could be extended, providing a similar service on television. In this domain, commercials are much more prone to increasing volume. Therefore, instead of changing the station, this system could be designed to automatically reduce the volume of the television. Taking this approach, as opposed to simply trying to constantly moderate the volume, would help prevent adjusting the volume from scene to scene in a given show or movie (which should not be interfered with).Another area of future interest involves different types of music. These techniques of discrimination should be examined in the context of classifying types of music (country vs. rap, for example). A motivation for this research would be to allow a listener to choose between channels based on type of music. Presumably in addition to removing commercials, this would ensure that a listener could listen to only the desired type of music, regardless of the fact that many stations play a variety of types.The general technique of combining a confidence-outputting classifier such as an SVM with the modified HMM inference is one that merits exploration in other domains. Testing in other areas and with other classifiers would lead to the discovery of whether this technique is a general improvement over regular HMMs or not.Any of these applications would require a broadening of the training set and improvement of the validation strategy. Alternative strategies for classification and feature extraction/selection could also be explored, possibly further improving the quality and practicalapplicability of the current classifier beyond the positive results that have been achieved as part of this study.References[1] Guo, Zhang and Li, "Boosting For Content-Based Audio Classification And Retrieval: An Evaluation," /503362.html[2] Lu, Li and Zhang, "Content-Based Audio Segmentation Using Support Vector Machines," /502778.html[3] Chen K., Wu T.Y., and Zhang H.J., “On the Use of Nearest Feature Line for Speaker Identification,” Pattern Recognition Letters 23: 1735-1746, (2002),.hk/~apnna/proceedings/iconip2001/papers/041a.pdf[4] Li, S. Z., "Content-Based Audio Classification and Retrieval Using the Nearest Feature Line Method," IEEE Transactions on Speech and Audio Processing, Vol. 8, No. 5, pp. 619--625, Sept. 2000, /~mjr59/reviews/nearest_line.pdf[5] S. Z. Li and G. Guo, “Content-based Audio Classification and Retrieval Using SVM Learning,” (invited talk), PCM, 2000,/china/papers/Content_Audio_Classification.pdf[6] T. Joachims, “Making large-Scale SVM Learning Practical. Advances in Kernel Methods - Support Vector Learning,” B. Schölkopf and C. Burges and A. Smola (ed.), MIT-Press, 1999, [7] E. Scheirer and M. Slaney, “Construction and Evaluation of a Robust Multi-featureSpeech/Music Discriminator,” Proceedings of the 1997 International Conference on Acoustics, Speech, and Signal Processing, 1997.[8] H. Franco, M. Cohen, N. Morgan, D. Rumelhart and V. Abrash, "Context-dependent connectionist probability estimation in a hybrid Hidden Markov Model – Neural Net speech recognition system," Computer Speech and Language, Vol. 8, No. 3, 1994.[9] B. Logan, “Mel Frequency Cepstral Coefficients for Music Modeling,” International Symposium on Music Information Retrieval, 2000.[10] P. Taylor, R. Caley, A. W. Black, S. King, “Edinburg Speech Tools Documentation,” 1999, /docs/speech_tools-1.2.0/.[11] “University of Leeds HMM Tutorial – Viterbi Algorithm,” /scs-only/teaching-materials/HiddenMarkovModels/html_dev/viterbi_algorithm/s1_pg1.html.。

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