英飞凌第四代REAL3
第1章嵌入式系统概述
2、SiM3U1xx（80MHZ USB）系列（M3）
1.4 STM32系列微控制器简介 STM32为意法半导体(ST)公司生产的ARM处理器。
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1.嵌入式系统简介
目前，对嵌入式系统的定义多种多样，但没有一种定义是全面的。下面给出两种 比较合理定义：
●从技术的角度定义：以应用为中心、以计算机技术为基础、软件硬件可裁剪、 适应应用系统对功能、可靠性、成本、体积、功耗严格要求的专用计算机系统。 ●从系统的角度定义：嵌入式系统是设计完成复杂功能的硬件和软件，并使其紧 密耦合在一起的计算机系统。术语嵌入式反映了这些系统通常是更大系统中的一 个完整的部分，称为嵌入的系统。嵌入的系统中可以共存多个嵌入式系统。
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IGBT保护分析(英飞凌)
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Th
TC
Heat conducting paste
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The correction factor could be different for different modules. It depends on the position of the NTC.
22.05.2007 For internal use only Copyright © Infineon Technologies 2007. All rights reserved. Page 4
Th - depends on the cooling capability and design for the power dissipation based on the real application. Tc – depends on case temperature, and determined by the nominal current definition (Tc=100oC,or 85oC) Tj - max. allowable operational Tvjop (125oC/150oC).
Current Ic Voltage Vce
Do Not Switch Off At This Moment
22.05.2007 For internal use only Copyright © Infineon Technologies 2007. All rights reserved.
Switch Off the SC!
22.05.2007 For internal use only Copyright © Infineon Technologies 2007. All rights reserved. Page 11
经典英文广告语120则经典英文广告语
经典英文广告语120则经典英文广告语经典英文广告语120则经典英文广告语 1．good to the last drop. 滴滴香浓，意犹未尽。
（麦斯威尔咖啡）2．obey your thirst.服从你的渴望。
（雪碧）3．the new digital era.数码新时代。
（索尼影碟机）4．we lead others copy.我们领先，他人仿效。
（理光复印机）5．impossible made possible.使不可能变为可能。
（佳能打印机）6．take time to indulge.尽情享受吧！（雀巢冰激凌）7．the relentless pursuit of perfection.不懈追求完美。
（凌志轿车）8．poetry in motion, dancing close to me.动态的诗，向我舞近。
（丰田汽车）9．come to where the flavor is marlboro country.光临风韵之境——万宝路世界。
（万宝路香烟）10．to me, the past is black and white, but the future is always color.对我而言，过去平淡无奇；而未来，却是绚烂缤纷。
（轩尼诗酒）11．just do it.只管去做。
（耐克运动鞋）12．ask for more.渴望无限。
（百事流行鞋）13．the taste is great.味道好极了。
（雀巢咖啡）14．feel the new space.感受新境界。
（三星电子）15．intelligence everywhere.智慧演绎，无处不在。
（摩托罗拉手机）16．the choice of a new generation.新一代的选择。
（百事可乐）17．we integrate, you communicate.我们集大成，您超越自我。
（三菱电工）18．take toshiba, take the world.拥有东芝，拥有世界。
英飞凌SPWM和SVPWM的讲解
AP16097XC164Different PWM Waveforms Generation for 3-Phase AC Induction Motor with XC164CSMicrocontrollersEdition 2006-08-04Published byInfineon Technologies AG81726 München, Germany©Infineon Technologies AG 2006.All Rights Reserved.LEGAL DISCLAIMERTHE INFORMATION GIVEN IN THIS APPLICATION NOTE IS GIVEN AS A HINT FOR THE IMPLEMENTATION OF THE INFINEON TECHNOLOGIES COMPONENT ONLY AND SHALL NOT BE REGARDED AS ANY DESCRIPTION OR WARRANTY OF A CERTAIN FUNCTIONALITY, CONDITION OR QUALITY OF THE INFINEON TECHNOLOGIES COMPONENT. THE RECIPIENT OF THIS APPLICATION NOTE MUST VERIFY ANY FUNCTION DESCRIBED HEREIN IN THE REAL APPLICATION. INFINEON TECHNOLOGIES HEREBY DISCLAIMS ANY AND ALL WARRANTIES AND LIABILITIES OF ANY KIND (INCLUDING WITHOUT LIMITATION WARRANTIES OF NON-INFRINGEMENT OF INTELLECTUAL PROPERTY RIGHTS OF ANY THIRD PARTY) WITH RESPECT TO ANY AND ALL INFORMATION GIVEN IN THIS APPLICATION NOTE.InformationFor further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office ().WarningsDue to technical requirements components may contain dangerous substances. For information on the types in question please contact your nearest Infineon Technologies Office.Infineon Technologies Components may only be used in life-support devices or systems with the express written approval of Infineon Technologies, if a failure of such components can reasonably be expected to cause the failure of that life-support device or system, or to affect the safety or effectiveness of that device or system. Life support devices or systems are intended to be implanted in the human body, or to support and/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the healthAP99007Revision History: 2006-07 V1.0 Previous Version: nonePage Subjects (major changes since last revision)Table of Contents Page1 Introduction (5)2 3-Phase AC Induction Motor Control Principle (6)2.1 Basic Operation Theory (6)2.2 VVVF Control (6)3 Different PWM schemes (8)3.1 General Theory of PWM (Pulse Width Modulation) (8)3.2 SPWM (Sinusoidal PWM) (8)3.2.1 Basic Principle (8)3.2.2 Implementation Method (9)3.2.3 Modulation Index (10)3.2.4 3-Phase SPWM (11)3.3 THIPWM (Third-Harmonic Injection PWM) (11)3.4 SAPWM (Saddle-wave PWM) (12)3.5 SVPWM (Space Vector PWM) (12)4 XC164CS Implementation (14)4.1 CAPCOM6 Unit Introduction (14)4.1.1 Overview (14)4.1.2 T12 Center-Aligned Mode (14)4.1.3 PWM Signals Generation (15)4.2 CAPCOM6 Initialization (16)4.3 Programming Consideration and Flow Charts (16)5 Experiment Results (18)5.1 Simulation Results (18)5.2 Oscilloscope measured waveforms (19)6 Conclusions (21)Introduction1 IntroductionIn this application note, the methods to generate different PWM waveforms for 3-phase AC induction motor using an Infineon 16-bit microcontroller XC164CS are introduced.For better understanding, the basic operation and control principle of 3-phase AC induction motors is described in Section 2. The content of Section 3 is the respective explanation of the theory of four popular PWM schemes frequently utilized in AC induction motor control, SPWM (Sinusoidal PWM), THIPWM (Third-Harmonic Injection PWM), SAPWM (Saddle-wave PWM), SVPWM (Space Vector PWM). In Section 4, XC164CS DAvE configurations and codes programming for different PWM schemes realization are discussed in detail and illustrated with flow charts. At the end of this article some experimental results including simulator waveforms and figures measured by oscilloscope are shown to validate the algorithms and some conclusions are drawn.Furthermore, two sets of source codes for SPWM and SVPWM with 10KHz carrier frequency and output frequency from 0 to 50Hz in 10 seconds respectively as well as an EXCEL file on calculation tables for all PWM modes are attached.3-Phase AC Induction Motor Control Principle 23-Phase AC Induction Motor Control Principle 2.1 Basic Operation TheoryThe 3-phase stators and 3-phase rotors are considered as two fundamental parts of a 3-phase AC induction motor. When the 3-phase stators are energized by the 3-phase AC power source, current flow is generated in the stators. The magnetic field synthesized by 3-phase stator current is always rotating incessantly with the variation of the current. This rotating magnetic field cuts the rotor and the current generated in it interacts with the rotating magnetic field and thus produces the magnetic torque which makes the rotor rotate (Figure 1). The rotating speed of the rotor n should be less than that of the rotating magnetic field n 0. Reverse rotating of the rotor will be realized by two of the 3-phase power source positions exchanged.Figure 1 Rotating start for AC induction motorThe rotating direction of the rotating magnetic field is consistent with the current phase and its speed is proportional to the power source frequency f and inversely proportional to the magnetic polar pair number P . Calculated per minute, the speed of the rotatingmagnetic field n 0 can be represented by this equation: pf n 600= (E-1) Where f is the frequency of power source and P is the magnetic polar pair number.2.2 VVVF ControlFrom the equation (E-1) two primary methods for speed control of 3-phase AC induction motor can be concluded: one is to change the magnetic polar pair numbers but the inflexibility and low efficiency of this method has limited its popularity of application. Another method is to regulate the stator current frequency. Usually a principle of popular practical implementation called “VVVF” is adopted on speed regulation.The 3-phase stators cutting the flux of the rotating magnetic field results in the back electromotive force generated and it can be calculated by the equation given below: M r N f k E Φ=111144.4 (E-2)3-Phase AC Induction Motor Control Principle Where k r1 is the winding structure related constant and N 1 is the number of turns of the stator winding per phase, f 1 is the stator current frequency, M Φis the main flux. Let aconstant 11144.4N k K r E =, we haveM E f K E Φ=111 (E-3) Since the voltage drop on the stators impedance only occupies relatively very small portion of the whole stator voltage U 1 and can be ignored, therefore11E U ≈ (E-4) Derived From (E-3) and (E-4), it holds 11f U K M Φ=Φ (E-5) Where 11E K K =Φ is also a constant. From (E-5) it can be concluded that if the value of 11/f U can be controlled to be a constant, M Φ remains unchanged. This control method toregulate frequency with voltage changed accordingly is usually called “VVVF”, i.e. Variable Voltage Variable Frequency.Different PWM Schemes for 3-Phase AC Induction Motor 3 Different PWM Schemes for 3-Phase AC Induction Motor3.1 General Theory of PWM (Pulse Width Modulation)PWM (Pulse Width Modulation) technology was put forward based on an important conclusion in the sample control theory that when two groups of pulses with the same impulse area but different waveforms are input to an inertial link, the effectiveness of these two groups of impulses are the same. The main principle of PWM technique can be briefly described as: Through ON/OFF control on the semiconductor switching components, a series of pulses with the same amplitude and different width are generated on the output port to replace the sinusoidal wave or other waveforms required. The duty cycle of the output waveform needs to be modulated by a certain rule and as a result both the output voltage and output frequency of the inverter can be regulated.The signals before PWM and after PWM are shown in Figure 2 and Figure 3 respectively. Compared with Figure 2, the frequency of the signal in Figure 3 is increased, the amplitude remains unchanged and therefore the average value of the signal is decreased. Therefore PWM just meets the requirement of VVVF described in section 2.2 and is adopted as the general method for AC induction motor control.Figure 2 Pulse before PWMFigure 3 Pulse after PWM3.2 SPWM (Sinusoidal PWM)3.2.1 Basic PrincipleAmong all PWM schemes, SPWM is one of the most popular and simple methods utilized in power inverter and motor control fields. Its main features can be summarized as sine-triangle wave comparison.As shown in Figure 4, a sine wave (modulated wave, magenta) is compared with a triangle wave (carrier wave, green) and when the instantaneous value of the triangle waveDifferent PWM Schemes for 3-Phase AC Induction Motor is less than that of the sine wave, the PWM output signal (orange) is in high level (1). Otherwise it is turned into the low level (0). The level switching edge is produced at every moment the sine wave intersects the triangle wave. Thus the different crossing positions result in variable duty cycle of the output waveform.Figure 4 SPWM Waveform Generation3.2.2 Implementation MethodIn terms of the basic principle of SPWM illustrated above, it’s easy to implement using analog circuit (Figure 5). Sine and triangle waves are respectively generated by specially designed circuits and then fed to the properly selected comparator which can output the desired SPWM signal. But the control precision and reliability of this scheme are always not so satisfying due to the complicated circuit structure as well as the instability of the parameters of all analog devices.With the development of the microcontroller, nowadays the software implementation for SPWM is absolutely mostly adopted to realize high precision control.Figure 5 Analog Scheme for SPWM ImplementationThe method utilizing the natural intersection points of sine wave with triangle wave to realize PWM is called “Natural Sampled Method”. It’s able to demonstrate the true moments the pulse is started and ended and the SPWM waveform is much closer to sineDifferent PWM Schemes for 3-Phase AC Induction Motor wave. This method is not adopted in most control applications due to the random intersection points of sine and triangle wave which results in complicated calculation and difficult real-time implementation.To overcome these disadvantages, another new method called “Regular Sampled Method” was put forward. It is widely used in engineering applications nowadays. It is based on the principle that a certain moment is selected in every cycle of the triangle carrier wave to find the corresponding value of the sine wave voltage which is introduced to sample on triangle wave and the sample result determines the ON/OFF moments of the power devices, ignoring whether the sine wave and triangle wave intersects in this moment or not. A more practical method named “Average Symmetric Regular Sampled Method” (illustrated in Figure 6) is applied in most control cases. In Figure 6, the sampled moment is given on the trough point of the triangle wave, then centered by the corresponding value of sine wave voltage, a horizontal line is drawn to intersect the triangle wave on both sides so the leading and trailing edges of PWM waveform are decided upon that. The leading edge is a little wider which just compensates the narrow trailing edge and therefore as an average consideration the effectiveness of this method is almost equivalent to that of the natural sampled method.Figure 6 Symmetric Regular Sampled Method3.2.3 Modulation IndexWhen the amplitude of the modulated sine wave is larger than that of the carrier triangle wave, over modulation occurs. Once the sine wave reaches the peak of the triangle, the PWM pulses will obtain the maximum width so the modulation will enter the state of saturation (Figure 7). Therefore the item “Modulation index” (represented by m) defined by the ratio of the amplitude of the modulated wave to that of the carrier wave is introduced to describe the modulation state. When 0<m<1, the linear relationship between the input and PWM output voltage is maintained. If the value of modulation index exceeds 1, this linear mode cannot be kept any more and the special control strategy for over modulation is required.Different PWM Schemes for 3-Phase AC Induction MotorFigure 7 SPWM Saturation3.2.4 3-Phase SPWMFor 3-phase AC induction motor control system, the SPWM signals to trigger the six power switches in the voltage source inverter is generated by comparison of the 3-phase sine waves with the same triangle wave (Figure 8).Figure 8 3-phase SPWM Waveforms (Pspice Simulation)3.3 SPWM (Sinusoidal PWM)If certain portion of the third harmonics wave is injected into the sine wave, the resulted modulated wave will appear as saddle-like shape (Figure 9) and the amplitude willDifferent PWM Schemes for 3-Phase AC Induction Motor obviously decrease. With keeping m <1, the amplitude of the fundamental wave can exceed that of the triangle wave and thus utilization ratio of the DC-bus voltage increases.Figure 9 3-phase THIPWM Modulated Wave3.4 SAPWM (Saddle PWM)SAPWM is one type of the optimized PWM methods and its modulated wave (Figure 10) can be exactly described by the mathematic equation (E-6): ⎪⎪⎩⎪⎪⎨⎧<<+<<=23 ),3sin(30 ,sin 3)(1111πωππωπωωt t t t t y (E-6) It can be concluded from the research results that the maximum output voltage of the inverter adopting SAPWM can reach the value of the input voltage of the electric net which is 15% higher than that of the inverter using SPWM. Furthermore, the SAPWM inverter has been improved in restraining harmonic current, reducing torque fluctuation andenhancing output torque.Figure 10 3-phase SAPWM Modulated Wave3.5 SVPWM (Space Vector PWM)Based on the 3-phase integrated generation effectiveness and for the purpose of approaching the ideal rounded track with constant amplitude of the rotating field formed by the gap flux, SVPWM waveform is realized by the combination of different switching modes of the inverter. In a 3-phase inverter, if “1” is defined as the positive half of the DC-bus voltage and “0” as the negative half (both are referred to the neutral point), there are totally 8 switch states for the six power switches (Figure 11). Therefore 8 voltage vectors (active vectors 1U r ~6U r and zero vectors 0U r , 7U r ) can be correspondingly defined to formDifferent PWM Schemes for 3-Phase AC Induction Motor the vector space which is divided into 6 sectors (Figure 12).1117=S0000=S 1005=S 1016=S 0011=S 0112=S 0103=S 1104=SFigure 11 Eight Switch States111=S r 0000=S r 1005=S 1016=S 0011=r r r 1104=S r 1r U rFigure 12 Voltage Vector SpaceThe voltage vector U r is generally decomposed into two nearest adjacent voltage vectors with zero vectors 0U r and/or 7U r as supplement. Thus the vectors in the six sectors I U r ~ VI U r are relatively calculated (assume that the zero vector operation time is halved by 0U r and 7U r ) as the equations (E-7):Different PWM Schemes for 3-Phase AC Induction Motor ⎪⎪⎪⎪⎪⎪⎪⎪⎩⎪⎪⎪⎪⎪⎪⎪⎪⎨⎧+−−++=+−−++=+−−++=+−−++=+−−++=+−−++=)(2)(2)(2)(2)(2)(2701611667065665570545544704344337032332270212211U U T T T T U T T U T T U U U T T T T U T T U T T U U U T T T T U T T U T T U U U T T T T U T T U T T U U U T T T T U T T U T T U U U T T T T U T T U T T U s s s s VI s s s s V s s s s IV s s s s III s s s s II s s s s I r r r r r r r r r r r r r r r r r r r r r r r r r r r r r r (E-7) For detailed SVPWM calculation and more related information, please refer to another application note (AP0803601, Title: “Space Vector Modulation and Over-Modulation with an 8-bit Microcontroller”, /microcontroller->Application Notes -8-bit Microcontrollers).XC164CS Implementation of Different PWM Generation 4 XC164CS Implementation of Different PWM Generation4.1 CAPCOM6 Unit Introduction4.1.1 OverviewThe CAPCOM6 unit of XC164CS provides 2 independent timers T12 and T13 for PWM signals generation, especially for AC induction motor control. Its block diagram is shown in Figure 13.Figure 13 CAPCOM6 Block DiagramThere are 3 capture/compare channels for Timer12 (16-bit) and each channel can be used either as capture or compare channel. Generation of a 3-phase Center- or edge-aligned PWM signals with dead-time control for each channel to avoid short-circuits in the power stage is supported (6 outputs, individual signals for lowside and highside switches) by T12. Timer13 (10-bit) has one independent compare channel with one output. It can be synchronized to T12 and supports single-shot mode. Fast emergency stop without CPU load via external signal CTRAP and related software interrupt process are both supported by CAPCOM6 unit to ensure the reliable protection under unexpected faults.4.1.2 T12 Center-Aligned ModeT12 block is the main unit to generate the 3-phase PWM. A 16-bit counter is connected to 3 channel registers via comparators, which generate a signal when the counter contents match one of the channel register contents. Besides the 3-phase PWM generation, the T12 block offers options for individual compare and capture functions as well as dead-time control.T12 can operate in Edge-Aligned mode or Center-Aligned mode. In this article the Center-Aligned mode is adopted in PWM implementation and its operation principle is illustrated in Figure 14.XC164CS Implementation of Different PWM GenerationFigure 14 T12 Center-Aligned Mode4.1.3 PWM Signals GenerationAs shown in Figure 15, each channel of T12 is connected to the T12 counter register via its individual equal-to comparator, which generates a match signal (CC6x_O) when the contents of the counter (CC6_T12) matches those of the associated compare register (CC6xR).Each channel consists of the comparator and a double register structure - the actual compare register CC6xR feeds the comparator and an associated shadow register CC6xSR is preloaded by software and transferred into the compare register when T12 shadow transfer (T12_ST) becomes active.CC6xST is a State Bit which holds the compare operation status of each channel. Bit CC6xPS/COUT6xPS selects the state of each channel, considered as the passive state during which the passive level (defined in register PSLR) is driven by the output pin: “0” represents that the compare output drives passive level while CC6xST is 0 and “1” defines that the compare output drives passive level while CC6xST is 1.Figure 15 PWM Signals Generated by T12XC164CS Implementation of Different PWM Generation 4.2 CAPCOM6 InitializationThe general initialization of CAPCOM6 for all the PWM schemes discussed in this article is summarized below (according to display order in DAvE): “Module Clock”: Enable module. “Pin Control”: Use pin CC60, CC61, CC62, COUT60, COUT61, COUT62 as output. “T12”: fcpu/4 (Resolution: 0.100us); Center-Aligned mode; T12 period 100us(carried frequency 10KHz); Start T12 after initialization; Enable interruptfor T12 period match (generating interrupt per carrier cycle). “T13”: No initialization is required for it, T13 isn’t used here. “Multi Ch.”: Disable multi-channel mode. “Channels”: Channel 0,1,2 should be individually configured as (x=0,1,2): CompareMode 3 (Use pins CC6x/COUT6x as output); Enable T12 modulation forCC6x; The compare output CC6x drives passive level while CC6xST is“0”; The compare output COUT6x drives passive level while CC6xST is“1”; The passive level of CC6x and COUT6x output are all “1”; Enabledead time generation. “Trap/INT”: The demo code is just focus on PWM algorithm implementation andverification, CTRAP function is not necessary to be configured. In“Interrupt Configuration” select “Enable T12 interrupts / node I2(IE)”. “Interrupts”: CCU6 I2 INT -> Level 15, Group 0 (For convenience it’s set to thehighest priority.) “Functions”: In “Initialization Function” select “CCU6_vInit”.4.3 Programming Consideration and Flow ChartsHere in all these PWM schemes 10KHz carrier frequency is adopted and thus T12PR value is 1F3H (499D ) with resolution set to 0.1us. As a result 499D is regarded as the PWM cycle value in the code programming and pulse width table calculation.For SPWM/THIPWM/SAPWM, all the modulated waveforms can be described with mathematic equations so pulse width tables can be directly pre-calculated for look-up. Here the 2π modulated wave is divided into 3000 points and therefore the 3-phase look-up pointers have 1000 points difference with each other. In the T12 period match interrupt service program, the increment of the 3-phase look-up pointers P inc are determined by: out PWM PWM-out carrier PWM-out PWM-out carrier inc F F F F T T P −=×=×=×=31010000300030003000 (E-8) The 3-phase look-up pointers are updated respectively with P inc increment every T12 Period-Match Interrupt. Hence the pulse width can be obtained by using these pointers to look up the precalculated table. Please refer to the attached EXCEL file for more details about the look-up table generation. It’s easy to generate THIPWM or SAPWM waveforms directly using their look-up tables to replace SPWM table in the attached SPWM demo code files.About detailed SVPWM algorithm programming, please refer to another two applicationXC164CS Implementation of Different PWM Generation notes (AP0801701, Title: “XC866 Constant V/f Control of Induction Motors Using Space Vector Modulation”; AP0803620, “Optimized Space Vector Modulation and Over-modulation with the XC866”. /microcontroller->Application Notes -8-bit Microcontrollers)Figure 16 shows the general flow chart of the T12 Period-Match Interrupt service subprogram in all these PWM schemes.Figure 16 Flow Chart of T12 Period-Match Interrupt Service RoutineExperiment ResultsResults5 ExperimentResults5.1 SimulationFigure 17 shows the simulated 3-phase SVPWM waveforms with dead-time control (P1L.0/CC60, P1L.1/COUT60, P1L.2/CC61, P1L.3/COUT61, P1L.4/CC62, P1L.5/COUT62) using KEIL uVision3 simulator.(a) 3-phase SVPWM signals(b) 3-phase SVPWM signals (Zoomed-in)(c) dead-time controlFigure 17 SVPWM Simulated WaveformsExperiment Results 5.2 Oscilloscope Measured WaveformsThe modulated waveforms of 3-phase SPWM, THIPWM (with 1/4 and 1/6 amplitude third harmonics injection respectively), SAPWM and SVPWM measured by oscilloscope are individually shown in Figure 18. All the signals are measured after being filtered respectively with three 3.3KΩ resistors and eight 1uF capacitors between CC6x/COUT6x and ground.Figure 19 displays the zoomed-in PWM output waveforms (directly measured between CC6x/COUT6x and ground) and the dead-time for these PWM schemes. For SPWM, THIPWM (1/4), THIPWM (1/6), SAPWM, SVPWM, the dead-time is correspondingly set to 1us, 2us, 3us, 4us, 5us.Figure 18 Modulateds Waveforms for Different PWM SchemesExperiment ResultsFigure 19 Output Signal for Different PWM SchemesConclusions 6 ConclusionsThe implementation of different PWM schemes for 3-phase AC induction motor control via Infineon 16-bit MCU XC164CS is discussed in this article. The high-performance CAPCOM6 unit dedicated for motor control provides an easy and fast way to realize various types of 3-phase PWM signals generation.。
英飞凌变频器设计IGBT选型指南-仿真工具
Dimensioning program IPOSIM for loss and thermal calculation of Infineon IGBT modules
Introduction IPOSIM performs an approximate calculation of switching and conduction losses for IGBTs and free-wheeling diodes in a three phase inverter configuration under the assumption of sinusoidal output currents at inductive loads. With this tool a quick selection of a suitable Infineon IGBT module for an application is possible taking into account its average losses and thermal ratings. Be sure to always have the latest IPOSIM version on-hand. The actual program is available on
T0 / 2
Psw,IGBT = f sw,IGBT ⋅
1 T0
∫ (E
0
on
+ E off )( t, ˆ i )dt
Using the measured turn-on and turn-off energy dissipation per switching pulse (given in the datasheets at nominal current Inom) the energy of the single switching event at a temporary current i can be assumed linear. Furthermore the applied DC-link voltage at several applications may vary from the nominal DC voltage used for the determination of the losses. The practice shows, that a linear adjustment of the losses within a certain limit of the nominal voltage (here ± 20% ) is permissible.
汽车的电动化、智能化、网联化
责任编辑：王莹汽车的电动化、智能化、网联化何芳 （英飞凌汽车电子事业部大中华区营销总监）1 汽车的电动化、智能化、网联化趋势英飞凌认为，汽车的电动化、智能化、网联化趋势将带动车内半导体含量的大幅增长。
电动汽车日渐流行，ADAS 渗透率稳步提升，用户对舒适性及驾乘体验的追求日益提高，都离不开半导体技术的支持。
而这三方面正是英飞凌汽车半导体业务所关注的核心应用。
1)在汽车电动化方面，核心痛点在于续航能力、动力性能和充电时间。
行驶里程不仅取决于电池的容量和性能，也跟整车系统的能源管理水平密切相关，特别是高性能的电机和电控系统。
这其中，功率半导体是电控系统的核心，主要包括IGBT 和MOSFET 。
硅基IGBT 技术相对比较成熟，市场竞争的重点在于产品性能的稳定性和可靠性。
业界的趋势是定制化模块封装以及双面冷却集成，以进一步提升IGBT 模块的综合性能。
虽然S i C 器件成本较高，但随着成品率和原材料利用率的提高以及SiC 对于整车系统的贡献，SiC MOSFET 的应用将很快在系统成本上取得优势。
保守预计至2025年，碳化硅技术在汽车电子功率器件领域的渗透率将超过20%2)ADAS 和智能网联方面，随着L2+到L3的演进，主要挑战在于法规和车内系统复杂性的增加。
从技术上来讲，在目前常见的三种传感技术中（摄像头、激光雷达和毫米波雷达），激光雷达的综合性能最优。
但无论采用哪种技术路线，都离不开高性能传感器以及传感器融合技术，同时还需要应对功能安全及信息安全等方面的新挑战。
3)在舒适性与用户体验方面，在网联化的驱动下，车身互联及安全性都需要很多新技术，以及很好的产品技术融合。
2 英飞凌的解决方案英飞凌是汽车功率半导体重要的供应商，在IGBT 和SiC MOSFET 方面有着深厚的技术积累和坚实的市场地位。
英飞凌在SiC 技术领域拥有25年的发展经验，针对多种新能源汽车系统已推出广泛的SiC 解决方案以及全方位的车规级产品系列，包括CoolSiC™车用肖特基二极管，CoolSiC™车用MOSFET ，全SiC 模组的HybridPACK™ Drive 等。
DECS-Ⅰ柴油机集成电子控制系统的设计
柴油机 Diesel Engine
Vol.28 （2006） No.3
电控与监测
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柴油机集成电子控制系统的设计
钢 !，庄 飚"
邱爱华 !，刘佳彬 !，李过房 !，吴建平 !，艾
（1. 七一一研究所，上海 200090； 2. 海军装备研究院舰船所，北京 100073） 摘 要：随着数字技术的进步，柴油机已越来越朝集成电子控制技术方向发展。在继承已有模 拟式电子调速器及
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自 20 世纪 60 年代开始，人们研制成 功了各
种形式的模拟式电子调速器。例如，美国 WOODWARD 公司 的 2301 EG-3P 型电 子调速 器、美 国 BARBER-COLMAN 公司的 DYN A 系 列电子调 速
收修改稿日期： 2006-01-25
2006 年 5 月
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:2; <,)=5!> ?2; @,5A,.!> ?2 B=/05.-!> C ; @,5.D,.!> <2 B5.-!> EF;<GB H,5/" （1. Shanghai Marine Diesel Engine Research Institute, Shanghai 200090; 2. Institute of Naval Academy of Armament, Beijing100073） <A+34573I With the improvement of digital technology, diesel engine has the development trend toward electronic control. Based on the available technology and CAN bus, DECS-Ⅰ diesel engine integrated electronic control system is developed. This ECS employs modular design, and achieves real-time communication of data with CAN bus, and thus is able to control the speed of engine and monitor the system. J*8K/46+I CAN bus; ECS; speed control; monitor
基于径向基神经网络的无位置传感器开关磁阻电机采煤机牵引系统
河北优控新能源科技有限公司EAXVA03技术手册说明书
EAXVA03 技术手册V1.0s河北优控新能源科技有限公司更改历史修改日期版本描述2019.08 V 1.0s 初始版本目录1.概述 (4)2.机械结构 (5)2.1机械尺寸 (5)2.2接插件 (5)2.3机械特性参数 (6)3.硬件描述 (7)3.1规格参数 (8)3.2产品外观 (8)4.开发工具 (10)4.1本地开发工具包 (10)4.2E CO SDK-XV (10)4.3E CO C ODER-ACU (10)4.4E CO C ODER (11)4.5E CO CAL (13)4.6E CO F LASH (13)1.概述EAXV A03产品是河北优控新能源科技有限公司针对自动驾驶系统开发的中央计算平台，内部主计算芯片为NVIDIA的Xavier芯片和英飞凌的TC297，整体支持ASIL-D最高安全等级要求。
开发者能够安全、便捷、高效的搭建出满足L4级别要求的自动驾驶系统。
Xavier芯片是NVIDIA专为包括自动驾驶系统在内的嵌入式智能系统而设计的。
Xavier拥有超过90亿个晶体管，每秒可执行30万亿次操作，但功率仅为30瓦，比上一代的TX2平台快了20倍。
Xavier有六种不同的处理器，这些处理器使其能够同时、且实时地处理数十种算法，以用于传感器处理、测距、定位和绘图、视觉和感知、路径规划。
其软件系统也是专门为自动驾驶系统而定制的，自下而上包含经过实时性优化的Linux操作系统，高性能运行时框架ROS，自动驾驶中间件Apollo Cyber RT等。
TC297包含300MHz工作频率的三核TriCore™架构，容量高达728KB + 2MB、带ECC（纠错编码）保护的RAM，基于ISO26262标准设计，支持ASIL-D最高安全等级要求。
其软件架构在设计时，严格按照AUTOSAR架构标准进行设计，分为应用软件层和基础软件层。
其中，基础软件层又分为微控制器抽象层、ECU抽象层、服务层和复杂驱动。
英飞凌单片机选型手册
XC166 40 MHz, Single Cycle
XC164CS Mid-range
XC164CM Low-end
XC167 High-end Motor Ctrl
XC161 High-end
C500 20 MHz, 12 Cycle
C505 44 Pin, CAN
C515 80 Pin, CAN
with 2 Nodes
–
MultiCAN with 2 Nodes
✓
✓
3 + 1-ch
✓
2
✓
✓
✓
PG-TQFP-64
✓
–
✓
✓
3 + 1-ch
✓
2
✓
✓
✓
PG-TQFP-64
✓
MultiCAN with 2 Nodes
✓
✓
3 + 1-ch
✓
2
✓
✓
✓
PG-TQFP-64
32-bit Microcontrollers
8
8-bit Microcontrollers
C505CA
10
C515C
12
C868
14
XC866
16
XC886/888CLM
18
XC886/888LM
20
Starter Kits for 8-bit Microcontrollers
22
16-bit Microcontrollers
C161K/O
24
✓
2
✓
✓
✓
PG-TQFP-48
✓
–
✓
✓
3 + 1-ch
英文的宣传广告词
英文的宣传广告词2. Never Stop Thinking! (探索未来，永无止境!) 英飞凌科技公司3. Feel the new space. 感受新境界。
(三星电子)4. The choice of a new generation. (Pepsi) 新一代的选择。
(百事可乐)5. We integrate, you communicate.(Mitsubishi) 我们集大成，您超越自我。
(三菱电工)6. Take TOSHIBA, take the world. 拥有东芝，拥有世界。
(东芝电子)7. Let's make things better. (Philips) 让我们做得更好。
(飞利浦电子)8. Fresh-up with seven-up. (Seven-up) 提神醒脑喝七喜。
(七喜)9. Connecting People. (Nokia) 沟通无处不在。
(诺基亚)10. A diamond lasts forever. (De Bierres) 钻石恒久远，一颗永流传。
(第比尔斯)11. Intel inside. (Intel Pentium) 给电脑一颗奔腾的“芯。
(英特尔)12. Focus on life. 瞄准生活。
(奥林巴斯相机)13. Good to the last drop. 滴滴香浓,意犹未尽。
(麦氏咖啡)14. A Kodak moment. 就在柯达一刻。
(柯达相纸/ 胶卷)15. Started Ahead. 成功之路,从头开始。
(飘柔洗发水)英文的宣传广告词欣赏1. Just do it. 跟着感觉走。
(耐克运动鞋)2. The taste is great. (Nestle coffee) 味道好极了。
(雀巢咖啡)3. Ask for more. (Pepsi) 渴望无限。
(百事流行鞋)4. Pepsi: Generation next 百事：新的一代5. Take time to indulge.尽情享受吧!(雀巢冰激凌)6. The taste is great. 味道好极了。
英飞凌IGBT及DIODE简介
Vbrces 650V @ 25°C Low Qg Low Coss / Eoss High Softness (H5) Co-pak diode robustness, low Qrr, high dI/dt, dV/dt
•Higher design margin •Cosmic radiation robustness in Solar •Low driving losses „cheap“ driver possible •High efficiency at light load •Low losses in resonant topology smaller resonant tank •low Rgon=Rgoff, Plug&Play replacement (H5) , easy of use. •Low EMI High Reliability in resonant topologies (ZVS/LLC) concerning hard commutation capability of anti-parallel diode
9/16/2013 Copyright © Infineon Technologies 2010. All rights reserved.
Turn-off controllability
Double Pulse characterization Turn-off
0.40
Vce=400V, Tc=25°C, Vge=15V, Ic=20A
25° C
Trade-off previous generation
150° C
TRENCHSTOP™5
TRENCHSTOP™5
At 150°C junction temperature the TRENCHSTOPTM5 offers: The same Vce(sat) value as the TRENCHSTOPTM family
英飞凌汽车电子器件选型
Lowbeam Indicator Park Optional Fog
55W
27W 10W 2x 55W
Park Indicator Lowbeam Highbeam
10W 27W 55W
65W
Highbeam 65W
Lowbeam 55W
Indicator 27W
Left Front-Light
Control
Right Front-Light
Control
LEDs
Relays
m n
Low-Side Driver
HITFET ™ BTS3110/18
BTS3134 BTS3160D
Interior Light
LED Driver
Basic LED Driver without Status
BCR40x
Basic LED Driver TLE424x
Power System ICs
C Smart Power C System Integration
– ABS/Airbag – Powertrain – Body
Infineon® Embedded Power ICs
C Single Package C Smart Power and
Controller Integration
System Basis Chip TLE826xE TLE826x-2E
Optional: DC/DC Regulator
TLF50281
Single or Dual High-Side Driver
Supply
Communication
32-bit Multicore/Lockstep
正确计算死区时间_英飞凌
AN2007-04 H o w t o c a l c u l a t e a n d m i n i m i z e t h e d e a dt i m e r e q u i r e m e n t f o r I G B T s p r o p e r l yPower Management and DrivesEdition 2008-05-07Published byInfineon Technologies AG81726 München, Germany© Infineon Technologies AG 2008.All Rights Reserved.Attention please!THE INFORMATION GIVEN IN THIS APPLICATION NOTE IS GIVEN AS A HINT FOR THE IMPLEMENTATION OF THE INFINEON TECHNOLOGIES COMPONENT ONLY AND SHALL NOT BE REGARDED AS ANY DESCRIPTION OR WARRANTY OF A CERTAIN FUNCTIONALITY, CONDITION OR QUALITY OF THE INFINEON TECHNOLOGIES COMPONENT. THE RECIPIENT OF THIS APPLICATION NOTE MUST VERIFY ANY FUNCTION DESCRIBED HEREIN IN THE REAL APPLICATION. INFINEON TECHNOLOGIES HEREBY DISCLAIMS ANY AND ALL WARRANTIES AND LIABILITIES OF ANY KIND (INCLUDING WITHOUT LIMITATION WARRANTIES OF NON-INFRINGEMENT OF INTELLECTUAL PROPERTY RIGHTS OF ANY THIRD PARTY) WITH RESPECT TO ANY AND ALL INFORMATION GIVEN IN THIS APPLICATION NOTE.InformationFor further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office ().WarningsDue to technical requirements components may contain dangerous substances. For information on the types in question please contact your nearest Infineon Technologies Office.Infineon Technologies Components may only be used in life-support devices or systems with the express written approval of Infineon Technologies, if a failure of such components can reasonably be expected to cause the failure of that life-support device or system, or to affect the safety or effectiveness of that device or system. Life support devices or systems are intended to be implanted in the human body, or to supportand/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the healthAP99007Revision History: 2007-08 V1.0 Previous Version: nonePage Subjects (major changes since last revision)FirstreleaseAuthor: Zhang Xi IFAG AIM PMD ID AETable of Contents Page 1Introduction (5)1.1Reason of IGBT bridge shoot through (5)1.2Impact of dead time on inverter operation (5)2Calculate proper dead time (6)2.1Basics for calculating the dead time (6)2.2Definition of switching and delay times (7)2.3Influence of gate resistor / driver output impedance (8)2.4Impact of other parameters on delay time (9)2.4.1Turn on delay time (9)2.4.2Turn off delay time (10)2.4.3Verification of calculated dead time (12)3How to reduce dead time (13)4Conclusion (14)Bibliography (15)1 IntroductionIn modern industry the voltage source inverter with IGBT devices is used more and more. To ensure proper operation, the bridge shoot through always should be avoided. Bridge shoot through will generate unwanted additional losses or even cause thermal runaway. As a result failure of IGBT devices and whole inverter is possible.1.1 Reason of IGBT bridge shoot throughThe typical configuration of a phase-leg with IGBTs is shown in the following figure. In normal operation two IGBTs will be switched on and off one after the other. Having both devices conducting at the same time will result in a rise of current only limited by DC-link stray inductance.Figure 1 Typical configuration of a voltage source inverterOf course no one will turn on the two IGBTs at the same time on purpose, but since the IGBT is not an ideal switch, turn on time and turn off time are not strictly identical. In order to avoid bridge shoot through it is always recommended to add a so called “interlock delay time” or more popular “dead time” into the control scheme. With this additional time one IGBT will be always turned off first and the other will be turned on after dead time is expired, hence bridge shoot through caused by the unsymmetrical turn on and turn off time of the IGBT devices can be avoided.1.2 Impact of dead time on inverter operationGenerally there are two types of dead time, the first one is control dead time and the second is effective dead time. The control dead time is the dead time, which will be implemented into control algorithms in order to get proper effective dead time at the devices. Target for setting control dead time is to ensure that effective dead time is always positive. Due to the fact that calculation of control dead time is always based on a worst case consideration, an effective dead time being a significant portion of the control dead time will result. Providing dead time can on one side avoid bridge shoot through but on the other side it has also adverse effect. To clarify the effect of dead time, let’s consider one leg of the voltage source inverter as shown in Figure. 2. Assuming first that output current flows in direction shown on the illustration IGBT T1 switches from ON to OFF and IGBT T2 switches from OFF to ON after slight dead time. During effective dead time both devices are off and freewheeling diode D2 is conducting output current. So negative DC link voltage is applied to the output, which is desired here. Consider the other case that T1 switches from OFF to ON and T2 from ON to OFF, then, with current in the same direction D2 still conducts the current during dead time, so that output voltage will be also negative DC link voltage, which is undesired here. The conclusion can besummarized as follows: during effective dead time output voltage is determined by the direction of output current but not the control signal.If we consider output current in the opposite direction than illustrated in figure 2, then we will gain a voltage if T1 switches from ON to OFF and T2 switched from OFF to ON. So in general output voltage and as a result also output current will be distorted with application of a dead time. If we choose a dead time unnecessary large, then in case of an induction motor the system will become instable and may cause some destructive effects [1]. So the process of choosing dead time is very important and should be performed very carefully.CurrentFigure 2One leg of voltage source inverterThis application note will explain how to measure delay time of IGBTs in practice and how to calculate the control dead time properly based on measurements.2 Calculate proper dead timeAs already mentioned, dead time should be chosen on one hand to satisfy the need of avoiding bridge shoot through, on the other hand dead time should be chosen as small as possible to ensure correct operation of voltage source inverter. So a big challenge here is to find out a proper dead time for a dedicated IGBT device and driver.2.1 Basics for calculating the dead timeFor calculation of control dead time we use the following equation:()[]2.1)(______×−+−=MIN PDD MAX PDD MIN ON D MAX OFF D dead t t t t t (1)Where Td_off_max : the maximal turn off delay time. Td_on_min : the minimal turn on delay time.Tpdd_max : the maximal propagation delay of driver. Tpdd_min : the minimum propagation delay of driver.1.2: safety margin to be multiplied.In this equation the first term td_off_max-td_on_min is the difference of the maximal turn off delay time andthe minimal turn on delay time. This term describes characteristic of IGBT device itself plus gate resistor which is used. Since fall and rise time is normally very short in comparison with delay time, they will be not considered here. The other term tpdd_max-tpdd_min is the propagation delay time difference (delay timemismatch) which is determined from driver. This parameter will be found normally in datasheet of driver from driver manufacturers. Typically this value is quiet high with opto-coupler based drivers.Sometimes dead time will be calculated from typical datasheet values just multiplying by a safety factor from field experience. This method will work in some cases but is not precise enough in general. With measurements shown here, a more precise approach will be presented.Because IGBT datasheet only gives typical values for standardized operation condition, it is necessary to obtain the maximal values for dedicated driving condition. For this purpose a series of measurements is done in order to obtain proper value for delay time and then to calculate dead time.switching and delay timesof2.2 DefinitionSince we will talk a lot about switching and delay times, it is necessary to give a clear definition here. Infineon Technologies defines the switching time of IGBT as follows:t d_on: from 10% of Vge to 10% if I c.t r: from 10% of Ic to 90% of I c.t d_off : from 90% of Vge to 90% of I c.t f: from 90% of Ic to 10% of I c.Figure 3 Definition of switching times.2.3 Influence of gate resistor / driver output impedanceThe choice of gate resistor will have significant impact on switching delay time. Generally to say, the larger the resistor is the longer the delay time will be. It is recommended to measure delay time with dedicated gate resistor in application. A typical switching time vs. gate resistor value diagram is shown in the following figures:Figure 4Switching times vs. Rg @25°CAll tests were done with FP40R12KT3 module, gate voltage is -15V/+15V, DC link voltage is 600V and switched current is nominal current of 40A.2.4 Impact of other parameters on delay timeBesides the gate resistor values, there are other parameters having significant impact on delay times: • Collector current.•Gate drive supply voltage.2.4.1 Turn on delay timeTo estimate this relationship, a series of measurements was done. First the dependence of turn on delay time and current was investigated. The results are shown in the next figure:Figure 6 The turn on delay time vs. switched current IcAll tests were done with a FP40R12KT3 module at a DC link voltage of 600V, gate resistor is chosen according to datasheet value.From results above it can be seen that turn on delay time is almost constant with variation of collector current Ic. With -15V/+15V gate voltage turn on delay time will get larger than with 0V/+15V gate voltage [2]. For further calculation of control dead time this variation will be neglected since it is quiet small and provides even additional margin.2.4.2 Turn off delay timeThe most important factor in the calculation of dead time is the maximal turn off delay time. Since this value determines almost entirely how long the final calculated dead time will be. So we will investigate this delay time in detail.In order to obtain the maximum turn off delay time following considerations have to be done:1. What and how long is the turn on delay time caused by IGBT device itself?To answer this question the following test based on a characterization driver board is done in laboratory. The characterization driver board is considered as an optimal driver, which means that this particular driver will cause no delay (which is almost true with an oversized driver), so the whole delay time is considered to be caused by the IGBT device itself. Following block diagram shows test setup:Figure 7 Block diagram of test with ideal driver2. What is the maximal turn off delay time if the threshold voltage of IGBT has the minimal value in datasheets? (this reflects the tolerance of Vth from module to module)To answer this question an additional diode is connected to simulate the reduced Vth voltage. The diode has a voltage drop of approximately 0.7…0.8V, which is quite similar to the Vth variation of FP40R12KT3 module. Following block diagram shows principle test setup:Figure 8 Block diagram of the test to simulate variation of Vth in worst case.3. What is the impact of driver output stage on switching times?To answer this concrete question the drivers on the market were splitted into two categories, one with mosfet transistor output stage and the other one with bipolar transistor output stage. For each category separate measurements were made.To simulate drivers with mosfet output stage, another additional resistor was connected and has been considered as the on state resistor Rds(on) of Mosfet transistor. The diode for simulation of Vth variation remained. The following block diagram shows the principle test setup:Figure 9 Block diagram of test to simulate variation of Vth and driver with mosfet output.4. What is the impact of the driver with bipolar transistor output stage?To answer the question an additional diode which simulated the voltage drop on bipolar transistor within output stage was connected. The following block diagram shows principle test setup:Figure 10 Block diagram of the test to simulate the variation of Vth and driver with bipolar transistor outputWith the configurations shown above the measurement of turn off delay time was done in our laboratory with module FP40R12KT3 and driver board which had been considered as optimal. Test conditions were Vdc=600V, Rg=27Ω. Results are shown in the next two figures:Figure 11 Turn off delay time vs. Ic @25°CFigure 12 Turn off delay time vs. Ic @125°CFrom the results we can see that there is a significant increase of turn off delay time with decrease of the switched current Ic. So just simply calculate dead time according to a chosen gate resistor is obviously not precise enough. Measuring the delay time under the dedicated driving condition then calculating dead time according to these values is a better and more precise way. Normally measurement until 1% of the nominal current would be enough to give a sufficient overview for calculating required dead time.Another point to be considered here is that the turn off delay time will increase with 0V/+15V gate drive, and the impact of output stage on switching times will be bigger with 0V/+15V switching. This means that with 0V/+15V switching voltage special care has to be taken by choosing the driver. Additionally, the increase of td_off with lower switched collector current Ic should be considered also.As an example: the HCPL-3120 driver IC will be considered here. This driver IC has a Mosfet output stage for switching off. From diagrams above we can read the value for td_off under 0V/+15V switching condition is roughly 1500ns. The td_on in this case is about 100ns. The tpdd_max-tpdd_min of this driver IC according to datasheet is 700ns. Applying these values to the formula (1) results in a dead time of about 2.5µs.2.4.3 Verification of calculated dead timeWith the discussion above and the formula (1) given in chapter 2.1 it is now possible to calculate the required dead time based on the measurements above. With the calculated dead time, a worst case measurement can then be performed to verify if the chosen dead time is enough or not.From the measurement it can be seen that the turn off delay time increases with temperature. From this reason it is preferable that the test should be done both at cold and hot condition.The schematic illustration of the test looks like following:Figure 13 Schematic illustration of test to check calculated dead time valueThe bottom IGBT has to be switched on and off, then the same for the top one. The time between the two pulses should be adjusted to be the value of calculated dead time for the dedicated driving condition. The negative dc-link current can then be measured and if the dead time is sufficient, a shoot through current should not be observed.Since there is no current through both IGBT, the described test represents the worst case condition for dead time calculation. From the discussion of turn off delay time it is known that dead time will be longer with decrease of collector current, so in case there flows no current, turn off delay time should be largest, which leads to a need of largest dead time. If there is no shoot through current at zero collector current then the chosen dead time is for dedicated driving condition sufficient.3 How to reduce dead timeFor a proper calculation of control dead time the dedicated driving condition should be considered: •What is the applied gate voltage to the IGBT?•What is the chosen gate resistor value?•What type of output stage does the driver have?Based on these conditions a test should be made, from the test results the control dead time can then be calculated using equition (1).Since dead time has a negative impact on the performance of inverter, it has to be minimized. Several methods can be taken.•Take a driver strong enough to sink or source the peak IGBT gate current.•Use negative power supply to accelerate turn off.•Prefer drivers based on fast signal transmission technology like Coreless Transformer Technology to drivers based on traditional opto-coupler technology.•If 0V/15V gate drive is used then consider use of separate Rgon/Rgoff resistor as described below.From measurements shown in chapter 2.3 a very strong dependence of Td_off and gate resistor value can be observed. If the Rgoff reduced then the td_off will be reduced as well as dead time. Infineon suggests reducing the Rgoff to 1/3 of the Rgon value if 0V/15V gate voltage is used. One possible circuit for separate Rgon and Rgoff is as follows:Figure 14 Suggested circuit with 0V/15V gate voltage.R1 should be chosen to satisfy the following relation:)(31int int 11g gon g gongon R R R R R R R +⋅=++⋅(2)int int 1221g gon g gon gon R R R R R R +−⋅⋅==>(3)From equation (3) it is to be noticed that the requirement Rgon>2Rgint has to be fulfilled to get a positivevalue of R1. However, with some modules this requirement can not be true. In this case, R1 can be omitted completely.The diode should be a schottky type diode.Another very important issue with 0V/15V gate voltage is the parasitic turn on effect. This issue can be also solved if suggested circuit is used. For more details on parasitic turn on please refer to AN2006-01[2].4 ConclusionIn this application note an approach of measuring switching times of IGBT and then calculating the control dead time is introduced. First dependence of switching time on gate resistor value was shown, and then influence of gate driver and collector current on switching times was discussed. Finally possible methods to reduce dead time were introduced.Bibliography[1] D.Grahame Holmes, Thomas A. Lipo: …Pulse width modulation for power converters: principles andpractice“, IEEE Press, 2003. ISBN 0-471-20814-0[2] Driving IGBTs with unipolar gate voltage./dgdl/an-2006-01_Driving_IGBTs_with_unipolar_gate_voltage.pdf?folderId=db3a304412b407950112b408e8c9000 4&fileId=db3a304412b407950112b40ed1711291。
全球知名品牌的英文广告标语_广告词
全球知名品牌的英文广告标语服装品牌一定需要广告语，广告语应该是体现品牌价值的灵魂。
那么有什么全球知名的英文广告语呢？下面是橙子为大家精心挑选的全球知名品牌的英文广告标语，希望对大家有所帮助。
全球知名品牌的英文广告标语(最新) 1. Impossible made possible. 使不可能变为可能。
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(ibm公司)4. Just do it. 跟着感觉走。
(耐克运动鞋)5. The taste is great. (Nestle coffee) 味道好极了。
(雀巢咖啡)6. Ask for more. (Pepsi) 渴望无限。
(百事流行鞋)7. Pepsi: Generation next 百事：新的一代8. Take time to indulge.尽情享受吧!(雀巢冰激凌)9. The taste is great. 味道好极了。
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11. Ask for more. 渴望无限。
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(雪碧)1 / 813. Olympus: focus on life 奥林巴斯：瞄准生活。
14. We lead，Others copy.我们领先，他人仿效。
(理光复印机)15. Impossible made possible.使不可能变为可能。
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如何正确读懂英飞凌的IGBT的资料
Switching parameters
! Gate charge (QG)
This value is specified at +/-15V, used to calculate driving power
! Cies, Cres
Cies = CGE + CGC: Input capacitance (output shorted) Coss = CGC + CEC: Output capacitance (input shorted) Cres = CGC: Reverse transfer capacitance (Miller capacitance) Required gate power at switching frequency f:
Infineon datasheet understanding
IFX AIM Zhou Yizheng
Infineon datasheet understanding
Current parameters Current parameters Voltage parameters Voltage parameters Switching parameters Switching parameters Diode parameters Diode parameters Thermal parameters Thermal parameters Module parameters Module parameters
∆VCE VCE ( 2) − VCE (1) = = I C ( 2) − I C (1) ∆I C
Basic data for conduction losses calculation
汽车电子自动变速箱-ADC测量和规范应用攻略
AudoNG, AudoFuture and AudoMax family静态误差参数1 引言这个应用笔记描述的是英飞凌微控制器中模数转换使用的测量方法。
它包含ADC的静态和动态误差参数的定义，以及用于描述ADC的测试建立和算法。
也简述了特征化/验证英飞凌的微控制器的ADC 时用的测试条件。
本文目的是帮助微控制器的ADC用户更好地理解其规范以及更有效地使用ADC。
2 术语和定义整篇文章中的小写字母代表了一个数据矩阵，而大写字母却用作信号值。
例如，tue表示整个数据矩阵，而 TUEmin 和 TUEmax 是用来说明最小和最大值。
3 静态误差参数3.1 理想ADC 转换曲线有两种理想ADC转换曲线。
对于一个实时ADC的误差参数的正确计算来讲，知道其相应理想转换曲线是很重要的。
图1表示3位AD转换的未补偿理想ADC转换曲线。
对所有代码其转换宽度是相同的。
量化误差的结果是从 0 到-1LSB。
图 1 3位AD转换器的未补偿理想转换曲线和量化误差另一个定义方法是，第一个代码宽度仅有½LSB，而最后一个代码宽度则是1½ LSB。
这种类型通常是如图2所示的½LSB补偿转换曲线，其量化误差是 ±½ LSB。
在英飞凌微控制器中，标准单端ADC采用½ LSB补偿转换曲线，而快速差分ADC采用1LSB补偿转换曲线。
注释:对于未补偿转换曲线，模拟输入范围的中点值经转换后位于代码2n/2-1到2n/2之间，而对于补偿转换曲线，其中间值经转换后则是位于代码2n/2处。
这两类转换曲线都有2n个代码值，但都只有2n-1代码跳变。
图 2 3位AD转换器的补偿理想转换曲线和量化3.2 总未调整误差实际ADC的转换曲线会偏离理想的ADC转换曲线。
实测转换曲线和理想转换曲线的偏差定义为未调整总误差（TUE）。
图3显示了3位ADC的例子，由于tue的计算是基于代码跳变的，因此在tue图上共有2n-1个tue值:tue[i]=real_code_transition[i]-ideal_code_tanstition[i]i=0→2n-1在转换过程中tue 的值通常以LSB来表示。
英飞凌16位单片机2287-DAVE演示文档-CAN
CAN_2
Init Main • fsys=80MHz • Init USIC
Init MultiCAN • Module • Node 0 • Node 1 • Message Objects • Start nodes
Page 15
HOT Exercise CAN_1 - DAvE Configurations A MultiCAN settings
Configure CAN Node 1 Baud Rate: ¬ Required baud rate : 500 Kbaud ¬ TSeg2 : 5 to get Real baud rate at 500 Kbaud
Configure the XC2000 with DAvE Configure USIC 0 Channel 0 as a UART Receive a character from a PC and generate a receive interrupt Transmit the character on CAN node 0 and receive the CAN message on CAN node 1 Transmit the value back to the PC and toggle one of the LEDs on the board on receipt of every character
Configure CAN Node 1 General:
¬ Select P2.4 for Receive Input and P2.2 for Transmit Output
嵌入式软件PIL自动化测试技术研究
10.16638/ki.1671-7988.2021.012.019嵌入式软件PIL自动化测试技术研究郭佳，金鑫，邓煜（陕西重型汽车有限公司，陕西西安710200）摘要：文章旨在提出一种PIL自动化测试方法，主要分析了当前汽车控制器开发中的主流测试流程，介绍了PIL 测试的必要性和测试原理，然后介绍了一种借助自动化软件测试工具TPT进行PIL测试的方法，通过一个实例详细介绍了在TPT中实现PIL自动化测试的过程。
通过分析可以看出通过自动化工具TPT进行PIL测试的方案可行性以及该方案对提升PIL测试效率的优势，为嵌入式软件开发的PIL自动化测试提供了一种新的测试方法。
关键词：TPT；PIL测试；汽车电子；V流程中图分类号：U461.99 文献标识码：A 文章编号：1671-7988(2021)12-64-05Research on Automatic Test Technology of Embedded Software PILGUO Jia, JIN Xin, DENG Yu( Shaanxi Heavy Duty Automobile Co., Ltd., Shaanxi Xi’an 710200 )Abstract: The purpose of this paper is to put forward a PIL testing methods, mainly analyzes the mainstream of the current car controller development testing process, introduces the necessity of PIL test, PIL test principle, then introduces a kind of automated software testing as a tool to TPT tested PIL, TPT is introduced by an example of realization process of PIL, through the analysis can be seen through an automated tool TPT PIL test feasibility and advantages of the scheme to improve PIL test efficiency, for car controller development of PIL test provides a new method of testing.Keywords: TPT; PIL test; Automotive electronics; VprocessCLC NO.: U461.99 Document Code: A Article ID: 1671-7988(2021)12-64-05引言当前汽车控制器的开发大都采用V流程开发模式。
曲轴、凸轮轴同步信号以及变转速信号模拟
}
示如果要改变转速，可以把上面程序添加一些程序，只赋给变量不同的转速， 其他参数单片机自动计算。 伪代码： Speed; //发动机转速单位 r/min
real_Littl_Period=1000000/Speed;//曲轴一个小波周期,单位us Bei_Littl_Period=10*real_Littl_Period;//为最小时钟间隔0.1us的倍数 GPTA0_LTCXR01 GPTA0_LTCXR02 GPTA0_LTCXR03 GPTA0_LTCXR04 GPTA0_LTCXR05 GPTA0_LTCXR06 ； 分别给变量不同的值（不同的转速） ，查看产生的曲轴和凸轮轴信号。 下面分别把转速设为 800r/min、1000r/min、2000r/min、3200r/min 情况 下，观察示波器波形，为了方便进行不同转速的模拟信号对比，示波器测量时， = = = = = = Bei_Littl_Period； Bei_Littl_Period /2； 3* Bei_Littl_Period； 2.5* Bei_Littl_Period； Bei_Littl_Period； Bei_Littl_Period /2
DAVE 配置： 时钟选择如下图
要用到 LTC0—LTC6.设置如下:
初 始 值 LTC0 0000 时钟 7 输入 输入敏 感 边沿敏 感（高 水平） 初 始 值 LTC1 LTC2 0x1388 0x09C4 无 无 输入 输入敏 感 无 无 SI 线使 能状态 低 低 reset set or copy LTC3 0x3A98 无 无 高 reset or copy LTC4 0x30D4 无 无 高 set or copy P0.0(连接 示波器的 通道 1) LTC5 LTC6 0x1388 0x09C4 无 无 无 无 总是 总是 Hold Hold 无 无 比较 比较 比较 无 比较 无 无 比较 比较 事件动作 输出 模式 中断 默认 SO 线 低 Hold 无 复位定时 有 事件动作 输出 模式 中断
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英飞凌第四代REAL3™飞行时间图像传感器将亮相
MWC2019
2019年2月28日，德国慕尼黑讯——英飞凌科技股份公司（FSE：IFX / OTCQX：IFNNY）在西班牙巴塞罗那2019年世界移动通信大会上展示第四代REAL3™图像传感器IRS2771C。
该3D飞行时间（ToF）单芯片器件旨在满足移动消费终端市场的需求，特别是满足利用小镜头支持更高分辨率的需求。
广泛的系列应用包括安全的用户身份验证（如人脸识别或手部识别等），用以解锁设备和确认付款。
此外，这款3D ToF芯片有助于提升增强现实技术、图像变形技术和照片（如bokeh）特效，并可用来扫描房间。
这款图像传感器的尺寸仅为4.6 x 5 mm，150 k（448 x 336）像素输出接近HVGA标准分辨率。
这使其分辨率比现今市场上大多数ToF解决方案的分辨率高四倍。
其像素阵列对940纳米红外光高度敏感，能带来无与伦比的性能。
这是通过每个像素处获得专利的“背景照明抑制”（SBI）电路实现的。
由于其集成度很高，每个IRS2771C图像传感器差不多是一个微型单片ToF摄像头。
这大大缩小了物料清单成本及摄像头模块实际尺寸，同时不影响性能，并能最大限度降低功耗。
市场领先的耐用性及节能优势。