APD偏压电路的最佳设计 - 外文翻译

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平板变压器设计详解

平板变压器设计详解

Design of PlanarPo wer T ransfo rmersContentsIntroduction 3Design procedure4Design examples-flyback8-forward 10Formulas13Layer design141FerroxcubeExploded view of a planar transformer2Ferroxcube3 FerroxcubeT able 1: Fit parameters to calculate the power loss density4Ferroxcube3. Determination of temperature rise in the PCB caused by the currentsThe final step is to check the temperature rise in the copper tracks induced by the currents. For this purpose the effective (= RMS) currents have to be calculated from the input data and desired output. The calculation method depends on the topology used. In the design examples this is shown for a conventional standard forward and flyback converter topology. An example of relations between the RMS currents and induced temperature rises for various cross sections of conductors in PCBs is shown in fig. 2. For single conductor applications or inductors which are not too closely spaced this chart can be used directly for determining conductor widths, conductor thickness, cross sectional areas and allowed maximum currents for various preset values of the temperature rise.Note:For groups of similar parallel inductors, if closely spaced, the temperature rise may be found by using an equivalent cross section and equivalent current. The equivalent cross section is the sum of the cross sectionsof the parallel conductors and the equivalent current is the sum of the currents in the inductor.A shortcoming in this design approach is that the induced heat in the windings is assumed to be caused by a DC current while in reality there is an AC current causing skin effect and proximity effect.The skin effect is the result of the magnetic field inside a conductor generated by the conductors own current. Fast current changes (high frequency) induce alternating fluxes which cause eddy currents. These eddy currents which add to the main current are opposite to the direction of the main current. The current is cancelled out in the centre of the conductor and moves towards the surface. The current density decreases exponentially from the surface towards the centre.The skin depth δ is the distance from the conductor surface towards the centre over which the current density has reduced by a factor of 1/e. The skin depth depends on material properties as conductivity and permeability and is inversely proportional to the square root of the frequency. For copper at 60 °C the skin depth can be approximated by: δ(µm) = 2230/(f [kHz])1/2 .When the conductor width (w t ) is taken smaller than 2δ, the contribution of this effect will be limited.This means a track width of <200 µm for a frequency of 500 kHz.If there is more winding width b w available for the concerned number of turns, the best solution from the magnetic point of view would be to split them up in parallel tracks.In practical situations there will be eddy current effects in the conductor not only due to the alternating field of its own current (skin effect) but also due to the fields of other conductors in the vicinity. This effect is calledthe proximity effect. When the primary and secondary layers are sandwiched this effect will be strongly decreased. Reason is that the primary and secondary currents flowin opposite directions so that their magnetic fields will cancel out. However there will still be a contribution to the proximity effect of the neighbouring conductors in the same layer.Empirical toolTemperature measurements on several designs of multilayer PCBs with AC currents supplied to the windings, show with reasonable accuracy that up to 1MHz each increaseof 100 kHz in frequency gives 2 °C extra in temperature rise of the PCB compared to the values determined for DC currents.Ferroxcube6Fig.2 Relation between current, dimensions of tracks in PCBs and temperature rise.7Ferroxcube8 FerroxcubeDepending on the heat generated by the currents the Array choice can be made between 35 or 70 µm copper layers. Between primary and secondary layers a distance of 400µm is required for the mains insulation. An E-PLT 18 combination has a minimum winding window of 1.8.mm. This is sufficient for the 35 µm layer design which results ina PCB thickness of about 1710 µm.To achieve a economic design we assumed a spacingof 300 µm between the tracks. Calculating the track widthfor the secondary winding with [5] returns 1.06 mm, inclusive mains insulation.Looking in fig 2. and using the calculated (see table 2) secondary RMS current of 1.6 A, results in a temperaturerise of 25 °C for the 35 µm layers and approx. 7 °C forthe 70 µm design.The temperature rise caused by the winding loss is allowedto be about half the total temperature rise, in this case17.5 °C. Clearly the 35 µm layers will give a too large temperature rise for an RMS current of 1.6 A and the70 µm layers will have to be used.The track widths for the primary winding turns can be calculated with [5] and will be approx. 416 µm. This track width will cause hardly any temperature rise by the primary RMS current of 0.24 A.Because the frequency is 120 kHz, 2 °C extra temperaturerise of the PCB is expected compared to the DC current situation. The total temperature rise of the PCB caused bythe currents only will remain below 10 °C.This design with 6 layers of 70 µm Cu tracks shouldfunction within its specification. The nominal thicknessof the PCB will be about 1920 µm which means that a standard planar E-PLT18 combination cannot be used.The standard E-E18 combination with a winding windowof 3.6 mm is usable. However its winding window is excessive, so a customized core shape with a windingwindow of approximately 2 mm would be a more elegant solution.Measurements on a comparable design with an E-E core combination in 3C90 material showed a total temperaturerise of 28 °C. This is in line with a calculated contributionof 17.5 °C temperature rise from the core losses and 10 °C caused by winding losses.The coupling between primary and secondary is good because the leakage inductance turns out to be only0.6 % of the primary inductance.9FerroxcubeReferences1. Mulder S.A., 1990Application note on the design of low profile high frequency transformers, Ferroxcube Components.2. Mulder S.A.,1994Loss formulas for power ferrites and their use in transformer design,Philips Components.3. Durbaum Th, Albach M, 1995Core losses in transformers with an arbitrary shape of the magnetizing current.1995 EPE Sevilla.4. Brockmeyer A., 1995Experimental Evaluation of the influence of DC premagnetization on the properties of power electronic ferrites, Aachen University of T echnology5. Ferroxcube Components technical note, 199625 Watt DC/DC converter using integrated Planar Magnetics.(9398 236 26011)Appendix 1: Formulas used for the calculations of the designformulas for flyback transformers formulas for forward transformersAppendix 2: Layer design for the planar E 14 forward transformerTop view of the example multilayer PCB。

光纤通信系统实验指导

光纤通信系统实验指导

ZY1804I光纤通信原理实验系统简介本实验系统是为配合《光纤通信》课程的理论教学,结合目前光纤通信工程技术最新进展,为了提高大专院校学生实际操作和动手能力而研制开发的。

一、产品的系统特点光纤I型实验系统注重产品的系统和功能组成,产品的设计着重体现系统性、先进性、实用性,并根据市场及客户实际需求,充分考虑工艺外观结构、产品的功能和性价比。

整个系统分中央控制器、备用环和光传输三大部分,各自独立又相互关联,所有模块在单独进行实验同时又可系统集联,实验灵活丰富,可设计、可比较、可操作、可观测性强。

整个系统采用2.048M传输速率,既有利于实验观测,又可以模拟实际光纤传输时的各种性能。

实验紧密结合光通信新技术的发展趋势,将波分复用、光时分复用和SDH传输网等新技术都通过实验演示出来,简单易懂。

采用大规模的现场可编程门阵列器件,使得产品的开放性、可升级性强。

同时为了实现自愈环(即备用环)功能以及使学生有更大的开发和操作空间,特意制作了二次开发板,并预留大量的I/O扩展口,可在开发板上独立完成二次开发设计。

所有实验大多采用开关控制,减小了实验操作时的繁琐性。

该实验系统融合了当今的光纤通信技术发展的一些新技术和新器件,并将其融入到光纤通信原理课程当中,同时与通信原理和程控交换课程的部分原理结合,其主要有以下特点:1、实验箱采用“整板+核心板”设计,特殊光器件玻璃罩保护,元器件贴片化,模块元件布局完全对称。

所有的测试钩和连接孔均有标识,深蓝色的电路板,白色丝印使得整个电路板层次性强、美观、大方。

2、实验箱和光纤通信原理教材紧密结合,实验项目和顺序与教材保持完全同步。

通过八个方面全面实验来了解光纤通信的全过程,八个方面分别是:光纤和光缆;通信用光器件(有源器件和无源器件);光端机(光发、光收端机);数字光纤通信系统;模拟光纤通信系统;光纤通信新技术;光纤通信测量技术;光纤通信网络。

3、系统采用整板上分模块的设计方式,除了核心板——中央控制器外,还配置了光发端机、光收端机、模拟信号源、数字信号源、数字终端、电话模块、串口通信模块等。

光电二极管偏置电路设计

光电二极管偏置电路设计

光电二极管偏置电路设计英文回答:Photodiode Bias Circuit Design.Introduction.A photodiode is a semiconductor device that converts light into electrical current. It is a type of photodetector that is used in various applications, such as optical communication, position sensing, and light measurement. The performance of a photodiode is greatly influenced by its bias circuit.Bias Circuit.The bias circuit of a photodiode is a circuit that provides the necessary voltage and current conditions for the operation of the photodiode. The main purpose of the bias circuit is to establish a reverse bias across thephotodiode, which allows the photodiode to generate a photocurrent when illuminated.There are two main types of bias circuits:Constant Voltage Bias Circuit: This circuit maintains a constant voltage across the photodiode, regardless of the light intensity. It is typically used in applications where a stable photocurrent is required.Constant Current Bias Circuit: This circuit maintains a constant current through the photodiode, regardless of the light intensity. It is typically used in applications where a wide dynamic range is required.Design Considerations.The design of a bias circuit for a photodiode involves several important considerations:Reverse Bias Voltage: The reverse bias voltage applied to the photodiode should be large enough to ensure that thephotodiode is fully depleted and operates in the reverse-biased region.Load Resistance: The load resistance connected to the photodiode determines the output current and voltage of the photodiode. A higher load resistance will result in ahigher output voltage and a lower output current.Dark Current: The dark current is a small current that flows through the photodiode even when it is not illuminated. The bias circuit should be designed tominimize the dark current.Responsivity: The responsivity of a photodiode is the ratio of the output current to the incident light intensity. The bias circuit should be designed to maximize the responsivity of the photodiode.Design Procedure.The design procedure for a bias circuit for aphotodiode typically involves the following steps:1. Determine the desired operating voltage and current for the photodiode.2. Select the type of bias circuit (constant voltage or constant current).3. Calculate the required reverse bias voltage and load resistance.4. Choose appropriate components for the bias circuit.5. Test and verify the performance of the bias circuit.Conclusion.The bias circuit is an essential part of a photodiode system. By carefully designing the bias circuit, it is possible to optimize the performance of the photodiode and meet the specific requirements of the application.中文回答:光电二极管偏置电路设计。

InGaAs_InPAPD探测器光电特性检测

InGaAs_InPAPD探测器光电特性检测
令bj=bj0+∆bj,bj0为给定初始值,利用Taylor展开,
将偏微分方程转化为可求解的线性代数方程组。
2.2 获得倍增因子M=1的 IP0 的方法 倍增因子定义为在完全相同的注入条件下,有
雪崩增益时通过器件的电流与无雪崩增益时通过器 件的电流之比。
在实际器件中,获得的最高直流倍增因子受串
联电阻的空间电荷效应限制,这些因素可以合并成
wi为权重,达到最小。使用Q→min为标准的拟合称 为最小二乘法。使用最小二乘原理处理非线性曲线
拟合,令权重wi=1,非线性曲线拟合的数学表达为: 已知一组数据{xi,yi},i=1,2,…, n,满足已知方程形 式f(xi, bj),j=1,2,…,m,求解{bj},使其满足
n
∑ Q = [ yi − f (xi ,bj )]2 → min ,即求解 ∂Q / ∂bj = 0 。 i =1
=
I0 P0
+ ∆IP0

3 测量
本文研究了台面型InP/InGaAs APD静态光电特
性。该APD的光敏面直径为500 µm,光照下的电流
与电压关系曲线及无光照下的暗电流与电压关系曲
线如图1所示,将有光照与无光照时候相同偏压下的
电流值相减得到的电流即为光电流。图1还显示了倍
增因子与偏压的关系,其中,实线对应由实验测量
中图分类号 TN312+.7
文献标识码 A
Measurement of the Static Optoelectronic Characteristics of InGaAs/InP Avalanche Photodiode
XIAO Xue-fang1, YANG Guo-hua1, GUI qiang1, WANG Guo-hong1, MA Xiao-yu1, CHEN Chao2, and CHEN Liang-hui1

APD偏压电路的最佳设计

APD偏压电路的最佳设计
由于实际使用中M是远远大于1的,暗电流可表示为 ≈ ,而APD过量噪声因子 =kM+(1-k)(2-1/M)≈2+kM。在这一近似条件下,当d( )/dM=0, 达到其最大值且倍增因子达到最佳,可表示为:
(2)
方程2中符号和方程1中符号的含义相同。我们知道,最佳倍增因子是外部温度、光信号功率、背景光功率,APD噪声、光谱灵敏度、放大器噪声和系统带宽的函数。此外,特别是APD内部结构决定了其倍增增益M随工作温度变化而变化。用温度系数 来描述这种影响。对于C30737系列的APD, 为0.6 V/℃,这意味着在相同条件下,当APD的工作温度增加1℃,为了维持APD倍增因子不变偏压需要增加0.6 V。
(9)
如果方程8中工作电压等于方程9的最佳值,温度补偿后APD获得最佳倍增因子,检测电路信噪比也能达到最高水平。
为了检查上述方法的有效性,我们在汽车半导体激光测距仪设计中运用此方案,并设定具体参数来检测150米远的目标。如图3所示,APD偏压 、背景光功率和串行电阻 之间的关系可通过对方程8和9进行数值计算和分析得到。通过同样的方式,如图4,检测电路当前的 、背景光功率和串行电阻 之间的关系可以通过方程1、8和9得到,这里 =30 nW, =0.35,B=35 M, =0~500nW。在背景光较弱时, =200nW, =8×10-11A,k=0.02, =400Ω, =1, =158.6V, =0.95VB.
APD偏压电路的最佳设计
孙纯生,秦世桥,王兴书,朱冬华
1.国防科技大学光电科学与技术学院,中国长沙410073
2.海军工程大学装备工程部,中国武汉430033
本文提出了一种基于温度补偿和负载电阻补偿的雪崩光电二极管(APD)反向偏压控制方法,并详细的分析了电二极管偏置电路的设计建立了一种理想的温度补偿和负载电阻补偿模型。据预测,这种控制方法特别适用于车辆使用的激光测距仪。实验结果证实,本文提出的设计可以很大程度的改善测距仪的性能。

apd 跨阻放大电路设计

apd 跨阻放大电路设计

apd 跨阻放大电路设计
APD(Avalanche Photodiode)跨阻放大电路是一种用于放大
光电二极管输出信号的电路,常用于光通信、光电检测等领域。

以下是一个常见的APD跨阻放大电路设计:
1. 选择适当的跨阻(Rbias)和电源电压(Vbias):根据APD
的规格参数和工作条件,选择适当的跨阻和电源电压。

一般来说,跨阻的大小决定了放大倍数,电源电压需要提供足够的偏置电压以确保APD正常工作。

2. 将APD连接到跨阻放大电路:将APD的阴极连接到跨阻的一端,将APD的阳极连接到电源的正极。

确保连接的稳固和
正确。

3. 将跨阻与负载电阻(Rload)连接:将跨阻的另一端连接到
负载电阻。

负载电阻的大小决定了电路的输出电流和电压。

4. 添加补偿电阻(Rc):为了补偿APD的零漂和温度变化引
起的电流变化,可以在跨阻和负载电阻之间添加补偿电阻。

补偿电阻的大小根据APD的特性曲线和设计需求确定。

5. 连接输出信号:将负载电阻连接到输出信号的接收器或放大器。

输出信号可以通过电流或电压进行传输和测量。

6. 添加滤波器和稳压器:根据设计需求,可以在电路中添加滤波器和稳压器来提高输出信号的质量和稳定性。

7. 进行测试和调整:在完成电路连接后,通过适当的测试和调整来确定电路的工作状态和性能。

可以采用示波器、光功率计等设备进行测试和测量。

以上是一个基本的APD跨阻放大电路设计步骤,具体的设计需要根据实际应用和需求进行调整和优化。

基于APD激光窄脉冲探测系统的研究

基于APD激光窄脉冲探测系统的研究

基于APD激光窄脉冲探测系统的研究崔一惟;贺伟【摘要】The laser narrow pulse detection system is mainly composed of transmitting and receiving parts. Since low⁃power laser narrow pulse signal is weak,it is urgent to improve the optical gain and the detection efficiency. The traditional method is to select a sensitive photoelectric detection device to solve the problem by amplifying circuit. The avalanche diode with internal gain is adopted in this paper as a photosensitive element. On the basis of this,the optical system is added to conduct amplifica⁃tion processing before detection,which can improve the efficiency of detection effectively.%激光窄脉冲探测系统主要是由发射和接收两部分组成,小功率的激光窄脉冲信号比较微弱,所以提高光增益增加探测效率是亟待解决的。

传统的解决手段是选择灵敏的光电探测器件,后续经过放大电路进行解决。

这里采用具有内部增益的雪崩二极管作为光敏元件,在此基础上增加光学系统,使探测前就进行放大处理,进而有效地提高探测效率。

【期刊名称】《现代电子技术》【年(卷),期】2015(000)004【总页数】4页(P135-138)【关键词】激光探测;APD;温控系统;光学系统【作者】崔一惟;贺伟【作者单位】西安邮电大学,陕西西安 710061;西安邮电大学,陕西西安710061【正文语种】中文【中图分类】TN312+.7-340 引言随着1960年第一个红宝石激光器的诞生,相应的激光探测技术也越来越受人们的关注与研究,激光测距、激光雷达、激光制导等国防级应用都离不开窄脉冲激光探测技术[1]。

基于ADL5317的APD偏压控制光功率监测电路的设计

基于ADL5317的APD偏压控制光功率监测电路的设计

1 引言目前,雪崩光电二极管(APD)作为一种高灵敏、能精确接收数据和测量光功率的光探测器件广泛应用于光纤传感、光纤通信网络中。

它借助于内部强电场作用产生雪崩倍增效应,具有极高的内部增益(可达102~104量级)。

然而,APD随温漂的变化严重影响其增益的稳定性.甚至引起测量精度的恶化。

理论上可以证明APD的增益是其偏压V和温度T的函数,二者共同决定APD工作时的增益,而且在维持APD增益比较恒定的条件下,其偏压和温度之间存在一定的关系。

因此。

可以控制APD的偏压使之随温度按一定的规律改变。

这样就可以维持APD增益基本恒定,保证其正常工作。

这就是对APD温度漂移的偏压补偿原理。

由此可知.施加在APD上的偏置电压必须能够精确受控是保证光纤系统性能的首要要求。

本文针对该要求。

采用ADL5317器件。

给出了一种具有高精度、宽动态范围的APD 偏压控制/光功率监测功能的核心电路。

2 引脚排列及功能ADL5317是ADI公司率先在业界推出的一款片上集成雪崩光电二极管(APD)偏置电压控制和光电流监测功能的器件。

ADL5317的主要特性如下:通过3V线性偏置控制电路,在6 V~75 V范围内精确设置雪崩二极管(APD)偏置电压;在106范围(5 nA一5 mA)内以5:1的比率监测光电流,其线性误差仅为0.5%;允许使用固定的高电压转换电路,降低传统APD偏置设计中对电源解耦和低通滤波的要求;过流保护和过热保护。

ADL5317采用3 mm*3 mm的16引脚LFCSP封装,其引脚排列如图1所示。

各引脚功能描述如表1所列。

3 内部结构及工作原理ADL5317的内部结构如图2所示。

其内部包括电流监测电路、偏置控制电路、GARD 电路、VCLH电路、过流和过热保护电路。

3.1 电流监测电路ADL5317的核心部分是一个具有电压跟随性质的精密电流衰减电路,为监测电路输入端提供精确偏置。

该电路采用了结型场效应管输入形式的放大器.驱动监测电路的两极,同时保持V APD端电压的稳定度及非常低的漏电流。

一种高性能APD反向偏压控制电路的实现

一种高性能APD反向偏压控制电路的实现

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低噪声APD偏置电路

低噪声APD偏置电路

-74-《国外电子元器件》2002年第9期2002年9月●MAXI M专栏低噪声APD偏置电路M AXIM公司M ehm et Nalbant徐继红1APD的特性雪崩光电检测器(APD)和PIN二极管通常被作为接收器用于光通信中。

其中APD适合于高灵敏度和高带宽的光接收电路。

但这种器件在工作时需要施加一个反向结压,这样,当接收到射线时产生的电子空穴对会被外加电场收集并转换为电流,其电流强度正比于射线强度。

另外,工作时施加在器件上的反向偏置电压会引发雪崩效应,其雪崩增益可通过改变偏压来进行调节。

这就有可能对光纤接收器的增益进行优化。

然而,要得到满意的雪崩增益,就必须给APD提供一个比较高的反向偏压。

很多APD 需要40V~60V的偏压,有些器件甚至要求高达80V 的反向电压。

另外,该增益还会随着温度的变化而改变,而且还要受到制造工艺的影响。

因此,在一个典型系统中,如果要求APD工作于恒定增益,其高压偏置电源必须能够改变,以补偿因温度和制造工艺而造成的增益变化。

同时要获得恒定的增益,APD 电源必须具有大约+0.2%/℃的温度系数(相当于约100mV/℃)。

图1给出了一个典型的a g ere1319型接收器所要求的偏置电压随温度的变化关系。

2APD电源有很多方法可使APD电源具有可变的输出电压,以补偿增益随温度的变化。

实际上APD模块内的温度测量元件本身就可以直接接入电源来调节输出电压。

在有些系统中,也可以由微控制器来读取电阻值,然后向电源发出指令以调节偏置电压。

图2所示是一个APD偏置电源的基本原理。

这个电路是基于M AX5026低噪声、固定频率PWM升压转换器而设计的,可工作于电感电流不连续模式。

该器件的开关速度被有意减慢的目的是便于降低高频电压毛刺。

同时开关速度的降低还可减小高频di/dt和dv/dt辐射。

作为辐射及耦合噪声的主要来源,它们会通过电流环以及管脚到管脚、管脚到PC B线条之间的寄生电容进入周边电路。

雪崩光电二极管APD直流偏压源设计

雪崩光电二极管APD直流偏压源设计

1 引言
雪崩光 电二极管 A D( vl ce P o i e 是一 种能实 P A a n h ht Do ) a o d 现光电转 换且具有 内部增益 的高灵敏度器件 。在 以硅或锗为材 料制成的光 电二极管的 P N结上 加上反 向偏压后 ,射人 的光被 —
P N结吸收后会形成光 电流 。加大反 向偏压会产生“ — 雪崩” 即光 ( 电流成倍地激 增 ) 的现象 , 因此 这种二极管被称 为“ 崩光 电二 雪 极管” 。目前 , 雪崩光电二极管( P 作为一种高灵敏 、 A D) 能精确接
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APD偏压的自适应电路设计

APD偏压的自适应电路设计

电子技术• Electronic TechnologyAPD 偏压的自适应电路设计文/朱斌本文介绍了雪崩光电二极管摘 偏压、增益、温度三者之间的关系。

要 设计出基于PIC 单片机的自适应 ■调节电路,介绍了电路的具体设计方案、硬软件实施方法,并在 理论分析的基础上进行了验证及 改进。

实验表明,该电路电压偏 差小于0. 5V,可满足工程化应用。

图1:电路设计方案原理框图【关键词】APD 偏压自适应电路设计1引言雪崩光电二极管(avalanche photodiode ,APD)具有体积小、灵敏度高、响应速度快等 特点,特别是在内部雪崩倍增时可将信号倍增 上百倍,且倍增后的噪声仅与运放本底噪声水平相当,从而极大地提髙了系统的信噪比,被 广泛应用于光纤通信、激光测距、星球定向和军事测控等领域。

APD 工作时的信噪比(SNR)为:2q(I p + I DA )BM !F + 2qI DS (J)式(1)中:M 为APD 的雪崩增益,I ”为M=1时的光电流,和输入光信号功率成正 比,I da 为参与倍增的暗电流,I ds 为不参与倍 增的暗电流,B 为带宽,F 为过剩噪声系数,K 为波尔兹曼常数,T 为绝对温度,陽为负载, q 为输入光信号功率。

通过式(1)可以看出, 在APDI 作时随着雪崩增益M 的增大,信噪 比也逐渐增加;M 继续增大信噪比反而会变 小,故存在一个最优雪崩倍增因子Mp :M | 2KT |小式(2)中,x 为APD 的过剩噪音指数,其大小取决于APD 的结构和制作材料的不同。

从式(2)中可知,APD 的最佳雪崩增益与温度、 输入信号光功率、器件自身的暗电流及负载大 小等有关。

其中温度的影响最为突出,温度的变化是影响最佳雪崩增益的关键因素。

因使用环境的不同,APD 不总是工作在一个恒温的 状态。

当温度变化时,最优雪崩倍增因子也随 之发生改变。

根据作者在理论和实验的研究中 发现,当APD 增益比较恒定时,其偏压Vb 与 温度T 之间存在一定的线性关系,该线性关系为:V b = ^L(0.51T-lI.98)+V BK ⑶式(3)中,Pp 是入注光功率,I ]是APD 的量子效率,V br 是PN 结的反向击穿电压。

电气类外文翻译

电气类外文翻译

1、外文原文(复印件)A: The Utility Interface with Power Electronic SystemIntroductionWe discussed various powerline disturbances and how power electronic converters can perform as power conditioners and uninterruptible power supplies to prevent these poweline disturbances from disrupting the operation of critical loads such as computers used for controlling important processes, medical equipment, and the like. However, all power electronic converters (including those used to protect critical loads) can add to the inherent powerline disturbances by distorting the utility waveform due to harmonic currents injected into the utility grid and by producing electromagnetic interference, To illustrate the problems due to current harmonics ih in the input current i s of a power electronic load, consider the simple block diagram of Fig. 1-6A-1. Due to the finite (non-zero) internal impedance of the utility source which is simply represented by Ls in Fig. l-6A-1, the voltage waveform at the point of common coupling to the other loads will become distorted, which may cause them to malfunction. In addition to the voltage waveform distortion, some other problems due to the harmonic currents are as follows: additional heating and possibly overvoltages (due to resonance conditions) in the utility's distribution and transmission equipment, errors in metering and malfunction of utility relays, interference with communication and control signals, and so on. In addition to these problems, phase-controlled converters cause notches in the utility voltage waveform and many draw power at a very low displacement power factor which results in a very poor power factor of operation.The foregoing discussion shows that the proliferation of power electronic systems and loads has the potential for significant negative impact on the utilities themselves, as well as on their customers. One approach to minimize this impact is to filter the harmonic currents and the electromagnetic interference (EMI) produced by the power electronic loads. A better alternative, in spite of a small increase in the initial cost, may be to design the power electronic equipment such that the harmoniccurrents and the EMI are prevented or minimized from being generated in the first place. Both, the concerns about the utility interface and the design of power electronic equipment to minimize these concerns are discussed here.Generation of Current HarmonicsIn most power electronic equipment, such as switch-mode dc power supplies, uninterruptible power supplies (UPS), and ac and dc motor drives, ac-to-dc converters are used as the interface with the utility voltage source. Commonly, a line-frequency diode rectifier bridge as shown in Fig.1-6A-2 is used to convert line frequency ac into dc. The rectifier output is a dc voltage whose average magnitude Ud is uncontrolled.A large filter capacitor is used at the rectifier output to reduce the ripple in the dc voltage Ud. The dc voltage Ud and the dc current Id are unipolar and unidirectional, respectively. Therefore, the power flow is always from the utility ac input to the dc side. These line-frequency rectifiers with a falter capacitor at the dc side were discussed in detail in other section.A class of power electronic systems utilizes line-frequency thyristor-controlled ac-to-dc converters as the utility interface. In these converters, which were discussed in detail, the average dc output voltage Ud is controllable in magnitude and polarity, but the dc current Id remains unidirectional. Because of the reversible polarity of the dc voltage, the power flow through these converters is reversible. As was pointed out, the trend is to use these converters only at very high power levels, such as in high-voltage dc transmission systems. Because of the very high power levels, the techniques to ffdter the current harmonics and to improve the power factor of operation are quite different in these converters, as discussed in other section, than those for the line-frequency diode rectifiers.The diode rectifiers are used to interface with both the single-phase and the three-phase utility voltages. Typical ac current waveforms with minimal filtering were shown in other section. Typical harmonics in a single-phase input current waveform are listed in Table 1-6A-1, where the harmonic currents Ih are expressed as a ratio of the fundamental current Il. As is shown by Table 1-6A-l, such current waveformsconsist of large harmonic magnitudes. Therefore, for a finite internal per-phase source impedance Ls, the voltage distortion at the point of common coupling in Fig. 1-6A-1 can be substantial. The higher the internal source inductance Ls, the greater would be the voltage distortion.Current Harmonics and Power FactorAs we discussed in other section, the power factor PF at which an equipment operates is the product of the current ratio Il / Is and the displacement power factor DPF:In Eq. (1-6A-I), the displacement power factor equals the cosine of the angle Φ1. The current ratio Il / Is in Eq. (1-6A-l) is the ratio of the rms value of the fundamental frequency current component to the rms value of the total current. The power factor indicates how effectively the equipment draws power from the utility; at a low power factor of operation for a given voltage and power level, the current drawn by the equipment will be large, thus requiting increased volt-ampere ratings of the utility equipment such as transformers, transmission lines, and generators. The importance of the high power factor has been recognized by residential and office equipment manufacturers for their own benefit to maximize the power available from a wall outlet. For example from a 120V, 15A electrical circuit in a building, the maximum power available is 1.8 kW, provided the power factor is unity. The maximum power that can be drawn without exceeding the 15A limit decreases with decreasing power factor. The foregoing arguments indicate the responsibility and desirability on the part of the equipment manufacturers and users to design power electronic equipment with a high power factor of operation. This requires that the displacement power factor DPF should be high in Eq. (1-6A-I). Moreover, the current harmonics should be low to yield a high current ratio I1 / Is in Eq. (1-6A- 1).B: A Three-phase Pre-converter for Induction HeatingMOSFETBridge InvertersIntroductionHigh frequency power supplies, based on MOSFET bridge inverters, are already widely used for induction heating applications. These units require dc input voltages of about 400V to allow efficient operation of the MOSFETs employed. This supply voltage is usually obtained by using a three-phase rectifier stage, appropriate smoothing components or by employing thyristor phase- angle control to the mains supply. This kind of mains frequency power supply allows output power control of the induction heater, but it suffers from highly distorted input current waveforms with a low power factor. New legislation has been proposed to limit the maximum magnitude of harmonics drawn from the mains supply and different strategies have been suggested to reduce mains pollution.Investigations have been made to replace mains frequency power supplies by switched mode pre-converters. Switched mode converters can be designed to draw sinusoidal input currents thus avoiding the need for large and expensive mains frequency filters. At the same time these converters provide output power control and implementation of a small size high frequency isolation transformer. Power factor corrected three-phase ac-dc switched mode converter systems have usually been obtained using three identical single-phase converters with a common output filter. These systems overcome problems of mains pollution, but suffer from the disadvantage of a relatively large number of components and the need for complicated control and synchronization circuits. To reduce component costs, a structure based on a boost converter with three-phase input diode rectifier has been suggested. However, when operated direct-off-line from a three-phase 415V mains supply, this structure leads to high output voltages above lkV.In this paper, a novel method to achieve power factor correction for three-phase ac to dc power converters is described. The proposed topology is based on the buck converter and allows therefore output voltages to be below the maximum input voltage. The proposed topology utilizes a three- phase diode rectifier at the mains input and a single active switching device. The active switching device operates underzero-current switching conditions, resulting in very high converter efficiencies and low RFI emissions.Zero-current switching technique allows semiconductor devices to be operated at much higher switching frequencies and with reduced drive requirements compared with conventional switched mode operation.The proposed single-ended resonant converter with three-phase diode rectifier offers good opportunities for medium power, ac to dc applications. It combines simplicity and ease of control with high converter efficiency and high output power capabilities. It will be shown in the paper, that these characteristics make the converter very suitable as a direct replacement for the conventional mains frequency power supply used to supply induction heating MOSFET bridge inverters.General DescriptionA block diagram of the proposed induction heating system is shown in Fig. 1-6B-1. Block 1 represents the pre-converter that produces the dc supply voltage to feed to the RF MOSFET bridge inverter. Its output voltage should be controllable over a wide range to control the output power of the inverter and it must be able to operate with a wide range of load resistance to compensate load changes of the induction heating inverter stage. The pre-converter should operate direct-off-line from a three-phase 415V mains supply, drawing sinusoidal input current waveforms with a power factor approaching unity.Block 2 shows the RF MOSFET bridge inverter.The required maximum supply voltage of the MOSFET bridge lies between 300V and 400V. Block 3 represents the control and protection circuit used to stabilise the output power and to allow reliable operation of the induction heater in an industrial environment.Principle of Converter OperationA circuit diagram of the proposed three-phase ac to dc converter topology is shown in Fig. 1- 6B-2. The converter input currents are filtered through the input inductors L1, L2, L3. These inductors are designed so that the converter input currents are approximately constant over a whole switching cycle.During the OFF time of switch S, all three capacitors are charged by the inputcurrents I1, I2,I3. Consequently the three capacitor voltages Uc1, Uc1, Uc1 begin simultaneously to increase at a rate proportional to their respective input currents. If discontinuous operation is assumed the initial voltages of all capacitors C1, C2, C3 are zero when the switch ceases conducting. Hence, the peak voltage across each capacitor at the end of the OFF interval is proportional to their respective phase input current during the same OFF interval. Since capacitor voltages always begin at zero, it means that their average values during OFF time are linearly dependent on the phase input currents.During the ON time of switch S the energy stored in the three input capacitors C1, C2 and C3 is discharged through the six rectifier diodes VD1 –VD6, the switch S and the resonant inductor Lr. The rate of current decrease is dependent on the phase currents I1, I2, I3 and the switch current I0. The average value of the capacitor voltages Uc1, Uc2, Uc3 during the ON time are not linearly dependant on their phase input currents.To draw sinusoidal input currents from the mains supply the converter must draw input currents averaged over each switching cycle which are proportional to the phase voltages. Assuming steady state converter operation, the average phase input voltages over each switching cycle must be equal to the appropriate average input capacitor voltages during the switch OFF time plus the average input capacitor voltages during the switch ON time.Average input capacitor voltages during the switch OFF time have been shown to be proportional to the phase input currents, but during the switch ON time this is not true. However, if the switch ON time of the converter is mucteshorter than the switch OFF time, then the shape of the phase input currents will approach a sinusoidal waveform with unity power factor.2、外文资料翻译译文A:效用界面与电力电子系统介绍我们之前介绍了许多种电力线的干扰情况和电力系统转换器是如何在作为电力调节器和电力电子变换器时,用来防止那些电力线扰动干扰操作的临界荷载,例如电脑用于控制重要步骤,医疗设备,以及类似其他情况。

单光子激光测距淬灭电路设计优化

单光子激光测距淬灭电路设计优化

单光子激光测距淬灭电路设计优化作者:陈雨羊毅郝培育李尊来源:《航空科学技术》2018年第12期摘要:随着对激光测距测程要求的提高,以量子探测和概率统计理论为基础的单光子激光测距技术逐渐成为发展的新方向,單光子测距灵敏度高、测程远,探测器常用盖革模式下的雪崩光电二极管。

盖革模式下,探测器一旦响应,电流成倍增大,需要加上淬灭电路。

目前主动淬灭方式较为常用,但是噪声较大,电路设计复杂。

优化设计了GHz的门控淬灭方式,将高频正弦信号加载在探测器两端,在正弦信号正半周期探测器处于盖革模式,负半周期淬灭探测器,同时门控信号的存在降低了电路的噪声。

把主动淬灭电路和门控淬灭电路进行了研究与仿真,结果表明,正弦门控电路死时间短,噪声低,探测效率高,性能较优。

设计了正弦门控电路。

关键词:单光子;主动淬灭电路;正弦门控淬灭电路,测距;激光中图分类号:TN958.98 文献标识码:A单光子测距测程远,能对微弱光信号产生很好的响应。

盖革模式雪崩光电二极管(Geiger Mode of AvalanchePhotodiodes,GM-APD)的工作电压高于雪崩击穿电压,对入射光子高量子效率转换和极高雪崩内增益放大,响应信号(或噪声)后,为了保证正常工作,必须采用淬灭电路将APD的工作偏压降低到雪崩击穿电压以下,来清除所有的自由载流子,再将APD的工作偏压提高到雪崩击穿电压以上,为探测下一个光子做好准备。

20世纪60年代,Haitz等在GM-APD雪崩击穿工作原理的研究过程中提出了被动淬灭电路[1,2],并提出了一个GM-APD电学模型。

1975年,意大利米兰理工大学Cova采用Haitz 提出的结构,针对GM-APD被动式淬灭死时间长的缺点,提出了主动淬灭电路[3]。

1981年,Cova证明了GM-APD皮秒级(ps)的分辨率及其应用在光学时间技术相关领域的潜力,提出门控式结合主动式的淬灭电路,缩短了 GM-APD死时间[4]。

APD单光子探测的电路设计

APD单光子探测的电路设计

APD单光子探测的电路设计王凡;蒋书波;胡佳琳【摘要】在气体分析领域,由于分子密度的减小,拉曼技术很难获得足够强的信号,为提高检测灵敏度,利用雪崩光电二极管设计了单光子探测器,来检测微弱的拉曼光。

系统围绕APD设计了3个主要模块:偏置/测试电源、温控模块、信号调理。

测试了系统的暗计数率,并用标准气校验了系统的准确度。

实验结果表明:标准差最大为0.905,按总量程计算可得重复性相对偏差为0.905%,而非线性误差取最大引用误差0.13%。

其多次测量结果的线性度很好,能够用于线性检测。

%In the field of gas analysis due to the reduced moleculardensity,Raman technique is difficult to obtain a suf⁃ficiently strong signal,a single photon detector is designed based on an avalanche photodiode to improve the detection sensitivity of the weak Raman light . The system has four main modules around APD:offset/test power supply,tempera⁃ture control module,signal conditioning,pulse output. The dark count rate of the system is tested,and by using the stan⁃dard gas the accuracy of the system was calibrated. The results showed that standard deviation is up to 0.905,accord⁃ing to the total process,reproducibility relative standard deviation can be calculated to 0.905%,while the non-linear er⁃ror to take as maximum reference error is 0.13%. Its good linearity measurements can be used for linear detection.【期刊名称】《电子器件》【年(卷),期】2016(039)005【总页数】5页(P1093-1097)【关键词】单光子探测;硅雪崩光电二极管;雪崩抑制;气体拉曼分析;偏置电源【作者】王凡;蒋书波;胡佳琳【作者单位】南京工业大学电气工程与控制科学学院,南京211816;南京工业大学电气工程与控制科学学院,南京211816;南京工业大学电气工程与控制科学学院,南京211816【正文语种】中文【中图分类】TP116.02近年来,光子计数技术发展迅速,在光谱测量、无损检测、高能物理、量子通讯、生物医疗、天文观测等领域有着广泛的应用[1-2]。

APD高压电路的设计

APD高压电路的设计

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在实际应用中,DC/DC Converter芯片的输出电压经分压得到的反馈电压 与芯片内部的参考电压进行比较,产生一个误差信号经由PWM控制器(根据 误差信号产生不同的占空比δ来控制信号)来控制功率管的开关。当误差放大 器EA的输出增加时,输出的开关电流增加;当EA的输出减小时,输出的开关 电流减小,从而实现高压输出的自动调节。
二、 APD高压电路
目前光模块的工作电压一般为3.3V或5V,而APD所需的工作电压高达几十伏。 为保证APD的正常工作,需要引入高压电路及相应的温度补偿措施。APD高压电 路主要包括升压电路、倍压电路 和温度补偿 三个部分。
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倍压电路
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1. 升压电路
光模块内部采用的升压电路一般都是非常成熟了的能够实现DC/DC转换功能的 专用升压芯片,如Linear的LT1930、Maxim的MAX1771等。通过DC/DC Converter 能将输入的电源电压(3.3V或5V)转换成20、30几伏的高压输出。
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APD高压 APD高压
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APD偏压电路的最佳设计 - 外文翻译APD偏压电路的最佳设计孙纯生,秦世桥,王兴书,朱冬华1 .国防科技大学光电科学与技术学院,中国长沙4100732 . 海军工程大学装备工程部,中国武汉430033提出了一种基于温度补偿和负载电阻补偿的雪崩光电二极管反向偏压控制方法,并详细的分析了背景光和负载电阻对雪崩光电二极管检测电路的影响。

为雪崩光电二极管偏置电路的设计建立了一种理想的温度补偿和负载电阻补偿模型。

据预测,这种控制方法特别适用于车辆使用的激光测距仪。

实验结果证实,提出的设计可以很大程度的改善测距仪的性能。

雪崩光电二极管的特点是具有很高的量子效率和教大的内部增益,这可以很大程度的降低对前置放大电路性能的要求,并能提高检测电路的信噪比(SNR)。

因此,它具有很广泛的用途,如光纤通信、激光测距仪、微弱信号探测器等。

为了使检测电路能获得最佳检测性能,APD的外部电压需要接近最佳倍增因子时的电压。

于最佳倍增因子是许多因数的复函数,如:外部温度、背景光通量、放大器噪声和系统带宽,因此需要设计一个复杂的反馈控制电路及时的调整雪崩光电二极管的偏压。

当然这就增加了开销。

介绍了一种简单的、避免高开销的方式,就是确保温度补偿的同时给APD偏置电路选择一个合适的负载电阻。

通过这种方式,背景光对雪崩光电二极管检测电路造成的不良影响可在一定程度上得到补偿,并且检测电路抗背景光能力得到了改善。

在这种方法基础上为汽车防撞设计的激光测距仪能很好地满足系统的要求。

APD激光检测电路的主要噪声源包括检测器噪声、负载电阻噪声、放大电路前端噪声,还有背景光电流和信号光电流造成的散粒噪声。

当前的信噪比可以按照下列方程式计算:方程1右边分子部分是光信号电流。

方程1右边分母部分是噪声电流,包括三个方面。

第一项是背景光电流和信号光电流造成的散粒噪声,第二项是检测器噪声,最后一项是负载电阻噪声和跟随放大电路的等效噪声。

在方程中,Ps代表检测器接收到的光信号功率,M是APD的倍增增益,Ro是当M=1时的电流灵敏度,e是电子的电荷量,等于×10-19C,B是检测电路的通频带宽,Pb是检测器收到的背景光功率,FA是APD的过量噪声系数,ids是APD表面漏电流,idb是负载漏电流,K是玻耳兹曼常数,等于× 10-23 JK-1,T 是检测器负载电阻的温度(K),Rl是检测器的负载电阻(Ω),Fn是放大电路的等效输入噪声系数。

M 于实际使用中M是远远大于1的,暗电流可表示为id≈idb,而APD过量噪声因子FA=kM+(1-k)(2-1/M)≈2+kM。

在这一近似条件下,当d(SNRi)/dM=0,SNRi达到其最大值且倍增因子达到最佳,可表示为:方程2中符号和方程1中符号的含义相同。

我们知道,最佳倍增因子是外部温度、光信号功率、背景光功率,APD 噪声、光谱灵敏度、放大器噪声和系统带宽的函数。

此外,特别是APD内部结构决定了其倍增增益M随工作温度变化而变化。

用温度系数CT来描述这种影响。

对于C30737系列的APD,CT为 V/℃,这意味着在相同条件下,当APD的工作温度增加1℃,为了维持APD倍增因子不变偏压需要增加 V。

从前面一段的分析,我们知道,电路温度和背景光补偿旨在控制偏压,以便在不同温度和背景光条件下电路仍能保持最佳的APD倍增因子。

目前有几种偏置电路控制方法:恒流偏置,温度补偿和恒虚报警控制。

恒流偏置是只适用于不变的背景光或无背景光情况。

温度补偿抗背景光的能力较差。

恒虚假控制可以保持最佳的倍增因子,但复杂的电路和高成本才换来较高的性能。

提出了一种新方法,为APD偏压电路设计了温度补偿以及串行电阻背景光补偿,实现高性能的同时保持低成本。

温度变化对APD偏置电路的影响主要在两个方面:一是温度变化使负载电阻噪声发生变化,因而改变了APD检测电路的最佳增殖因子;另一方面,温度变化改变了APD载流子和晶格之间的碰撞频率和强度,这也改变了APD的倍增因子。

以下就是分析这两个因数的影响。

APD倍增因子M和其反向偏置电压V之间的关系可以用下式描述:其中V是APD的反向偏置电压,VB是某一确定温度时的击穿电压,n介于1和3之间,它半导电材料、半导体掺杂分配和辐射源的波长决定。

在方程3中,当M达到最佳值Mopt 时反向偏置电压达到最佳Vopt。

从方程2和3我们能够得到最优偏置电压Vopt、工作温度和接收到的背景光功率Pb之间的关系如下:方程4只包括APD偏置电路的温度对负载电阻噪声的影响,例如上文提到过的一个方面。

温度变化对APD倍增因子影响可表示为温度系数Ct。

以最佳工作电压V22 为22℃作为参考点,温度变化引起的最佳偏置电压的变化可以描述为:方程5右边前面两项的和ΔV1表明负载电阻噪声对最佳偏置的影响。

方程5右边第三个项ΔV2表明工作温度对APD偏置的影响。

公式5显示了当温度变化时如何控制偏置电压优化倍增系数。

这仅仅是APD偏置电压工作的温度补偿模型。

从方程5,我们知道ΔV不仅取决于APD的工作温度T,而且还取决于击穿电压VB、接收光信号功率和背景光功率。

因此,方程5是不符合实际工程的。

实际中,ΔV2>>ΔV1,所以方程5可以近似为:APD的温度系数和22 ℃时的最佳工作电压可在设备手册里获得,工作温度可用温度传感器测出。

因此,APD偏压相对于V22的补偿可通过方程6得到。

温度补偿模拟温度传感器、模拟数字转换A / D转换器、微控制器和可调直流电源供应电路组成,其电路框图如图1所示:模拟温度传感器置于APD附近监测其工作温度。

A / D 转换器将模拟温度信号变量转换成数字信号变量。

单片机的作用是将A/D转换器提供的数字温度信号转变成相应的控制信号,并根据方程6和控制方程调整数字电位器的输入电阻值调整直流电源供应。

通过调整输入电阻值来调整直流适配器,这样就能输出合适的APD偏压。

温度补偿、背景光补偿可以通过适当选择的负载电阻实现。

因此检测电路的抗背景光干扰能力可以大大改善。

图2显示了APD检测电路的偏置电路及其外围电路,其中V0是V图1.温度补偿电路原理图温度补偿后输入的可调的直流电压,π型滤波器电容C1、C2和电阻RC组成,输出脉冲信号的读出电路电容C3, 电阻Rf和运算放大器AV组成。

Rl是APD偏压电路的直流负载电阻。

我们将在下面分析APD偏压电路负载电阻对抗背景光能力的的影响。

从图2,我们知道,APD偏置电压的性能可表示为:图2.APD的外围电路方程3和7,偏置电压Vapd、背景光功率P b和负载电阻RC间的关系可以表示为:基于温度补偿,并假设只考虑背景光功率PVopt和Pb 对APD最佳偏置电压Vopt的影响,b之间的关系给出如下:如果方程8中工作电压等于方程9的最佳值,温度补偿后APD获得最佳倍增因子,检测电路信噪比也能达到最高水平。

为了检查上述方法的有效性,我们在汽车半导体激光测距仪设计中运用此方案,并设定具体参数来检测150米远的目标。

如图3所示,APD偏压Vapd、背景光功率和串行电阻RC之间的关系可通过对方程8和9进行数值计算和分析得到。

通过同样的方式,如图4,检测电路当前的SNRi、背景光功率和串行电阻RC之间的关系可以通过方程1、8和9得到,这里Ps=30 nW, R0=, B=35 M, P在背景光较弱时,Pb=0~500nW。

b=200nW, idb=8×10-11A, k=, Rl=400 Ω, Fn=1, VB= V, V0= VB.图3显示了Rc取不同值时APD偏压和的背景光之间的关系,其中实线代表最佳偏压虚线代表工作偏压Vapd和背景光之间的关系。

该图显示了Vopt和Vopt和背景光之间的关系。

Vapd随着Rc变化的补偿。

因而可以找到一个最佳的Rc 使Vopt和Vapd保持一致。

图4对应于图3,显示了Rc取不同值时检测电路当前SNR和背景光之间的关系,其中实线呈现了最佳SNR和背景光间的关系,虚线显示了Rc取不同值时实际SNR和背景光间的关系。

该图显示了APD检测电路和最佳偏压因Rc不同而产生不同的偏移时的实际SNR。

为了优化检测电路的性能,认真选择Rc对保持实际SNR和最佳SNR 恒等非常重要。

从图3和4,我们知道背景光对检测电路造成的不良影响可以通过选择合适的Rc得到一定的补偿,并能够改善检测电路的性能。

图3.Rcs不同时Vapd和Pb间的关系图4.Rcs不同时SNRi和Pb间的关系这种雪崩光电二极管激光检测电路已广泛地运用于汽车防撞激光测距仪中,其性能测试的方法有两种。

途径之一是在恒定的背景光下改变雪崩光电二极管的负载电阻,这时激光测距仪测距能力范围变化很明显。

在明亮的背景光下,Rc为Ω时的测量范围比Rc等于5 MΩ或100 kΩ的范围大10 %~20 %左右。

另一种方式是在相同的测量范围下改变APD负载电阻,这时测量的抗背景光干扰能力具有明显的差距。

在同一测量范围,Rc为Ω对比Rc等于5 MΩ或100 kΩ抗背景光能力增加了20 %~30 %。

从以上理论分析和实验结果,我们发现基于负荷电阻的温度补偿和背景光补偿的APD偏压控制方法可以大大地提高相同条件下APD检测电路的检测能力范围和抗背景光能力。

这种方法的电路设计特点是结构简单,成本低,所以它是一个实际可行的项目。

APD偏压电路的最佳设计孙纯生,秦世桥,王兴书,朱冬华1 .国防科技大学光电科学与技术学院,中国长沙4100732 . 海军工程大学装备工程部,中国武汉430033提出了一种基于温度补偿和负载电阻补偿的雪崩光电二极管反向偏压控制方法,并详细的分析了背景光和负载电阻对雪崩光电二极管检测电路的影响。

为雪崩光电二极管偏置电路的设计建立了一种理想的温度补偿和负载电阻补偿模型。

据预测,这种控制方法特别适用于车辆使用的激光测距仪。

实验结果证实,提出的设计可以很大程度的改善测距仪的性能。

雪崩光电二极管的特点是具有很高的量子效率和教大的内部增益,这可以很大程度的降低对前置放大电路性能的要求,并能提高检测电路的信噪比(SNR)。

因此,它具有很广泛的用途,如光纤通信、激光测距仪、微弱信号探测器等。

为了使检测电路能获得最佳检测性能,APD的外部电压需要接近最佳倍增因子时的电压。

于最佳倍增因子是许多因数的复函数,如:外部温度、背景光通量、放大器噪声和系统带宽,因此需要设计一个复杂的反馈控制电路及时的调整雪崩光电二极管的偏压。

当然这就增加了开销。

介绍了一种简单的、避免高开销的方式,就是确保温度补偿的同时给APD偏置电路选择一个合适的负载电阻。

通过这种方式,背景光对雪崩光电二极管检测电路造成的不良影响可在一定程度上得到补偿,并且检测电路抗背景光能力得到了改善。

在这种方法基础上为汽车防撞设计的激光测距仪能很好地满足系统的要求。

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