HFSS 与腔体滤波器设计

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hfss腔体滤波器设计实例

hfss 腔体滤波器设计实例在微波带通滤波器的设计中，我们经常采用腔体交指型结构。

它具有插损小、带外抑制度高、结构紧凑、体积小等优点。

对于腔体交指型带通滤波器的设计，现在比较广泛的的思路是：只考虑相邻两耦合杆之间的耦合关系，忽略相邻杆以外的边缘电容的影响，因而采用两个沿结构传输的TEM 正交模来描述，即奇模和偶模。

而实际在这种滤波器结构中所有的谐振杆之间都存在耦合，因此这种方法只是一种简化的近似设计。

采用这种方法设计的产品性能差，表现在带内插损和波纹大，矩形系数不好等，一般无法满足现在通讯的要求，我们还要花大量的精力对滤波器进行调整，以提高其性能。

甚至需要重新加工再生产，这大大增加了产品的研制成本和周期。

因此我们必须对滤波器进行精确的设计，即在工程设计中将所有谐振杆的耦合都考虑进去，而这不是传统的手工计算可以完成的，必须借助计算机软件进行辅助设计。

自上世纪70 年代以来，CAD 工具在微波工程领域得到越来越广泛的应用。

经过多年的发展，目前国内外已有多种微波CAD 软件，而以Ansoft公司的HFSS 效果最佳。

通过该软件我们可以方便的得到各种物理模型，进而对该模型进行电磁场的仿真。

计算结束后我们就可以得到所需的场结构和相关的S 参数，也就知道了该滤波器的电性能情况。

本文用一个实例介绍了一种设计思路，借助计算机利用Ansoft 公司的HFSS 软件对腔体交指型滤波器进行精确设计，实验表明用这种方法设计的滤波器有通带平坦、插损小、精确度高等特点。

hfss 腔体滤波器设计实例下面通过一个S 波段的五级滤波器的设计实例加以说明。

首先我们通过简化的近似计算得到该滤波器的几何数据的初值，由于这类滤波器的粗略设计的方法已经很成熟，因此这里不进行详细介绍，直接给出（详细情况可参看《现代微波滤波器的结构与设计》）。

但这一步也是非常重要的，初值的好坏直接关系到我们利用软件计算优化的快慢。

我们知道，对交指型滤波器的理论分析由平行耦合线滤波器演化而来。

基于HFSS的微调谐腔体带通滤波器设计

基于HFSS的微调谐腔体带通滤波器设计发表时间：2016-10-12T14:41:17.417Z 来源：《电力设备》2016年第14期作者：李婷婷[导读] 针对微调谐腔体带通滤波器设计制造中存在的问题，介绍了腔体带通滤波器的总体设计。

（广州海格通信集团股份有限公司）摘要：针对微调谐腔体带通滤波器设计制造中存在的问题，介绍了腔体带通滤波器的总体设计；论述了需要解决的问题，如优化计算、提高仿真精度和简化调谐结构，并对二端口网络等效替换、整体仿真和微调谐关键技术进行了分析。

关键词：腔体带通滤波器；微调谐；免调谐；HFSS传统的微波腔体带通滤波器的设计过程中，参数计算量大，仿真存在误差，调谐过程耗时费力。

随着腔体带通滤波器在微波通信设备中的广泛应用，其设计方法有待改进。

通过设计参数求取方法的改进和对原理图的完善补充以及采用合理的仿真过程，确保了滤波器设计的精确度。

在此基础上，摒弃传统的用调谐螺钉调谐的方式，采用微调谐结构的腔体来实现滤波器的微调谐，配合线切割加工工艺，最终实现腔体带通滤波器的精确微调谐设计，一定的相对带宽条件下，可实现免调谐设计。

一、总体设计微波腔体带通滤波器的设计过程大体分为3步：一是按设计要求求取设计参数；二是进行滤波器模型的仿真；三是进行滤波器的调谐。

求取设计参数一般先根据设计要求选择合适的切比雪夫低通原型滤波器，因为较之最大平坦型滤波器，切比雪夫滤波器有更优异的带外抑制，较之椭圆函数滤波器更易于实现。

进行模型仿真前，还需要得到以下3个参数：一是单端输入最大群时延；二是谐振器间耦合系数；三是谐振腔的谐振频率。

滤波器模型的仿真分为2步：第一步要在HFSS中建立滤波器的三维微调谐模型；第二步就是进行HFSS的模型仿真。

在HFSS中建立滤波器腔体模型后，对其先后进行单谐振器本征模仿真、双谐振器本征模仿真和单端输入最大群时延仿真，分别得到单谐振器谐振频率、相邻谐振器间耦合系数和单端输入最大群时延等参数的仿真值。

HFSS高性能平行耦合微带带通滤波器设计与仿真攻略

HFSS高性能平行耦合微带带通滤波器设计与仿真攻略实现射频带通滤波器有多种方法，如微带、腔体等。

腔体滤波器具有Q值高、低插损和高选择性等特点，但存在成本较高、不易调试的缺点，并不太适合项目要求。

而微带滤波器具有结构紧凑、易于实现、独特的选频特性等优点，因而在微波集成电路中获得广泛应用。

常用的微带带通滤波器有平行耦合微带线滤波器、发夹型滤波器、1/4波长短路短截线滤波器、交指滤波器等形式以及微带线的EBG (电磁带隙)、DGS(缺陷地结构)等新结构形式。

而平行耦合微带带通滤波器具有体积小、重量轻、易于实现等优点。

01平行耦合带通滤波器的基本原理平行耦合带通滤波器是一种分布参数滤波器滤波器，它是由微带线或耦合微带线组成，其具有重量轻、结构紧凑、价格低、可靠性高、性能稳定等优点，因此在微波集成电路集成电路的供应商中，它是一种被广为应用的带通滤波器。

滤波器的基础是谐振电路，它是一个二端口网络，对通带内的频率信号呈现匹配传输，对阻带频率信号失配而进行发射衰减，从而实现信号频谱过滤功能。

微波带通滤波器在无线通信系统通信系统中起着至关重要的作用，尤其是在接收机前端。

滤波器性能的优劣直接影响到整个接收机性能的好坏，它不仅起到频带和信道选择的作用，而且还能滤除谐波，抑制杂散。

02平行耦合带通滤波器结构与模型的创建平行耦合带通滤波器原理平行耦合单元由两根相互平行且有一定间距的微带线组成，其结构图包括介质层、接地层和微带线如图 3.3 所示。

图中每根微带线的宽度和厚度分别为为W 和t；两根微带线的间距为S；介质层厚度和介电常数分别为h 和Er。

两根微带线通过接底层产生了耦合效应，随之产生了奇模和偶模特征阻抗。

平行耦合带通滤波器通过级联平行耦合线元件得到。

平行耦合带通滤波器的相对带宽BW 与中心频率、上边频和下边频有关，而奇模和偶模特征阻抗由低通滤波器参数g、滤波器输入输出端口特征阻抗Zo和耦合单元组成。

可由以下公式得到：平行耦合带通滤波器参数计算与设计本节中所设计的平行耦合带通滤波器指标如下表所示：根据表中滤波器指标，选择0.1dB纹波的切比雪夫滤波器来设计，阶数为5阶。

腔体式带通滤波器的研究与设计

西安科技大学硕士学位论文腔体式带通滤波器的研究与设计姓名：***申请学位级别：硕士专业：通信与信息系统指导教师：***2011论文题目：腔体式带通滤波器的研究与设计专业：通信与信息系统硕士生：刘健（签名）指导老师：刘新良（签名）摘要近年来，随着移动通信、电子对抗和导航技术的飞速发展，对新的微波元器件的需求和现有器件性能的改善都提出了很高的要求。

微波带通滤波器作为一种重要的微波元器件在近几年来也得到了大力的发展。

因此，对微波滤波器理论和设计方法的研究，已经引起了国内外器件工程师的极大兴趣。

本文以腔体式带通滤波器为研究的对象，采用综合法的经典公式与计算机仿真工具相结合的方法简化了设计过程，提高了设计和加工的准确性。

在整个研究的过程中，概括起来主要做了以下几个方面的工作：1. 从滤波器的网络设计理论入手，在耦合谐振腔带通滤波器的理论基础上，研究了从低通原型滤波器到耦合谐振腔可调带通滤波器的设计过程。

2. 针对腔体式带通滤波器的设计，研究分析了滤波器频率变化和滤波器性能参数之间的关系，得出实际设计时所需参数和滤波器结构的设计公式。

3. 依据设计指标，明确采用切比雪夫函数带通滤波器，并利用HFSS仿真软件对几何尺寸参数的初值进行了仿真、优化，以得到滤波器几何尺寸参数的终值，使其能够满足最初的设计指标要求，最终的仿真结果说明了这种方法的可行性和实用性。

关键词：带通滤波器；微波滤波器；同轴腔；切比雪夫滤波器；HFSS研究类型：应用研究Subject : The Cavity Asana Band-pass Filter Research and Design Specialty ：Communication and Information SystemName ： Liu Jian (Signature)Instructor：Liu Xin-liang (Signature)ABSTRACTWith the rapid development of the mobile communication industry，the electronic countermeasure and the technologies of navigation in recent years，the demand of new microwave components and the requirement of improving the quality of the existing microwave components are very high．Therefore the Microwave Band-pass filter, as an important microwave component, is well developed in recent years. Domestic and foreign engineers are very interested in the research of Microwave filters theory and practical design for the microwave filters．This paper chooses cavity asana band-pass filter as research object, combining the method of the synthetic classic formula with computer simulation tools to study, therefore simplifies the design process, and improve the accuracy of designing and machining. In the process. The study mainly includes several aspects:1.Starting from the network design theory of filter, based on coupling resonatorband-pass filter theory, this paper studies the design process developed from Low-pass prototype filter to Coupling resonance cavity adjustable band-pass filter.2. To design cavity band-pass filter, this paper researches and analyzes the relationshipbetween filter frequency variation and filter performance parameters. As a result, it finds out the designing formula.3. According to the design index, this research makes use of Chebyshev functionband-pass filter, and uses HFSS simulation software to simulate and optimize the initial geometric parameters to get the final value of geometric parameters of filter, so it can meet its original design requirements. The final simulation results demonstrate the feasibility and practicability of this method.Key words：Band-pass filter Microwave filter Coaxial-cavity Chebyshev HFSS Thesis ： Application Research1 绪论1 绪论1.1 滤波器概述当前，无线通信技术高速发展，业务范围不断扩大，人们对无线产品的需求迅速增长。

HFSS高性能平行耦合微带带通滤波器设计与仿真攻略

HFSS高性能平行耦合微带带通滤波器设计与仿真攻略HFSS（High Frequency Structural Simulator）是一款广泛应用于高频电磁场仿真的软件工具，具有高效准确的计算能力，广泛应用于微波通信、天线设计、微带滤波器设计等领域。

在微带带通滤波器设计中，HFSS软件可以帮助工程师快速准确地设计出性能优异的滤波器，提高设计效率和准确性。

本文将介绍HFSS软件在高性能平行耦合微带带通滤波器设计与仿真中的一般步骤和攻略。

一、平行耦合微带带通滤波器原理平行耦合微带带通滤波器是一种结构简单、性能良好的微带滤波器，通常由一组垂直耦合微带谐振器和几个开路微带谐振器组成。

通过合理设计电路结构中的微带谐振器的长度、宽度和耦合间隔等参数，可以实现所需的滤波特性。

平行耦合微带带通滤波器通常具有较低的插入损耗、较高的带宽和较好的阻带衰减等性能。

二、HFSS平行耦合微带带通滤波器设计步骤1.确定滤波器的工作频率和性能指标，如通带中心频率、通带带宽、阻带衰减等；2.设计滤波器的电路拓扑结构，包括微带谐振器的种类和数量、耦合方式等；3.利用HFSS软件建立滤波器的三维模型，并设置仿真参数，如工作频率、网格精度等；4.通过HFSS软件进行电磁场仿真，分析滤波器的传输特性和谐振器的工作状态，调整设计参数以满足性能指标；5.优化滤波器的结构设计，如微带谐振器的长度、宽度和耦合间隔等参数；6.在HFSS软件中进行频域和时域仿真，验证滤波器的性能指标是否满足设计要求；7.在满足性能指标的前提下，进一步优化滤波器的结构设计，以降低损耗和提高性能；8.导出最终的滤波器设计文件，用于制作和验证实际器件性能。

1.合理选择HFSS软件版本和许可证类型，确保软件功能和性能满足设计需求；2.熟练掌握HFSS软件的操作界面和基本功能，包括建模、设置仿真参数、网格划分、分析结果等；3.在建立滤波器的三维模型时，注意设计精度和模型简化，提高仿真效率和准确性；4.在仿真过程中，结合HFSS软件的参数优化功能，快速有效地调整设计参数，实现滤波器性能的优化；5.结合HFSS软件的频域和时域仿真功能，全面分析滤波器的传输特性和动态响应，确保性能指标的准确性；6.在滤波器设计的不同阶段，及时保存和备份仿真文件和结果，方便后续验证和分析；8.最终，通过HFSS软件的仿真和验证结果，确定滤波器的结构设计方案，并导出制作文件进行实际器件的制作和测试。

3.4GHz梳状线腔体滤波器的设计.

本科生毕业论文设计题目： 3.4GHz 梳状线腔体滤波器的设计系部学科门类工学专业电子信息工程学号姓名指导教师年月日装订线3.4GHz梳状线腔体滤波器的设计摘要在当今通信领域中，微波滤波器在通信设备中占有重要的地位，在微波毫米波通信、卫星通信、雷达、导航、制导、电子对抗、测试仪表等系统中，有着广泛的应用。

梳状线滤波器具有小体积、高Q值、高功率容量等优点，是微波滤波器中常见的腔体形式，工程实用性较强，广泛应用于通信及其它领域。

本文从滤波器的工作原理出发，分析了梳状线带通滤波器的结构特征，并利用软件Ansoft HFSS进行仿真，最后基于仿真结果制作出实物并进行了调试，使其最终达到预期的指标。

关键词：梳状线滤波器仿真调试ABSTRACTIn the field of current communication, Comb-line filters occupies an important position in communication equipment. Microwave filters has a wide range of applications in microwave communication, millimeter wave communication, satellite communication, radar, navigation, guidance, electronic against, testing instruments system. Comb-line filters have small size, high Q value, high power capacity etc, and is common in microwave filters of the recessed forms, therefore it widely used in communications and other fields . Based on the theory of filters, the structure characters of comb-line band-pass filter have been analyzed and the typical parameters have been calculated. Then the filter is simulated with software Ansoft HFSS. At last, I have manufactured a practicality based on the results of simulation and debugged it for the purpose of achieving anticipative targets.Key words：Comb-line Filter Simulation Debug目录一绪论 (1)1.1 课题来源与意义 (1)1.2 国内外发展状况 (1)1.3 课题的研究内容、方法及手段 (1)二梳状线滤波器的综合介绍 (3)2.1 梳状线滤波器的特点 (3)2.2 梳状线滤波器的结构 (3)2.3 梳状线滤波器的工作原理 (3)三梳状线滤波器的设计 (4)3.1 梳状线滤波器设计思路 (4)3.2 梳状线滤波器的技术指标 (4)3.3 梳状线滤波器的归一化原型 (4)3.4 频率变换 (5)3.5 相关的理论计算过程 (5)四运用Ansoft HFSS进行仿真设计 (7)4.1 单腔模型及仿真结果 (7)4.2 双腔模型及仿真结果 (8)五梳状线滤波器的实物制作与测试 (11)六总结与结论 (12)参考文献 (13)一绪论1.1 课题来源与意义本课题来源于科研生产。

基于HFSS设计同轴腔调谐滤波器

基于HFSS设计同轴腔调谐滤波器贾建蕊;韩军【摘要】同轴腔调谐滤波器在军事通信设备中具有广泛应用.论述了该类滤波器的设计原理,详细分析了腔间耦合孔和输入、输出耦合环的位置和大小的设计,在此基础上应用高频结构仿真软件(HFSS),对L波段调谐滤波器的实例进行仿真设计.结果表明仿真拟合准确,说明应用HFSS仿真软件能够很好的描述调谐滤波器的关键设计内容.【期刊名称】《无线电工程》【年(卷),期】2011(041)001【总页数】4页(P44-46,60)【关键词】滤波器;调谐;耦合孔;耦合环【作者】贾建蕊;韩军【作者单位】中国电子科技集团公司第五十四研究所,河北,石家庄,050081;中国电子科技集团公司第五十四研究所,河北,石家庄,050081【正文语种】中文【中图分类】TN4540 引言调谐滤波器既能够实现频率抗干扰又能够满足最佳收、发滤波器要求,其通带频率能够随着工作频率要求的改变而相应改变,在通信系统抗干扰技术中的作用举足轻重。

腔体调谐滤波器设计主要存在以下2个难点:①在调谐过程中,滤波器的响应特性和绝对带宽随着中心频率的变化而发生显著变化,通带内插损增大,阻带抑制度下降;②在调谐过程中,中心频率随着耦合结构的变化呈非线性变化,耦合结构难以实现。

在此根据滤波器设计的基本原理,利用HFSS进行预仿真,很好地克服了上述2个缺点,实现了实例设计。

1 总体设计以L波段滤波器设计为例,主要设计参数如表1所示。

由于所设计的调谐滤波器的调谐范围为1 300～1 600MHz,为了便于照顾整个频段的相对带宽,选择中间点1 450 MHz为设计频点。

利用网络综合法,选取切比雪夫函数作为逼近函数,切比雪夫响应函数[1]。

在通带内是等波纹型的,通带内损耗LAr要足够小,取LAr=0.03。

经查表计算确定滤波器阶数n=4,对应的低通原型参数 gn可通过公式计算得:g0=1,g1=0.868 1,g2=1.275 4,g3=1.506 2,g4=0.735 1,g5=1.181 0。

hfss腔体滤波器设计实例

hfss腔体滤波器设计实例HFSS（High Frequency Structure Simulator）是一种用于电磁场仿真和分析的软件工具。

它广泛应用于高频电磁场的建模和分析，可用于设计各种射频（RF）和微波器件，如天线、滤波器、耦合器等。

本文将以HFSS腔体滤波器设计实例为题，介绍如何利用HFSS软件进行腔体滤波器的设计。

我们需要明确腔体滤波器的基本原理。

腔体滤波器利用腔体的谐振模式和谐振频率来实现信号的滤波。

通过调整腔体的几何参数和材料特性，可以实现对特定频率范围内的信号进行滤波。

因此，腔体滤波器的设计关键在于确定合适的腔体结构和参数。

接下来，我们将以一个实际的设计例子来具体介绍HFSS腔体滤波器的设计流程。

假设我们要设计一个工作在2.4GHz频段的微波腔体滤波器。

首先，我们需要选择合适的腔体结构。

常见的腔体结构有矩形腔体、圆柱腔体等，根据设计要求选择合适的结构。

在HFSS中，我们可以通过绘制几何模型来定义腔体结构。

绘制完成后，我们需要定义腔体的材料属性，包括介电常数、磁导率等。

这些参数将直接影响腔体的谐振频率和模式。

接下来，我们可以利用HFSS的求解器进行电磁场仿真。

在仿真前，我们需要设置仿真的频率范围和精度。

根据设计要求，选择合适的频率范围，并设置适当的网格精度。

仿真完成后，我们可以通过HFSS的结果分析工具来分析仿真结果。

主要包括频率响应、S参数、电场分布等。

根据设计要求，对仿真结果进行评估和调整。

如果需要改善滤波器性能，可以通过调整腔体的几何参数和材料特性来实现。

在设计过程中，需要注意以下几点。

首先，腔体的尺寸和几何参数应该合理选择，以满足设计要求。

其次，材料的选择和特性对滤波器性能影响很大，需要选择合适的材料并设置正确的特性。

最后，仿真结果的准确性和稳定性也需要重视，可以通过调整网格精度和求解器参数来提高仿真结果的准确性。

HFSS是一种强大的工具，可以用于腔体滤波器的设计和分析。

基于HFSS设计同轴腔滤波器

第30卷　第2期2007年4月电子器件Chinese J ournal Of Elect ron DevicesVol.30　No.2Ap r.2007Design of Coaxial Filters B ased On HFSSL I U Pen g 2y u1,2,Z H A N G Yu 2hu 2,S H EN H ai 2gen11.School of Elect ronic I nf ormation and Elect ric Engineering ,S hanghai J iao Tong Universit y ,S hanghai 200240,China;2.S hanghai S pacef li ght I nstit ute of T T &C and Telecommuniation ,S hanghai 200086,Chi naAbstract :Coaxial filters is widly used in microwave circuit s.we research how to analysis and design coaxial filters used by a 3D f ull 2wave field solver ,HFSS.The 3D f ull -wave field analysis includes t he effect s of t uning screw ,interstage coupling apert ure and inp ut/outp ut coaxial excitation.Base on t hess analysises ,we work out a S -band coaxial filter aided by simulating and optimizing in HFSS.The result of t he experi 2mentation matched well wit h t he result of simulation ,and f ulfiled technic target s.The coaxial filter has been used in a spaceflight project successf ully.The way of combining t he t raditional t heory wit h t he ad 2vanced comp uter technology has great practical value ,it can save much time and co st .K ey w ords :microwave filters ;coaxial resonator ;coupling apert ure ;HFSS EEACC :1320基于HFSS 设计同轴腔滤波器刘鹏宇1,2,张玉虎2,沈海根1(1.上海交通大学电子信息与电气工程学院,上海200240;2.上海航天测控通信研究所,上海200086)收稿日期:2006204217作者简介:刘鹏宇(19782),男,工作于上海航天测控通信研究所,工程师,主要研究方向为射频与微波电路设计,pengyu_liu @ ;摘　要:同轴腔滤波器在微波电路中有着广泛的应用,在此研究如何利用3D 全波场分析软件HFSS 分析设计同轴腔滤波器.该分析包括谐振腔调谐螺钉、腔间耦合孔及输入输出激励的影响效应.基于上述分析,借助HFSS 仿真优化得到一S 波段滤波器.其实测结果与仿真相符,满足指标要求,并已成功应用于某航天工程中.这种结合传统理论和先进计算机技术的方法可以大大节省研制周期和生产成本,具有非常大的实用价值.关键词:微波滤波器;同轴谐振腔;耦合孔;HFSS 中图分类号:TN 713 文献标识码:A 文章编号:100529490(2007)022******* 传统的微波滤波器设计方法已经非常成熟,但其中一些参数需要反复试验来获得.这势必要增加产品的设计周期,对于当前研制周期紧、产品数量大的要求是一个制约.利用仿真工具进行辅助设计成为目前一种非常有效的解决途径.本文即介绍如何借助H FSS 设计同轴腔滤波器.1　HFSS 简介HFSS 是ANSO F T 公司开发的一个基于物理原型的EDA 设计软件.使用H FSS 建立结构模型进行3D 全波场分析,可以计算.①基本电磁场数值解和开边界问题,近远场辐射问题;②端口特征阻抗和传输常数;③S 参数和相应端口阻抗的归一化S 参数;④结构的本征模或谐振解.依靠其对电磁场精确分析的性能,使用户能够方便快速地建立产品虚拟样机,以便在物理样机制造之前,准确有效地把握产品特性,被广泛应用于射频和微波器件、天线和馈源、高速IC 芯片等产品设计中.H FSS 有本征模解(Eigenmode Solution )和激励解(Driven Solution )两种求解方式.选择Eigen 2mode Solution 用于计算某一结构的谐振频率以及谐振频率点的场值和腔的空载Q0值.选择Driven Solution用于计算无源高频结构的S参数和特性端口阻抗、传播常数等.本课题的研究中,将用到本征模解求解单同轴腔特性和腔间耦合系数;激励解求解有载品质因数Q L值和滤波器响应特性.2　同轴腔滤波器工作原理及设计2.1　工作原理同轴腔滤波器主要用于米波、分米波段.传输TEM模,无色散、场结构简单稳定、空载品质因数高[1].其基本结构由谐振腔、腔间耦合、输入输出激励组成,如图1所示即为一个三腔同轴滤波器.输入信号通过闭合圆环耦合到谐振腔中产生谐振,能量在谐振腔之间由耦合孔进行逐级耦合,再经图1　三腔同轴滤波器结构模型(a=3.25mm,b=9mm,l=29mm,l1=l2=14mm)过输出端的闭合圆环耦合输出.各腔均工作在同一谐振频率附近,只有该谐振频率附近的电磁波有效传输,形成一带通滤波器.2.2　集总参数网络设计下面以S波段滤波器设计为例,主要技术指标见表1.表1　滤波器技术指标技术参数工作频率f0插入损耗L A带宽(4f3dB)通带波动L Ar阻带抑制L As(f0±15M Hz)输入输出阻抗Z o指标要求2.0～2.15GHz≤2dB≥8M Hz≤±0.3dB≥25dB50Ω 利用网络综合法[2],选取切比雪夫函数作为逼近函数,查表或计算[3]确定滤波器阶数n=3,对应的低通原型参数:g0=g4=1,g1=g3=1.0316,g2=1.1474,由此得到腔间耦合系数K ij和外部品质因数Q L.K ij=bwg i・g j=0.0036(i=1,j=2;i=2,j=3)(1)Q L=g1bw=266.3(2)2.3　微波结构设计2.3.1　同轴腔为减小体积和便于安装,本滤波器采用内圆外方的1/4λ缩短电容同轴腔结构.依据谐振腔结构尺寸参数选取三个原则[1]:①避免高次模,(a+b)≤λmin/π;②满足功率容量,b/a=1.65时功率容量最大;③损耗要小,b/a=3.6时Q0值最高,损耗最小.b/a一般选择在2.0～3.6之间.在此选取内导体半径a=3.25mm,外导体内半径b=9mm.内导体长度l、调谐螺钉最大调谐距离t的设计既要考虑能够满足所需的调谐范围,同时还要考虑到内导体缩短会降低Q0值[4]的因素,一般选择内导体长度为1/4λ的65%以上,在此选取l=29mm,t=3mm.谐振腔的调谐范围将通过HFSS进行仿真验算.2.3.2　耦合考虑到本滤波器属于窄带滤波器,腔间耦合[5]采用圆孔实现,输入输出耦合采用闭合半圆环实现.耦合圆孔、半圆环需要确定的参数是中心位置和半径大小.滤波器带宽基本上由级间耦合决定.设计一个在某个频率范围内可调谐的滤波器时,若要保持固定的带宽,则必须控制带宽对频率的敏感性,即要保持d(Δf)/d f=0.Cohn[6]研究得出,当耦合孔中心离腔短路端距离l1在中心频率电长度36°附近时,耦合带宽最大且随频率变化缓慢.则取l1=14mm.半圆环的几何位置通常与耦合孔保持一致,所以也取l2=14mm.关于耦合孔径的大小,下面通过HFSS仿真腔间耦合系数K ij和外部品质因数Q L获取.3　HFSS仿真分析3.1　单谐振腔仿真根据选定的结构尺寸(a=3.25mm,b=9mm, l=29mm),在H FSS中对单谐振腔建模(图2),不需要加载激励,进行Eigenmode分析,获取在不同间距t的加载电容下对应的谐振频率.仿真结果(图3)得出,当t在0.25～3mm之间调整,对应谐振频率范围在1619～2171M Hz之间变化,可以满足要求.图2　单谐振腔模型　图3　谐振频率与加载电容关系3.2　腔间耦合系数K ij仿真腔间耦合的电性能用耦合系数K ij表示.当两134第2期刘鹏宇,张玉虎等:基于H FSS设计同轴腔滤波器个相邻的谐振腔耦合在一起、并且对源和负载具有非常小的耦合时,K ij 与相邻腔谐振频率f 1、f 2存在如下关系[7]:K 12=2(f 2-f 1)/(f 2+f 1)(3)因此,对两个相邻谐振腔在不接源和负载(图4)情况下进行Eigenmode 分析(modes =2),得到在不同圆孔半径下对应的谐振频率f 1、f 2,从而绘制出对应的腔间耦合系数曲线(图5).结果表明耦合孔越大,耦合越强.图4　腔间耦合系数仿真模型图5　耦合圆孔与耦合系数关系3.3　有载品质因数Q L 仿真当单个谐振腔耦合源和负载时,有载品质因数Q L 与谐振频率f o 及3dB 带宽Δf 3dB 存在如下关系[7]:Q L =f o /Δf 3dB(4)建立模型对单谐振腔加载源和负载(图6),进图6　有载品质因数　图7　耦合圆环与有载品质仿真模型因数关系行Driven Terminal 分析,得到在不同耦合圆环半径下对应的有载品质因数Q L 曲线(图7).耦合环越大,耦合越强,Q L 值越低. 根据公式(1)、(2)中计算结果,对照以上仿真分析图表,即可选取适当的结构参数,在H FSS 中完成整个滤波器的建模(图1),经过进一步优化,获取理想的特性曲线,确定最终的结构尺寸:r _apert ure =3.18mm ,r _loop =2.6mm .4　实测结果与分析综合上述设计及优化结果,并考虑到为实物调试时留有一定的调整余量,耦合孔和耦合环半径均取的略小一些,确定最终的加工尺寸见表2.表2　同轴腔滤波器结构加工尺寸结构参数a bltl 1l 2r _loop r _aperture尺寸/mm 3.25929314142.53 按照表2结构尺寸机械加工,进行适当的谐振频率和耦合调整,获得了满意的特性曲线(图8),达到技术指标要求(表3).结果表明,插入损耗、带外抑制实测结果比与仿真结果要差一些.这是可以理解的,因为HFSS 仿真是在理想边界条件下进行的,而滤波器实物是由三个单谐振腔和输入输出端口组合在一起的,,还有腔体内部镀银表面不光滑,这些都会引入损耗[8],导致Q 0值降低,使得插损、带外抑制指标略有变差.图8　实测(粗线)与仿真(细线)滤波器响应表3　滤波器测试数据技术参数工作频率f 0插入损耗L A带宽(Δf 3dB )通带波动L Ar 阻带抑制L A s(f 0±15M Hz )驻波比指标要求2.065GHz 1.75dB8.5M Hz 0.15dB33.6dB1.345　结束语本文利用ANSO F T HFSS 仿真软件对同轴腔滤波器中的谐振腔、腔间耦合及输入输出激励进行了优化设计,确定了滤波器实际结构尺寸,测试结果与仿真一致.该方法可以有效并准确地替代传统试验方法,也可以应用在其它的微波滤波器设计中.参考文献:[1]　廖承恩,陈达章.微波技术基础[M].北京:国防工业出版社,1979.[2]　甘本,吴万春.现代微波滤波器的结构与设计[M].北京:科学技术出版社,1973.[3]　Hong Jia 2Sheng.Microstrip Filters for RF/Microwave Applications ,ncaster C opyrightc 2001John W illy &S ons ,Inc.pp.29261.[4]　K urzrok R.M.Design of C omb 2Line Band 2Pass Filters (C orrespon 2dence )[J ].T ransactions on Microwave Theory and T echniques ,Jul.1966,T 2MTT 214(7):3512353.[5]　姚毅,黄尚锐.调谐滤波器的腔间耦合结构研究[J ].微波学报,1994(1):16222.[6]　K urzrok R M.Design of Interstate C oupling Apertures for Narrow 2Band T unable C oaxial F ilters[J ].(C orrespondence )IRE T rans.on M i 2crowave ’Theory and T echniques ,March ,1961,MTT 210:1432144.[7]　Randall W.Rhea ,HF Filter Design and Computer Simulation[M ].Mc Graw 2Hill ,Inc.,1995.[8]　高葆新.波导带通滤波器的设计[J ].国外电子测量技术,2001(1):34237.234电　子　器　件第30卷。

HFSS和serenade滤波器设计

Efficient Design of Chebychev Band-Pass Filters with Ansoft HFSS and SerenadeTutorialDr. B. Mayer and Dr. M.H. VogelAnsoft CorporationContentsAbstract 31. Introduction 32. Circuit Representation of the Filter 53. Relationships between Circuit Components and Physical Dimensions in the Microwave Filter 104. Initial Filter Design in HFSS 145. Curve Fitting in Serenade 166. Corrected Filter Design in HFSS 207. Additional Information from the 3D Field Solver 21 7a. Effects of Internal Losses 21 7b. Maximum power-handling capability 23 7c. Mechanical Tolerances 24 References 25 Appendix A Derivation of the Circuit 26 Appendix B The Physical Meanings of K and Q 43AbstractAn efficient method is presented to design coaxial Chebychev band-pass filters. The method involves a 3D full-wave field solver, Ansoft HFSS, teaming up with a circuit simulator, Serenade. The authors show how for a practical case, a 7-pole band pass filter with a ripple of only 0.1 dB, an accurate design is obtained in a matter of days, as opposed to weeks for traditional methods.The method described is also applicable to even more challenging designs of elliptic filters and phase equalizers realized in dielectric, waveguide or coaxial technology.1 IntroductionIn this paper, we will describe an efficient method to design a filter. The method involves a 3D full-wave field solver teaming up with a circuit simulator. The basic idea has been explored by others [1] but a different circuit was used in the circuit simulator. We will explain our procedure by presenting in detail how we design a Chebychev band pass filter with the following specifications:Center frequency 400 MHzRipple bandwidth 15 MHzRipple 0.1 dBOut-of-band rejection 24 dB at 390 MHz and at 410 MHzIn order to achieve the out-of-band rejection, we will need seven poles.The desired filter characteristic is shown in Fig. 1.Fig. 1 Desired filter characteristicAs the basic geometry for this filter we have chosen a cavity with seven coaxial resonators, as shown in Fig. 2. In the figure, the “buckets” have been drawn as wire frames for clarity, to show that the cylinders don‟t extend all the wa y to the bottom.Fig. 2 Basic filter geometryThis geometry is symmetrical with respect to the central cylinder. In this kind of filter, the walls of the cavity, the long cylinders, the buckets under the cylinders and the disk-shaped objects near the first and last cylinder are all made of metal. The long cylinders are connected to the top of the cavity; the buckets are connected to the bottom of the cavity. Cylinders and buckets don‟t touch. The disk-shaped objects near the first and last resonator are connected to the input and output transmission lines and provide the necessary coupling to the source and the load. We will call these objects antennas in this document. They are near the first and last cylinders, but never touch them. Each cylinder-bucket combination is a resonating structure. At this stage, without restricting ourselves, we can choose many dimensions in the filter relatively freely. We make the following choices:Cavity dimensions 280 x 30 x 120 mmResonator diameter 10 mmBuckets‟ inner diameter12 mmBuckets‟ outer diameter 16 mmBuckets‟ height15 mmAntennas‟ diameter26 mmAntennas‟ thickness 6 mmSix dimensions remain, and these six will be crucial in obtaining the desired filter characteristic:The length of the first and last resonating cylinder (both have equal length)The length of the five interior cylinders (all five have equal length)The distance between an antenna and its nearest cylinderThree distances between neighboring cylinders (remember the filter is symmetric) With traditional filter design methods, obtaining the correct dimensions is a time-consuming task that commonly takes several weeks. Filter design with a circuit simulator, on the other hand, is relatively straightforward. Filter theory provides the values for the lumped inductors and capacitors that are needed to obtain the desired filter characteristic. First, we will show how to design a circuit that not only has the desired filter characteristic, but also lends itself to implementation with microwave components. In such a circuit, we use series L and C for each resonator, i.e. the cylinder-and-bucket combinations, and impedance invertors to represent the distances between adjacent resonators. Second, we will show how one can determine relationships between components in the circuit and dimensions in the physical filter. Third, we will present an iterative procedure between the electromagnetic field solver and the circuit simulator to optimize the design. The procedure converges very quickly.2 Circuit Representation of the FilterIn order to design an order-seven band-pass filter around 400 MHz with a 0.1 dB ripple, filter theory tells us to start with an order-seven low-pass filter, normalized to 1 radian/s. The normalized filter is to have a 0.1 dB ripple, like the desired band pass filter. The source and load impedances of the normalized low pass filter are normalized to 1 Ohm. This circuit is shown in Fig. 3 and its characteristic in Fig. 4. Filter theory provides us with the values for the inductors and the capacitors, denoted by g1 through g7 in the figure. These values are in our caseg1=g7=1.1812 Hg3=g5=2.0967 Hg2=g6=1.4228 Fg4=1.5734 F.Fig. 3 Normalized low-pass filter circuit, starting point for design procedureFig. 4 Filter characteristic for the normalized low-pass filter in Fig. 3The step-by-step procedure from this normalized low-pass filter circuit to the final band-pass filter circuit is presented in detail in Appendix A. Here, we show an outline of the major steps.An important step is the replacement of shunt capacitors by series inductors and impedance inverters. Basically, an impedance inverter transforms impedances in the same way as a quarter-wave-length transmission line, but independent of frequency. The resulting circuit is shown in Fig. 5. This is still a normalized low-pass filter with the same characteristic as the circuit in Fig. 3. The reason for this change is that at microwave frequencies it is often impossible to realize the ladder circuit consisting of series inductors and shunt capacitors. Depending on the basic structure either series elements or shunt elements are easily realizable but often not both in the same structure. Taking advantage of impedance inverters, it is possible to transform shunt capacitors into series inductors. In the physical filter these impedance inverters will be realized by couplings between the coaxial resonators.Fig. 5 Normalized low-pass filter without shunt capacitorsFollowing a standard procedure, we take the following steps to derive the desired band-pass filter model:(1)De-normalize the low-pass cut-off angular frequency from 1 rad/s to bw rad/s.(2)Transform the low-pass filter to a band-pass filter with a relative bandwidth ofbw and a center angular frequency of 1 rad/s by inserting a 1 F capacitor inseries with every 1 H inductor.(3) De-normalize the center frequency to 400 MHz by choosingL=1/(2×π×4E8) H and C = 1/(2×π×4E8) F.(4) De-normalize the port impedances from 1 Ohm to the usual 50 Ohm byintroducing impedance inverters at the input and output with coupling coefficients of √50.(5) Introduce finite quality factors to the individual resonators by adding a seriesresistor to each resonator.(6) Introduce individual resonant frequencies to the first and last resonators to beable to be able to take the frequency shift due to the coupling antennas intoaccount.(7) Add a homogeneous transmission line of length ZUL between filterinput/output and port 1 / port 2 to be able to adjust the phase due to the connectors. This gives us the filter shown in Fig. 6. The procedure outlined above is presented in more detail in Appendix A.Fig. 6 Final filter circuit, representing the desired band pass filterIn this circuit, every LC pair resonates at 400 MHz. Further K12, K23, K34 and Q L have been defined asK ij= bwg i g j(1)andQ= g1bw(2)where bw is the relative bandwidth and g i is the i th g value from filter theory.Notice that, since the g values are known from filter theory, we still know the values of the all the components in the circuit, even through the components have changed considerably in the process.Filter theory [2] tells us that K i,i+1 and Q L have important physical meanings. K i,i+1 is known as the coupling constant between adjacent resonators. If we have just two resonators in the cavity, with a very light coupling to the source and the load, then the relation between coupling constant K12 and resonant frequencies f1 and f2 is given byK12 = 2(f2-f1) / (f2+f1) . (3)Q L is known as the loaded Q of the circuit. If we have just one resonator in the cavity, coupled to source and load, the relation between Q L , resonant frequency f R and 3-dB band width BW3dB is given byQ L = f R / BW3dB(4)In the next section, we will link the components of this circuit to dimensions in the physical geometry of the filter.3 Relationships Between Circuit Components and Physical Dimensions in the Microwave FilterAs explained in the previous section, every LC pair resonates at 400 MHz. In the microwave filter, we must choose the length of each resonator such that it resonates at 400 MHz. That will determine the length of each of them.Further, K i,i+1 (i=1,2,3) are the coupling coefficients between adjacent resonators. Therefore, these three coefficients are related to the distances between adjacent resonators.Finally, Q L is the loaded Q of the circuit. Therefore, in an otherwise lossless circuit, it is directly related to the distance between the first or final resonator and the antenna that couples it to the source or the load.3a Relation Between Resonator Spacing and Coupling CoefficientThe model in Ansoft HFSS that was used to determine the coupling coefficient K as a function of resonator spacing is shown in Fig. 7. Two resonators have been placed in a closed metal cavity. This cavity has the same height and the same front-to-back depth as the cavity to be used in the real filter. The left-to-right width has been chosen large enough to make its influence on the results negligible. There are no transmission lines nor ports for signal input and output, since the resonances of this structure are to be determined through an eigenmode simulation. In the Setup Materials menu, resonators are modeled as perfect conductors; the cavity is filled with air. Further, symmetry has been exploited through the use of a Perfect-H boundary condition. As can be seen in the figure, this cuts both the resonators and the cavity in half.Fig. 7 Model used to determine the coupling coefficient KBy embedding this HFSS project in Optimetrics, dimensions can be varied easily. First, the length of the cylinders was adjusted such that the resonances are centered at 400 MHz. Then, the distance between the resonators was varied, and for each distance the eigenmode solver in HFSS computed the two eigen frequencies and obtained K. In order to get very accurate results, more accurate than necessary, twelve adaptive passes were run in each simulation, resulting in models with 63,000 tetrahedra. Total run time foreach point was 38 minutes on a 1.2 GHz PC. The simulation of the lossless structure required 557 MB of RAM. The relation between the resonator spacing and K is shown graphically in Fig. 8.Fig. 8 Relation between resonator spacing and coupling coefficient K With this graph, for any coupling coefficient required by filter theory, the spacing to be applied between resonators in the physical model can be readily determined.3b Relation Between Antenna Distance and Loaded Q The model in Ansoft HFSS that was used to determine the loaded Q as a function of antenna spacing is shown in Fig. 9. An antenna-resonator combination has been placed in a closed metal cavity. The 50-Ω transmission line is present, but in order to perform an eigenmode analysis, it has been terminated by a Perfectly Matched Layer (PML) of absorbing material. This was done by replacing the final 20 mm of dielectric in the coaxial cable by PML material. A macro, named pmlmatsetup, in the Materials-Setup menu supplies the material parameters. This construction will give use the same resonant frequency and loaded Q as the corresponding structure with a real 50-Ω load would. The cavity has the same height and the same front-to-back depth as the cavity to be used in the real filter. Again, the left-to-right width has been chosen large enough to make its influence on the results negligible, and symmetry has been exploited through the use of a Perfect-H boundary condition.Fig. 9 Model used to determine the loaded QThis HFSS project has been embedded in Optimetrics. The antenna distance and the cylinder length were varied simultaneously, since both influence the resonant frequency and the loaded Q. As an example of the results, the relation between antenna spacing and loaded Q is shown graphically for a constant cylinder length of 113.4 mm. In order to get very accurate results, maybe a little more accurate than strictly necessary, fifteen adaptive passes were run for each point. This results in simulations with 50,000 tetrahedra, requiring 830 MB of RAM. Total run time per point was 50 minutes on a 1.2 GHz PC.Fig. 10 Relation between antenna spacing and loaded Q at a resonator length of 113.4 mmWith results like these, for any loaded Q and resonant frequency required by filter theory, the antenna spacing and cylinder length to be applied in the physical model can be readily determined.4 Initial Filter Design in HFSSNow that we have the circuit and we know the relations between circuit components and physical dimensions, we can construct the filter in the field solver, Ansoft HFSS. Filter theory tells us we need to achieve the following parameters: Resonant frequency of the outermost resonators f R1= 400 MHzResonant frequency of the inner resonators f R2=400 MHzLoaded QCoupling coefficients K12=0.02893 , K23=0.02171 , K34=0.02065The calibration projects above tell us that the dimensions of the filter, as shown in Fig. 2, are to beLength of the two outermost resonators = 113.399 mmLength of the five inner resonators = 114.69 mmAntenna distance = 1.879 mmDistances between resonators are 25.513 mm , 28.291 mm , 28.767 mmThis filter has been modeled and simulated in Ansoft HFSS. The model is shownin Fig. 11.Fig. 11 Initial design in HFSSNotice that only half the geometry is actually simulated. Symmetry has been exploited through the use of a Perfect-H boundary condition. Further, all materials and boundaries in the model are lossless for now. This requires less RAM and less CPU time. The resulting filter characteristic is shown in Fig. 12. Notice that the center frequency and the ripple bandwidth are almost perfect. We see the correct number of ripples, but theripple is 0.3 dB rather than 0.1 dB.Fig. 12 S21 results for the initial design in HFSS5 Curve Fitting in SerenadeThe HFSS results, shown in the previous section, have been exported to Serenade, the circuit simulator. This was done through Post Process / Matrix Data / File / Export / Touchstone. In Serenade, we can determine through curve fitting what the actual parameters of this initial design are. This curve fitting is done through the Serenade setup shown in Fig. 13. All the variables are defined in the top level schematic. Circuit model and HFSS results are defined via the sub circuits denoted as MODEL and MEASU, respectively. To utilize all the available information for the optimization process, the optimization goal is to end up in a complex S-Matrix identical for the model and the S-matrix resulting from the HFSS simulation. This is defined in Serenade via an OPT block and the goal definition S=MEASU in the sub circuit defining the circuit model as shown in Fig. 14. To match the phases of the S parameters of the Serenade and HFSS simulations, homogeneous transmission lines of length ZUL are attached to ports 1 and 2 in the Serenade model. Optimization is done by starting with the optimum filter parameters given in Fig. 6 and performing 1500 iterations with the random optimizer. The solution was found without any manual interaction.It took 35 minutes on a PC with a clock speed of 400 MHz. Figs. 15 through 17 show curve fitting results. Notice, in Fig. 15, that there is still a few hundredths of a dB difference between the HFSS results and the best fit in Serenade. This indicates that thisdesign method is accurate to a few hundredths of a dB.Fig. 13 Serenade setup used for curve fitting: top level schematicFig. 14 Serenade setup used for curve fitting: Model definition as sub circuitFig. 15 Result of curve fitting, magnitudeFig. 16 Result of curve fitting, phaseFig. 17 Results of curve fitting, complex S11 Blue and green lines are S_11 and S_22 from HFSS, which have the same magnitude but slightly differentphases; red line is the best fit.The result of the curve fitting procedure is as follows: we have built a filter with Resonant frequency of the outermost resonators f R1= 400.058 MHzResonant frequency of the inner resonators f R2=399.926 MHzLoaded Q Q L=30.368Coupling coefficients K12=0.02825 , K23=0.02173 , K34=0.02068Notice that the largest discrepancies occur in K12 and Q L. Apparently, the calibration project that determines the coupling coefficient by simulating two identical resonators is not quite representative of the two outermost pairs of resonators, where one resonator is coupled through an antenna to the source or the load. Also, the calibration project that determines the loaded Q by simulating one resonator-antenna combination is not perfectly representative of the real situation where this resonator is coupled to a neighboring one.Nevertheless, the calibration projects tell us how much correction is needed to achieve the desired characteristic. For example, noticing that Q L is too low by a certain amount, we will aim for a Q L that is higher by this amount the second time. Caution is needed when adjusting the antenna distance, since that also changes f R1. We have to change antenna distance and resonator length simultaneously, and aim for the correct Q Land f R1.Keeping this in mind, with the aid of the calibration projects we find that the dimensions of the filter are to beLength of the two outermost resonators = 113.44 mmLength of the five inner resonators = 114.684 mmAntenna distance = 1.928 mmDistances between resonators are 25.286 mm , 28.3 mm , 28.78 mm Hence, the dimensions that undergo the largest changes are the antenna distance and the distance between the first and second resonator.6 Corrected Filter Design in HFSSThe corrected filter was modeled and simulated in Ansoft HFSS. The resulting characteristic and the corresponding Smith chart are shown in Figs. 18a and 18b. Note that the ripple, which was 0.3 dB in the initial design, is better than 0.13 dB now. The target is 0.1 dB.Fig. 18a S21 results in HFSS for the corrected designFig. 18b Smith chart in HFSS for the corrected designIn order to obtain this result, the mesh was refined adaptively until it had 180,000 tetrahedra. With a mesh that size, the calculation of each frequency point required 1.28 GB of RAM and 9.5 minutes real time on a 1.2 GHz PC with one processor. Seventeen frequency points were needed for an interpolating frequency sweep, bringing the total time needed for the sweep to two hours and forty minutes. An identical model with only 119,000 tetrahedra (see below) provided results within a few hundredths of a dB in the pass band and saved almost half the time.7 Additional Information from the 3D Field Solver7a Effects of Internal LossesAll simulations thus far have been performed with lossless filters. A simulation without loss results in computations with real numbers only, as opposed to computations with complex numbers. This reduces the RAM requirement and the CPU time significantly. Once the design has been finalized, however, one can easily change the material parameters and boundary conditions to go from perfectly-conducting metals to lossy metals like copper or silver. The software enables you to select materials from a database or specify the conductivity. A plot comparing a lossless and a silver filter is shown in Fig. 19. Notice that, due to the seven consecutive resonances, even with a very good conductor like silver the insertion loss will be between 0.5 and 1 dB.Fig. 19 Comparison lossless filter and silver-plated filterAlso note that the center frequency of the silver filter is slightly lower than the center frequency of the perfect filter. A careful inspection of the data shows that this shift is between 0.07 and 0.08 MHz. A model with just one resonator shows the same shift. Further investigation reveals that this shift is due to the imaginary part of the surface impedance of the silver. According to electromagnetic theory, the conductivity of the silver translates into an equivalent surface impedance, provided that the metal thickness is much larger than the skin depth. This surface impedance has a real and an imaginary part, which are both equal to √ (πfμ0μR/σ), where f is the frequency, μ0μR is the permeability of the material, and σ is the conductivity of the material. In the case of silver at 400 MHz the surface impedance is Z surface= 5(1+j) mΩ/square. A simulation in HFSS with Z surface = 5 mΩ/sq uare shows no frequency shift at all relative to the perfect conductor case, while a simulation with Z surface= 5j mΩ/square shows the same shift as in Fig. 19.Curve fitting with Serenade shows that replacing perfect conductors by silver in HFSS is equivalent to introducing an unloaded Q of 2,800 in each resonator in Serenade. According to filter theory, the introduction of an unloaded Q shifts the resonant frequency downward by f r/(2Q), which in this case equals 0.07 MHz. Hence, HFSS has predicted this frequency shift very accurately.In order to account for this shift, designers should first determine the magnitude of the shift with an HFSS simulation involving just one resonator. Then, they should design a perfectly lossless filter around a frequency that is higher by this amount. The center frequency of the filter with internal losses will thus come out just right.The computer requirements were as follows. These computations have been performed with a model with 119,000 tetrahedra. In the lossless case, this took 810 MB RAM, 207 MB disk, and 86 minutes real time on a PC with a clock speed of 1.2 GHz and one processor. In the lossy case, it took 1,332 MB RAM, 1,600 MB disk and 396 minutes real time. The large time difference is due to the change from a real to a complex solver and to the time needed for disk access. The disk access in this case is …spill logic‟, which is a deliberate process, performed under the software‟s control. It is not to be confused with the very inefficient …swapping‟ which is done by the operating system when a process is too large for the available RAM.7b Maximum Power-Handling CapabilityIt is important to know how much power the filter can handle. The maximum power handling capability can be obtained easily with the help of a field plot. Fig. 20 shows a close up of the fields around a resonator in the region where they are strongest.Fig. 20 Fields around a resonatorThe HFSS 3D Fields Post Processor tells us that, with 1 W input power, the electric field strength between the cylinder and the bucket is 105 kV/m. You can change the input power in the post processor (Data/Edit Sources). The filter would cease to operate when the fields are strong enough to cause arcing in the air. This phenomenon occurs at 3 MV/m, although, with a wide safety margin, 1 MV/m is commonly used as the maximum acceptable field strength. Therefore, the fields can be allowed to be 9.5 times as strong as they are now, which implies that the maximum power handling capability is 9.5×9.5 = 90 W.7c Mechanical TolerancesOnce the dimensions are known that provide a filter with the desired specifications, it is important to establish mechanical tolerances. With the HFSS model fully parameterized in Optimetrics, it is an easy task to explore the effects of smalldimensional changes on the filter characteristic. An example is shown in Fig. 21. There,the distance between two resonators was made 0.08 mm larger and 0.08 mm smaller. The original characteristic and the two modified ones are shown. In this case, a manufacturing inaccuracy of 0.08 mm in the distance between two resonators results in a change of up to 0.05 dB in the filter characteristic. This way, mechanical tolerances, depending on the accuracy requirements of the filter characteristic, can be specified.Fig. 21 Example of the effects of manufacturing tolerancesReferences[1] Daniel G. Swanson and Robert J. Wenzel, “Fast Analysis and Optimization of Combline Filters Using FEM”, presented at the IEEE MTT Soc iety 2001 International Microwave Symposium, May 2001.[2] Randall W. Rhea, “HF Filter Design and Computer Simulation”McGraw-Hill, Inc., 1995ISBN 0-07-052055-0Appendix A Derivation of the CircuitIn this appendix, the well known low-pass prototype method for filter designs is repeated. Only the necessary facts for the present example are given. For more details a standard book on filter designs should be used [A1]. The starting point of this method is the low-pass prototype as shown in Figs. A1 and A2. These filters are normalized to a cut-off angular frequency of 1 rad/s and a generator impedance of 1 Ohm. In the case of a Chebychev filter some care has to be taken regarding the order of the filter. For an odd number of elements the load impedance is also 1 Ohm. For an even number of elements, however, the load impedance depends on the order of the filter and the ripple. Therefore, the two cases are treated separately.g = 1g N-1g Ng 3g = 1 N+1Fig. A1Low-pass prototype for the case N=oddg = 1g Ng 3g N-1g = 1N+1Fig. A2Low-pass prototype for the case N=evenIn the past, the prototype values g i (i = 1 … N+1) were read from tables, but nowadays it is more convenient to use a filter design program. Chebychev filters aredefined by the filter order N and the in-band ripple. Closed-form expressions exist for the g-values. An example is given in Fig. A3. This MathCAD program is valid for even as well as odd order filters for any ripple value. Essentially the g-values are defined by a recursive relation. Only for the last value a special treatment for the even and odd order case is necessary. This is considered by an if –statement with the mod –function ascondition. Exact definitions of these functions are given in the MathCAD handbook [A3]. The response of this filter is shown in Fig. A4.N 6:=ripple 0.1:=eps 10ripple 101-:=K i ()4sin 2i 1-()⋅1-2N⋅π⋅⎡⎢⎣⎤⎥⎦sin 2i 1-()⋅1+2N⋅π⋅⎡⎢⎣⎤⎥⎦⋅sinh asinh 1eps ⎛ ⎝⎫⎪⎭N ⎛ ⎝⎫⎪⎪⎭⎛ ⎝⎫⎪⎪⎭2sin i 1-()π⋅N ⎡⎢⎣⎤⎦⎡⎢⎣⎤⎥⎦2+⋅:=G i ()if i 12sin π2N ⋅⎛⎝⎫⎪⎭sinh 1N asinh 1eps ⎛ ⎝⎫⎪⎭⋅⎛ ⎝⎫⎪⎭⋅,K i (),⎛⎝⎫⎪⎪⎪⎭:=g01:=gi i ()if i1G i (),G i ()gi i 1-(),⎛ ⎝⎫⎪⎭:=gnplus1if mod N 2,()01eps 1eps2++()2,1,⎡⎢⎢⎣⎤⎥⎥⎦:=g0nplus1i ()if i 0g0,gnplus1,():=g i ()if i N 1+()i 0()+g0nplus1i (),gi i (),[]:=j 0N 1+..:=g j ()=j =Fig. A3MathCAD program to derive the g-values for Chebychev filter responses. Example: N = 6, ripple = 0.1 dB.。

HFSS,天线,滤波器,学习记录

1）根据腔数和整体尺寸确定大致腔体尺寸2）单腔仿真，确定谐振杆和调谐杆的半径r1,r2，3）根据元件值计算理论耦合系数,然后做双腔仿真固定2）中得到的参数不变，对两腔间距W作参数扫描调整，输出K-W曲线，使得W满足K要求4）计算理论需要的Qe，再做单端口Qe仿真，调整连接引线接在谐振杆上的位置T直至符合要求5）根据以上得到的数据整体仿真6）得到的曲线很不理想，再调整获得合适的中心频率,带宽，但是通带衰减过大的问题始终无法解决随后对T调整，发现T越大反而通带衰减越小，而以前看到资料上说，中心抽头接入的位置应尽量靠近谐振杆的短路端，我现在选T=1.8mm，通带衰减最好才-13分你要用软件仿真腔体滤波器得到一个理想的结果是比较困难的,一般只要仿真出来有波形的样子,并且保证中心频率和带宽满足要求就可以加工了.一般都是能实调出来的.如果你非要在软件中调个好的波形出来,那就要不断的调整耦合以及有载Q值.其中影响最大的是K12和有载Q值,你调试的主要精力需要放在改变一二腔的距离,抽头高度,以及第一腔的加载螺钉上.过程是比较烦琐的,祝你早日成功!很多问题可以直接再论坛里搜索,比百度,好对哦了1、看下频率（因为这是后面HFSS或者CST仿真要用的单腔频率）2、看带宽和近端抑制点以及插损（这个可以用相关软件仿比如MA TLAB或者COUPLEFILA 仿真下需要几阶，几个传输零点以及交叉耦合的方式。

一般阶数越少，插损越好，抑制越插）3、再根据带宽所需要的耦合系数用HFSS或者CST仿真下，看谐振杆的间距或者耦合窗口应该定多大。

4、开始排腔，以及投入初样（一般开始做初样前还可以拿Desinger把电路仿真下，因为Desinger里面可以改变每个腔的Q值等，进行验证，看设计是否有明显的错误）5、调试，这个其实就是看个人的水平了，多动手多思考第四步排完腔一般我会用HFSS或者CST仿下Q值，看能否达到第二步用解析软件计算时预设的Q值，如果达不到就要重新考虑方案了看懂规范书抑制损耗回波功率互调温补要了解，先看通带曲线确定节数几传输零点个数零点实现形式和对应位置以及Q多少满足综合指标，仿单腔确定频率和Q值，观察几个元件间距（影响功率因素），后布局几点重要建议：布局的空间合理性和结构紧凑，生产可操作性，各个通道（单腔大小）分配均匀，功率要求尽量内部各个间距加大，互调高要对连接器表面处理材料光洁度做要求温补要考虑材料的不同环境下发生形变对指标的影响另外选用几种形式：交指梳状平行耦合，这就要看个人喜好了对于窄带滤波器来说，仿真频率必须放在中心频率上，收敛：maximum number 设置个几十，maximum delta s:0.02.看过一些资料，对耦合系数和端口外部Q值的计算都已了解，现在在仿真上有些问题，向大家请教一下第一个就是耦合会使谐振频率下降，所以仿真时会让单腔的谐振频率稍微高一些，那么一般应该高多少呢？第二个就是比如1、2两个腔的耦合窗尺寸已经调好了，耦合系数K12在中心频率和理论值差不多，接着在仿真2、3两个腔的时候，调节2、3腔之间耦合窗口大小使耦合系数K23与理论差不多的时候，谐振频率已经偏离了中心频率，这种情况接着怎么处理呢？需要调节什么参数呢？第三个就是在HFSS里用本征模仿真外部Q值的时候发现Q值与理论值一样的时候，此时的谐振频率与中心频率不一致，这种情况该如何处理呢？一，一般缩个15%~20%,原则上你能调回来就好二，改变谐振杆高度调频率啊，尽量在中心频率下算窗尺寸三，还是改变谐振杆的高度吧正耦合系数（磁耦合）可以很简单的通过腔与腔之间各种形状的开孔实现，《现代微波滤波器的结构与设计》里面有对应的相关公式。

基于HFSS的带通滤波器设计论文

本科毕业设计（论文）题目基于HFSS的带通滤波器设计学院物理与电子工程学院年级08 专业电子信息工程班级 2 学号*********学生姓名刘建指导教师施阳职称讲师论文提交日期基于HFSS的带通滤波器设计摘要本文介绍了矩形杆交指型带通滤波器的原理和设计方法,根据交指滤波器的设计理论获得矩形杆的自电容和互电容,利用计算机辅助设计工具结合图表得到滤波器的初始尺寸。

给出了一个中心频率为1.5GHz,带宽为1GHz的矩形杆交指滤波器的设计过程,利用HFSS 软件仿真,提高了设计效率和精度,仿真结果和理论结果吻合良好,证明了设计方法的可行性。

关键词：HFSS软件带通滤波器交指型带通滤波器HFSS-based band-pass filter designAbstractThis paper gives the particular description of a theoretical analysis and practical design for rectangular -rod inter -digitalband-pass filter. First, the self - capacitance and mutual capacitance of the inter-digital band-pass filter is obtained in accordingwith the design theory of the inter-digital band-pass filter. Based on computer aided design, the parameters of filter are given. Filterwith pass-band centered at 1.5GHz and bandwidth is 1GHz, has been designed and simulated by HFSS-software, whichgreatly improves the efficiency and accuracy of the design. The simulation result shows good agreement with theoretical result, whichproves the method is valid.Key words:HFSS-software；band-pass filter;Cross-finger-type band-pass filter目录第一章绪论1.1HFSS软件简介HFSS是ANSOFT公司开发的一个基于物理原型的EDA设计软件.使用HFSS建立结构模型进行3D全波场分析,可以计算.①基本电磁场数值解和开边界问题,近远场辐射问题;②端口特征阻抗和传输常数;③S参数和相应端口阻抗的归一化S参数;④结构的本征模或谐振解.依靠其对电磁场精确分析的性能,使用户能够方便快速地建立产品虚拟样机,以便在物理样机制造之前,准确有效地把握产品特性,被广泛应用于射频和微波器件、天线和馈源、高速IC芯片等产品设计中.HFSS有本征模解( Eigenmode Solution)和激励解(Driven Solution)两种求解方式.选择Eigenmode Solution 用于计算某一结构的谐振频率以谐振频率点的场值和腔的空载Q0值.选择Driven Solution用于计算无源高频结构的S参数和特性端口阻抗、传播常数等。

八腔带通腔体滤波器的设计

作者简介：陈建囡，女，１９８４年生，工程师，研究方向：射频设计与雷达接收；李卉，女，１９８９年生，工程师，研究方向：射频设计与雷
达接收；马雯莹，女，１９８４年生，工程师，研究方向：射频设计与雷达接收；吴梅，女，１９８２年生，工程师，研究方向：射频
设计与雷达接收。
— ４６ —
陈建囡等八腔带通腔体滤波器的设计
根据反射函数和传输函数，运用网络综合的理论
得到输入输出耦合系数Ｒ＝［Ｒ１，Ｒ２］和耦合矩阵Ｍ＝
带宽、腔体数、回波损耗等指标，相关结果便在相应模
块窗口中显示出来。调整滤波器的谐振腔个数，结合
幅频特性窗口中的相关指标情况，便能确定出合适的
腔体滤波器腔数值。［１］如果腔数设置太少，则带外抑
Ｄｅｓｉቤተ መጻሕፍቲ ባይዱｎｏｆａｎｅｉｇｈｔ⁃ｃａｖｉｔｙｂａｎｄ⁃ｐａｓｓｃａｖｉｔｙｆｉｌｔｅｒ
ＣＨＥＮＪｉａｎ⁃ｎａｎ，ＬＩＨｕｉ，ＭＡＷｅｎ⁃ｙｉｎｇ，ＷＵＭｅｉ
（Ｎｏ．７２４ＲｅｓｅａｒｃｈＩｎｓｔｉｔｕｔｅｏｆＣＳＩＣ，Ｎａｎｊｉｎｇ２１１１５３）
Ａｂｓｔｒａｃｔ：Ａｎｅｉｇｈｔ⁃ｃａｖｉｔｙｂａｎｄ⁃ｐａｓｓｃａｖｉｔｙｆｉｌｔｅｒｉｓｄｅｓｉｇｎｅｄ，ａｎｄｔｈｅｔｉｍｅａｎｄｃｏｓｔｏｆｄｅｓｉｇｎｉｎｇｍｉ⁃
ＨＦＳＳ的三维场仿真，能极大减少设计微波腔体滤波器的时间和成本。通过对模型的仿真和优
化，最终设计出来的滤波器实验测试结果与仿真结果基本一致，验证了方法的准确性。
关键词：腔体滤波器；ＨＦＳＳ；八腔；仿真
中图分类号：ＴＮ７１３．５文献标志码：Ａ文章编号：１００９⁃０４０１（２０１９）０４⁃００４６⁃０４

基于HFSS的滤波器设计流程

滤波器设计流程：1.确定设计指标要求2.查阅资料，确定形状3.建模，仿真4.优化结果5.版图，加工，测试本例设计一个带通滤波器，通过微带线结构实现，工作频率覆盖。

选用基板材料为Rogers 4350,其相对介电常数为，厚度为h=0.508mm，金属覆铜厚度h1=0.018mm，表1 模型初始尺寸W1=0.8mm L1=7.2mm h=0.508mmh1=0.018mm S1=0.14mm S2=0.6mm L2=7.1mm W2=1.1mm W0=1.1mm L0=5mm a=6mm b=17mm设计步骤（以为例）一开始（一）建立工程1.在HFSS窗口中，选择菜单File->New2.从Project菜单中，选择Insert HFSS Design（二）设计求解模式1.选择菜单HFSS->Solution Type2.在Solution Type窗口，选择Driven Modal,点击OK二建立3D模型（一）定义单位并输入参数表1.选择菜单Modeler->Units2.设置模型单位：mm，点击OK3.选择菜单栏 HFSS->Design Properties再弹出的窗口中，点ADD添加参量，将上面模型的参数表中的变量全部添加进去，如下图：（二）创建金属板R11.在菜单栏中点击Draw->Box,创建Box12.双击模型窗口左侧的Box1，改名为R1，再点击Material 后面按钮，选择Edit，选择Copper，点击确定。

3.双击左侧R1的子目录Createbox，修改金属板大小及厚度。

Position输入坐标（0mm,0mm,0mm），金属板长L1=7.2mm，宽W1=0.8mm，厚h1=0.018mm。

点击确定。

（三）创建金属板R1_11.在菜单栏中点击Draw->Box,创建Box22.双击模型窗口左侧的Box2，改名为R1_1，再点击Material 后面按钮，选择Edit，选择Copper，点击确定。

HFSS13微带滤波器教程

HFSS13微带滤波器教程HFSS（High-Frequency Structure Simulator）是一种基于有限元法的电磁场仿真软件，广泛应用于高频电磁场分析与设计领域。

本教程将介绍如何使用HFSS13来设计微带滤波器。

第一步：创建新工程在打开HFSS13后，点击"File"，然后选择"New"来创建一个新的工程。

在弹出的对话框中，填写工程名称和保存路径。

接下来，选择“Microstrip”作为工程类型，然后点击“OK”。

第二步：添加设计在创建工程后，点击左侧的“Design”选项卡，在弹出的对话框中选择“Insert”并选择“Design”。

然后在弹出的对话框中，填写设计名称，并选择“SI”（Sheet of Integrated Circuits）作为模板。

点击“OK”来添加设计。

第三步：定义基底材料接下来，点击左侧的“Design”选项卡，在弹出的对话框中选择“Edit”并选择“Mesh Settings”。

在弹出的对话框中，选择“Substrate”选项，并定义基底材料的相关参数，如介电常数、介质损耗等。

点击“OK”来确定。

第四步：创建微带线结构在设计窗口中，点击左侧的“Drawing”工具栏中的“Microstrip”按钮来创建微带线结构。

根据设计要求，在工作区内绘制微带线的几何形状，并指定尺寸参数。

通过右键单击微带线，可以选择“CreateCoplanar Waveguide”来添加同轴线连接。

第五步：定义仿真设置点击左侧的“Analysis”选项卡，在弹出的对话框中选择“Setup”并选择“General”。

然后在弹出的对话框中，填写仿真设置的相关参数，如频率范围、端口设置等。

点击“OK”来确定。

第六步：添加端口激励在设计窗口中，点击左侧的“Excitations”工具栏中的“Waveport”按钮来添加端口激励。

根据设计要求，在工作区内选择微带线的起始和终止点，并指定激励参数。