broadband-noise

合集下载

Ultra-Wideband Low-Noise Amplifier超宽带低噪声放大器

2012 International Conference on Solid State Devices and Materials ScienceUltra-Wideband Low-Noise Amplifier Kaizhuo Lei, Jiao Su, Jintao Shang, Quanshun Cui and Haibo YangCollege of Marine EngineeringNorthwestern Polytechnical UniversityXi’an, Shaanxi Province, China 710072AbstractUltra-Wideband (UWB) Low-Noise Amplifier (LNA) is an essential part of the digital TV and UWB signalprocessor, but what makes it hard to design is the comprehensive consideration of bandwidth, noise and gain controlperformance. A new solution of high performance amplifier with low-noise, UWB and direct current (DC) ispresented (Fig.1), which is composed of a precision pre-amplifier with AD797, a stepped gain controller withVCA810 and a digital potentiometer, an eight-order Bessel low-pass filter with LC network, a zero-drift correctorwith the digital compensation method. The test results (Tab 1-3 & Fig.6) show that the gain of amplifier can beadjusted from 0 to 80dB by step, the fluctuation of the pass band from DC to 10MHz is less than 0.87dB, stop-bandattenuation reaches -42dB/2fc, the equivalent input noise voltage is less than 7.2μVrms. This design successfullysolves some high challenging contradictions, such as ultra-wideband and low-noise, stop-band attenuation andpass-band fluctuation, precise gain control and DC zero-drift correction.©2012 Published by Elsevier B.V. Selection and/or peer-review under responsibility of Garry LeeKeywords :amplifier; ultra-wideband; low-noise; gain control; zero-drift correction1 Introduction1Ultra-wideband (UWB) Low-Noise Amplifier (LNA) is widely used inthe mid-frequency and videoamplifiers. This kind of circuit is not only used to amplify the video signal, impulse signal and RF signalwith the bandwidth ranging from DC to several MHz or even tens of MHz [1], but also widely applied inthe signal processing [2]. In recent years, with the rapid development of ultra-wideband in the covertcommunication [3] and target detection [4], higher requirements for the bandwidth are claimed by theUWB signal, thus the front-end preprocessing circuit of the receiver must be a low-noise amplifier [5][6] with UWB [7]-[9].The performance [10] of the ultra-wideband amplifier directly influences the precision of signaldetection and processing. As a consequence, the design of low-noise, low zero-drift and ultra-widebandbecomes the key point which is of great engineering significance and application value [11]. In other references, the typical gain of UWB LNA was 12-20dB [12] and there was also a contradiction between performance and feasibility. For example, Ref.[13] [14] proposed the amplifier which solved the problem of ultra-wideband and low-noise but it couldn’t avoid zero-drift and high NF.This paper designed and realized a low-noise wideband amplifier made up with the low-noise amplifier, high performance filter network [15], and digital program control circuit for zero-drift correction [16], MCU control system and high precision power supply. Several contradictions such as the ultra-wideband and low-noise, the high stop-band attenuation and low pass-band fluctuation, the high precise gain control and the compensation of DC zero-drift, etc. were successfully solved [17]. The design of our machine got superior parameters and reliable performance together with better promotion value.2 Solution Of Low-Noise And Ultra-widebandThe functional block diagram of the low-noise wideband DC amplifier is shown in Fig. 1. Theamplifier system contains five parts: the primary amplifier, filter network, zero-drift correction circuit,control system and high performance power supply. The primary amplifier consists of low-noise precise pre-amplifier, gain control, mid-amplifier and power driver circuit outputted by the final push-pull. Thelow-noise precise pre-amplifier adopts the ultra low-noise integrated operational chips, realizing the low-noise for the whole system. Voltage gain is adjusted by the MCU. Mid-amplifier consists of the low-noise, high speed integrated amplifier in order to increase the system gain. Final end power driver adopts the dual op-amp consisting of the pull-push output to increase the loading ability of the system. The high performance filter adopts the passive filter proposal to realize the 0~5MHz and 0~10MHz dual channel, eight-order Bessel low-pass filter with the switchable wave band. The zero-drift correction hastwo proposals: analog revised and digital revised, and here we adopt the digital one to increase the correction precision. The control system is to realize the gain and zero-drift digital control with the MCU AT89C52 [18] as the centre. Power supply adopts the mixed regulator, through the decoupling filter, secondary regulator and precise regulator in order to provide the precise low-noise DC power for the whole system.3 Design Of Circuits And Calculation Of Parameters3.1 The design of low-noise and ultra-widebandDecreasing the output noise is the key to the wideband amplifier. By using the Friis Formula we can get the noise coefficients of the cascade amplifier [19]:Where NF1、NF2…NFn are the noise coefficients of each amplifier, and Kpa1、Kpa2…Kpa(n-1) are the gains of each amplifier. From the Friis Formula we can see what affects the cascade amplifier most is the first stage amplifier, so we should try to get an amplifier of smaller noise coefficient and larger gain in the low-noise design.The design chooses the ultra low-noise integrated op-amp AD797 as the pre-amplifier matching the appropriate source impedance. The peripheral devices consist of high performance low-noise metal film resistors and each stage adopts low-noise chips. The LC low-pass filter with bands of 0~5MHz and 0~10MHz is designed. And also the analog and digital grounds are separated in the PCB layout and masking technique is also adopted in the preceding stage in order to decrease the output noise voltage. The low-noise pre-amplifierconsisting of AD797 is shown in Fig. 2.3.2 Program gain controlGain controller is a kind of control method with the amplifier gain changing along with the external control signal. In this system, the program gain control is realized easily by using the external keyboard to set the gain, the voltage gain control amplifier VCA 810 is selected as gain controller.With the control of MCU, the digital potentiometer X9C103 adjust the output voltage ranging between 0~2V, which is added to the VCA810 gain control pin. In this way, we can reach the system with the gain of 0~80dB and the 1dB step adjustable. The principle circuit of gain controller is shown in Fig. 3.3.3 DC zero-drift correctionDC zero-drift is that the operating point of DC amplifier irregularly, slowly and gradually changes.The greater the gain and the more magnification series, the more serious the phenomenon will be, even making the op-amp work badly when the zero-drift reaches the saturation. So a DC zero-drift correction circuit must be designed in order to guarantee the stability of DC amplifier. Through A/D sampling, the DC zero-drift detected in the final stage is sent into the MCU, then we can realize the automatic set of zero by choosing the proper reference voltage and using MCU to control digital potentiometer X9C102 with a compensation voltage adding to the zero regulating end. The zero-drift correction circuit is shown in Fig. 4.3.4 High performance filter networkThe filter is mainly used to reduce the noise, filter band interference and improve system stability. In this design, two low-pass filter pass-bands are 0..5MH and 0..10MHz, with the additional requirements of the pass-band fluctuation less than 1 dB, and stop-band attenuation 40dB/2fC, so precise capacitance and inductance are used to achieve the eight-order passive LC low-pass filter. In order to realize the linear phase, the Bessel filter has to be adopted. As for the complicated calculation and hardship in setting the parameters of LC filter, we can use the software named Filtering Solutions to do some computer aided design. The high performance filter is shown in Fig. 5.4 System performance test4.1 The measurement of the system’s self-noiseThe system is plugged in ±15V DC power supply with the input shorted to GND. The amplifier gain was adjusted to Av=40dB、60dB、80dB. The oscilloscope was used to observe the output noise waveforms of each amplifier and the Agilent 34401A was used to measure the RMS of noise voltage, the measure results are shown in TABLE I.4.2 The test of zero-drift correctionKeeping the input shorted and use the MCU to control the digital potentiometer, adding the compensation voltage by a step of 20 dB, adjusting the gain of amplifier in order to suppress the DCzero-drift .Use Agilent 34401A to measure the correction voltage under different magnifications, the results are shown in TABLE II.4.3 The calibration of the gainSet the working frequency band at 0~10MHz and the input signal frequency fi = 2MHz. Respectively,set the gain of amplifier separately at 0、20dB、40dB、60dB、80dB and input the appropriate signal amplitude Vi, use dual-channel oscilloscope to observe the input and output, record the output signal amplitude, calculate amplifier real gain and make comparisons with the set, the results are shown in TABLE III.4.4 Amplitude-frequency characteristics of the systemFix the amplitude of the input signal Vi = 100mVpp, AV=40dB, adjust the signal frequency between0~20MHz, then use the oscilloscope to observe VPP of the output signals with different frequency input signals and record them. Draw the curve of amplitude-frequency characteristic with MATLAB [20],which is shown in Fig.6.5 ConclusionIn this paper, the key technology of the ultra-wideband low-noise DC amplifier was researched. A high performance amplifier based on ultra-low-noise pre-amplifier, LC filter network, digital program gain control and zero-drift correction circuit was presented. The D/A converter was adopted to control the low-noise wideband amplifier VCA810, and the dynamic voltage gain range 0~80dB was achieved, the linear phase low-noise filter with band 0~10MHz was realized with the passive wideband Bessel low-pass filter, which composed by the inductance and capacitance, matched up with the low-noise preamplifier AD797, the equivalent input noise less than 7.2 μVrms was guaranteed. The MCU was used to control the digital potentiometer X9C102 to add compensation voltage to the zero-set end in order to realize the auto-adjustment of DC zero-drift. The test results show that the amplifier designed works with low-noise, small offset, high cost-effective, great stability and reliability.6 AcknowledgmentThe authors would like to thank Tiande Gao and Linwei Tao for help in the experiment, Zengxiang Fu and Hai Huang for advice in English writing.。

初级听觉皮层预测相关神经机制

目录缩略语表 (1)英文摘要 (2)中文摘要 (8)论文正文初级听觉皮层预测相关神经机制 (12)前言 (12)第一部分听皮层神经元节律性声音预测信号的检测 (15)1.1 材料与方法 (15)1.2 结果 (19)1.3 讨论 (37)第二部分听皮层节律性声音预测性信号的行为学表征 (39)2.1 材料与方法 (39)2.2 结果 (41)2.3 讨论 (46)第三部分听皮层活动在节律性声音预测性行为中的关键作用研究 (48)3.1 材料与方法 (48)3.2 结果 (49)3.3 讨论 (58)第四部分节律性声音预测性行为不依赖于听觉丘脑的活动 (60)4.1 材料与方法 (60)4.2 结果 (61)4.3 讨论 (67)全文总结 (71)参考文献 (72)文献综述双光子荧光成像技术在神经科学中的应用 (79)参考文献 (88)攻读博士期间发表的论文 (92)致谢 (93)缩略语表英文缩写英文全名中文全名Au1 Primaryauditorycortex 初级听觉皮层Cal-520 AM Cal-520 acetoxymethyl ester 卡尔-520乙酰氧基甲酯OGB-1AM Oregon Green BAPTA-1AM 俄勒冈州绿色四乙酸乙酰氧基甲酯SPL Sound pressure level 声压级ISI Inter-stimulus-interval 刺激间隔L2/3 Layer2/3 第2/3层S2 Secondary somatosensory cortex 次级体感皮层ArchT Archaerhodopsin 古紫质ChR2 Channelrhodopsin-2 视紫红质-2Cre Cis-acting regulatory element 顺式作用元件调节因子loxP Locus of X-over P1 噬菌体P1交换位点PV Parvalbumin 小清蛋白rAA V Recombinant adeno-associated virus 重组腺相关病毒YFP Yellow fluorescent protein 黄色荧光蛋白GFP Green fluorescent protein 绿色荧光蛋白GAD Glutamatedecarboxylase 谷氨酸脱羧酶WT Wildtype 野生型MGB Medial geniculate body 内侧膝状体MGBv Ventral division of medial geniculate body内侧膝状体腹侧部MGBd Dorsal division of medial geniculate body 内侧膝状体背侧部SR101 Sulforhodamine101 磺酰罗丹明PFA Paraformaldehyde 多聚甲醛CTB Cholera-toxinB 霍乱毒素B亚基EEG fMRI ElectroencephalographyFunctional magnetic resonance imaging脑电图功能磁共振成像Primary auditory cortex is required foranticipatory motor responseAbstractThe ability of the animals to respond to environmental information properly and timely is crucial for their living. One aspect of this ability isto predict future events based on the pattern of recent sensory experience. This is critical for animal’s cognition, decision making and guiding behavior. For example, following rhythmic sensory stimulation that normally triggers a movement, animals often “over-react” after the cessation of simulation by moving once or more at the expected time of the next stimulus. In addition, impaired ability to anticipate stimuli in the immediate future, as indicated by both brain activity measurements and neuropsychological performance, is often found in psychiatric disorders, including, for example, schizophrenia and anxiety. However, comparatively little is known about the underlying neural circuitry and the brain regions that are required for both coding expectation and generating anticipatory motor behavior. Such information may be relevant for understanding circuit malfunction in these disorders. Neocortical circuits for immediate processing of sensory stimuli are extensively studied, but their contributions to the anticipation of upcoming sensory stimuli remain less understood. We therefore used in vivo cellular imaging, fiber photometry and optogenetic activation and inhibition to elucidate the role of primary auditory cortex (Au1) in processing anticipated stimulation.Objectives:1. To explore the neuronal anticipatory response in the Au1 and its behavioral relevance following rhythmic sound sequence.2. To explore the causal relationship between Au1 neuronal anticipatory response and anticipatory behavior response.3. To explore the necessity of the auditory thalamus in the generation of anticipatory behavior response.Methods:1. Two-photon Ca2+ imaging in Au1The highly sensitive fluorescent Ca2+ indicator Cal-520 AM was used for multicell bolus loading in Au1. Ca2+ imaging was performed ~1 hour after dye injection and lasted for up to 8 hours. The body temperature of mouse was kept between 36.5–37.5 °C throughout the experiments. For two-photon imaging in head-fixed awake mice, the mouse underwent head-fixation training for 3-5 days (from 1 to 4 hours per day).2. Behavioral taskWe developed a simple sound-licking task for behavioral experiments. A drop of water (duration 20 ms) was provided from a spout 100 ms after the end of each sound stimulus (broadband noise, 76 dB SPL, duration 100 ms). The mouse was trained to associate sound with water supply and thus licked the water spout in a short time window after each sound. The mouse was used for experiments if its success rate was above 80%.3. Simultaneous optical-fiber and behavior recordingAn optical fiber was placed on the dura of each cortical site and then fixed on the skull with dental cement. To excite the fluorescent dye (OGB-1AM), continuous laser light was delivered at the tip of fiber. The emitted fluorescent light was collected via the same fiber. The licking behavior was monitored with a camera (frame rate: 30 Hz) under infrared illumination at the same time.4. Optogenetic activation and inhibitionThy1-ChR2-YFP or virally-injected mice were trained for the sound-licking task before optogenetic experiments. Two optical fibers were placed into L5 of bilateral auditory cortices or bilateral MGB regions under isoflurane anesthesia. The behavioral experiments started at least 2 h after the end of anesthesia. The light stimulus consisted of a single pulse of 473 nm blue light or 593 nm yellow light, which was applied with a duration of 100 ms, followed 100 ms later by a drop of water (20 ms).Results:1. Observation of echo responses following rhythmic sound stimulation in Au1 neurons1.1 Au1 L2/3 neurons show echo responses following rhythmic sound stimulation in anesthetized and awake MiceWe first explored the activity of L2/3 neurons of Au1 using two-photon Ca2+ imaging with Cal-520 AM. Simultaneous electrophysiological recordings of somatic action potentialsin Cal-520-labeled neurons revealed that Cal-520 signals could reliably report single action potentials with near 100% success rate. The calcium transients of Au1 L2/3 neurons were recorded before, during and after isochronous broadband noise stimuli (duration 100 ms, 76 dB sound pressure level, inter-stimulus-interval of 2 s) in anesthetized or awake state under two-photon microscope. Following the end of the last sound stimulus, about 15% (anesthetized state) or 21.5% (awake state) of neurons produced 1-3 extra responses at the approximate repetition rate of the sequence, although no sound was played. We referred to these extra responses as echo responses. Rhythmic sound stimuli at another inter-stimulus-interval (ISI), 4 s, also induced echo responses (362 imaged neurons from 3 mice). In contrast, stimuli delivered at alternating ISIs of 2 s and 3 s did not induce significant echo response (669 imaged neurons from 5 mice). In addition to the ISIs of 2 s and 4s, we also tested more ISIs, including 1 s, 3 s, 6 s and 10 s with 20 rhythmic stimuli. We found that stimuli at an ISI of 3 s significantly induce echo responses. In contrast, we did not observe significant echo response at an ISI of 1 s, or 6 s or 10 s. In addition to broadband noise stimulation, we also tested the sequence of pure tones at three different sound intensities. We found that all pure tones at different intensities were able to evoke significant echo responses, indicating no dependence on stimulus type and intensity. Furthermore, we tested sequences of 5, 10 or 30 stimuli, and found that at least 10 stimuli were necessary for producing reliable echo responses. Thus, neurons in L2/3 of Au1 exhibited significant echo responses corresponding to the ISI of the preceding sound sequence in both anesthetized and awake mice.1.2 Au1 L5 and L4 neurons show echo responses following rhythmic sound stimulationTo achieve optimal imaging quality for deep layer neurons, we minimized fluorescence from superficial layers by injecting Cal-520 AM at a depth of ~600 µm. Ca2+ imaging was restricted to the neurons located between 550 to 750 µm, corresponding to the cells within L5. We found echo responses in ~33.5% L5 neurons following rhythmic stimuli with an ISI of 2 s. We next imaged the neurons within the main thalamorecipient layer, L4, to rhythmic sound stimulation. 1934 imaged neurons from 13 mice were obtained at depths in the range of 400 to 500 µm. We observed echo responses in only 8% of these imaged neurons, with a latency approximately equal to the ISI.2. Mice show echo licking responses in the sound-licking task following rhythmic sound stimulationTo explore potential behavioral consequence of the neuronal echo responses in Au1, we developed a simple sound-triggered licking task, in which mice were trained to detect auditory stimulation (broadband noise, duration 100 ms, 76 dB SPL) and report sound by licking a spout for water reward. We then tested behavioral echo responses in trained mice with a train of 10 rhythmically repeated sound stimuli at an ISI of 3 s or 4 s. All the mice tested showed one more licking response at the time point that closely corresponded to the ISI of the preceding sound sequence. Similar to the cellular echo responses in Au1, five repeated sound stimuli were insufficient for producing echo licking responses. Next, we simultaneously monitored licking behavior and neuronal activity in Au1 using fiber photometry that allows for the fluorometric detection of action potentials of a group of neurons. For control, we placed the second optical fiber onto the rostral secondary somatosensory cortex (rostral S2). In 8 trained mice, we found that 98% ± 1% of the sound-evoked licking responses and 84% ± 7% of the echo licking responses were associated with Ca2+ responses in Au1. Similarly, 94% ± 4% and 88% ± 2% of the Ca2+ responses in Au1 were associated with licking responses during repeated sound stimuli and at the first echo time point respectively. Besides, according to a signal detection theory analysis, we can see that, in ~80% of all echo licking trials, the echo licking responses can be predicted by the echo Ca2+ responses measured by photometry in Au1. On the other hand, sound stimulation was not associated with Ca2+ responses in rostral S2. Rostral S2 did not show any response associated with licking at the first echo time point either (n = 4 mice). These results suggest a strong correlation between behavioral echo responses and neuronal activity in Au1.3. The key role of Au1 in behavioral echo responses3.1. Optogenetic inhibition of Au1 impairs behavioral echo responsesTo determine the necessity of Au1 neuronal activity in producing behavioral echo responses, we performed two optogenetic silencing experiments. First, we suppressed the activity of excitatory neurons in Au1 by activating the light-driven outward proton pump archaerhodopsin (ArchT), causing rapid and effective inhibition of neuronal activity. We then measured the licking responses during and after rhythmic sound stimulation. We found that light presentation produced a significant reduction in the rate of echo licking responses ascompared to the control mice, reaching a level slightly higher than chance level (probability of observing licking activity within a time window of the same duration preceding sound stimulation). In the second optogenetic silencing experiment, we used an alternative approach for inactivating Au1. We took advantage of Cre-loxP recombination to express channelrhodopsin-2 (ChR2) in parvalbumin (PV)-expressing inhibitory neurons. We bilaterally injected rAA V-EF1α-DIO-ChR2-mCherry to the Au1 of Pvalb-cre transgenic mice. In these virally-injected mice, blue light reduced the success rate of echo licking responses to chance level. In control mice, blue light presentation did not affect echo licking responses. Together, these two experiments suggest that Au1 is a critical hub in the circuit that generates anticipatory motor responses following rhythmic auditory stimulation.3.2 Optogenetic activation of Au1 induces behavioral echo responsesWe tested whether a direct activation of principal neurons in Au1 using ChR2 can drive mouse echo licking behavior in our auditory associative learning task. Since L5 neurons provide the major source of outputs from Au1 and also show echo responses following rhythmic sound stimulation, we used a previously reported mouse line, in which high levels of ChR2 and YFP are expressed mainly in L5 neurons under the Thy1 promoter. In mice that were already trained for the licking task, optogenetic activation of L5 neurons in Au1 bilaterally, substituting for sound stimulation, evoked reliable licking response in all animals tested. For control, neither naïve mice that expressed ChR2 (ChR2 naïve mice; n = 6 mice) nor trained mice that expressed no ChR2 (WT trained mice; n = 5 mice) showed licking in response to the same optogenetic stimulation. Importantly, in ChR2-expressing mice that were trained to lick following auditory stimulation, rhythmically repeated light stimuli readily produced rhythmic licking behavior as well as echo licking responses after the end of the stimulation sequence. The latency of echo licking responses following the last light stimulus closely matched the ISI. In 5 trained mice, we also combined the recordings of licking behavior with optical fiber-based measurements of population neuronal activity in Au1 L5 while delivering rhythmic light stimuli. We found that both immediate and echo licking responses induced by optogenetic stimulation were tightly associated with the neuronal responses in Au1. In addition, we bilaterally injected rAA V-CaMKIIα-ChR2-mCherry to Au1. As in the Thy1-ChR2-YFP transgenic mice, delivering blue light bilaterally through two fibers placed above the virally transduced Au1 neurons (~500 µm deep from the corticalsurface) evoked reliable licking responses in all trained mice (n = 7 mice). As expected, in CaMKIIα-ChR2-mCherry-expressing trained mice, rhythmically repeated light stimuli evoked clear echo licking responses following the end of the light sequence. The latency of the echo licking responses following the last light stimulus closely matched the ISI (n = 7 mice). These results show that anticipatory motor response can be effectively triggered by optogenetic activation of a subgroup of Au1 neurons.4. The auditory thalamus is not necessary for echo responsesIn both MGBv and MGBd, we reliably observed sound-evoked responses to a train of 10 broadband noise stimuli with an interval of 4 s. However, we did not detect any echo response following the end of sound stimulus sequence in these two MGB divisions. Furthermore, we conducted optogenetic silencing experiments by inactivating ArchT-expressing MGB neurons during the licking task. We found that delivering light stimulation at a wavelength of 593 nm through two fibers (400 µm diameter) bilaterally placed above the transfected MGB regions had no significant effect on the echo licking responses. Therefore, these results suggest that auditory thalamus plays no or minor (if any) role in the generation of behavioral echo responses.Conclusions:1. Anticipatory responses exist in Au1 L2/3, 4 and 5 neurons.2. Au1 anticipatory neuronal response is a key element in the generation of anticipatory behavior response.3. The auditory thalamus is not necessary for anticipatory responsesKey words: anticipatory motor response; primary auditory cortex; predictive coding;two-photon calcium imaging; rhythmic sound stimulation;初级听觉皮层预测相关神经机制摘要及时地对环境信息做出合适的行为反应是动物在进化过程中获得的一项基本能力，其中一个很重要的方面就是动物基于过去新近感觉经验，提炼出规律，进而根据这个规律来预测即将到来的感觉信息。

RF2484资料

4 Pins 1 and 9 are fused. 5 Package Warpage: 0.05 max.
GND VCC
PD RF OUT
GND
Optimum Technology Matching® Applied
Si BJT
üGaAs HBT
GaAs MESFET
Si Bi-CMOS
SiGe HBT
Parameter
LO Input
Frequency Range
Power Level
5
Input Impedance
Modulation Input
Frequency Range Reference Voltage (VREF) Input Resistance Input Bias Current
• W-CDMA Base Stations • WLAN and WLL Systems • GMSK,QPSK,DQPSK,QAM Modulation
MODULATORS AND UPCONVERTERS
Product Description
The RF2484 is a monolithic integrated quadrature modu-
Sideband Suppression>35dBc over temperature with highly linear operation • Noise Floor better than -152dBm/Hz from 800MHz to 2200MHz • Single 5V Power Supply
LO=-5dBm at 1960MHz; Single sideband testing unless otherwise noted For ACPR =-72dBc; I&Q Amplitude=1.2VPP (single-ended) Channel Power=-13dBm; see Test Setup for detailed information T=25°C; POUT=-13dBm; optimized I,Q DC offsets Temperature cycled from -40°C to +85°C after optimization at T=25°C; POUT=-13dBm T=25°C; POUT=-13dBm; optimized I,Q amplitude and phase balance Temperature cycled from -40°C to +85°C after optimization at T=25°C; POUT=-13dBm

Guidelines for the reduction of underwater noise from commercial shipping to address adverseimpacts

E
4 ALBERT EMBANKMENT
LONDON SE1 7SR
Telephone: +44 (0)20 7735 7611
Fax: +44 (0)20 7587 3210
MEPC.1/Circ.833 7 April 2014
GUIDELINES FOR THE REDUCTION OF UNDERWATER NOISE FROM COMMERCIAL SHIPPING TO ADDRESS ADVERSE IMPACTS ON MARINE LIFE
3.2 Given the complexities associated with ship design and construction, the Guidelines focus on primary sources of underwater noise. These are associated with propellers, hull form, onboard machinery, and operational aspects. Much, if not most, of d by propeller cavitation, but onboard machinery and operational modification issues are also relevant. The optimal underwater noise mitigation strategy for any ship should at least consider all relevant noise sources.
4.3 Underwater noise, or the underwater-radiated noise level, for the purposes of these Guidelines refers to noise from commercial ships*.

电信术语词典

电信术语词典A∙A和B比特（A & B Bit）∙ A 链接（A Links）∙ A 和 B 比特信令（A&B Bit Signaling）∙ABAM 电缆（ABAM cable）∙呼叫取消（Abandoned Call）∙无人接听回应（Abandonment）∙缩写的地址（Abbreviated address）∙缩写地址呼叫（Abbreviated Address Calling）∙缩位拨号（Abbreviated Dialing）∙ABCD 信号（ABCD signaling）∙ABCD 信号比特（ABCD Signaling Bits）∙ABCD 音调（ABCD Tones）∙ABDN：号码接线模块（Attendant Blocking of Directory Number）∙外出用户电话服务（Absent Subscriber Service ）∙绝对延误（Absolute delay）∙吸收频带（Absorption Band）∙吸收系数（Absorption coefficient）∙吸收频谱（Absorption Spectrum）∙AC/DC 铃声（AC/DC Ringing）∙ACA：自动电路保证（Automatic Circuit Assurance）∙ACB：自动回电（Automatic Call Back）∙接入试呼叫（Access Attempt）∙使用费（Access Charge）∙接入编码（Access Code）∙接入失败（Access Failure）∙接入组群（Access Group）∙接入线（Access Line）∙接入链接（Access Link）∙接入号码（Access Number）∙接入保护（Access Protection）∙接入速度（Access Rate）∙接入中继（Access Tandem）∙帐号代码（Account Code）∙Accunet光谱数码服务（Accunet spectrum digital service）∙ACD 回电话信息（ACD Call Back Massaging）∙ACD 呼叫人直呼路由（ACD Caller Directed Call Routing）∙ACD 中央电话总机（ACD Central Office）∙ACD有条件的路由（ACD Conditional Routing）∙ACD 数据直接呼叫路由（ACD Data Directed Call Routing）∙ACD 智能呼叫处理（ACD Intelligent Call Processing）∙ACD 号码（ACD Number）∙ACD：自动呼叫分配器（Automatic Call Distributor）∙直流－交流振铃（AC-DC Ringing）∙ACELP：代数码激励线性预测（Algebraic Code Excited Linear Prediction）∙音频连接（Acoustic Connection）∙ACSB：振幅压缩扩展单边带（Amplitude Compandored Sideband）∙ACTA：美国电讯运营商协会（America's Carriers Telecommunications Association）∙正在通话中的来电（Active Call）∙积极的活动（Active Campaign）∙正常工作的线路（Active Line）∙自动噪音控制（Active Noise control）∙主动接听（Active Participation）∙ACTS：高级通讯技术和服务（欧洲）（Advanced Communications technologies and Services (Europe)）∙ACTS：自动电话投币服务（Automatic Coin Telephone Service）∙ACU：自动呼叫单元（Automatic Calling Unit）∙ACX：异步交叉连接（Asynchronous Cross-connect）∙ADA：平均放弃延长时间（Average Delay to Abandon）∙ADACC：自动完成目录查询协助呼叫（Automatic Directory Assistance Call Completion）∙ADAD：自动拨号和回应装置（Automatic Dialing and Announcing Device）∙适应性的通道分配（Adaptive Channel Allocation）∙ADAS：自动地址目录查询帮助服务（Automated Directory Assistance Service）∙ADC：模拟信号到数字转化器（Analog-to-Digital Converter）∙允许第三方加入的电话功能（Add-on Conference）∙外加的数据模块（Add-on Data Module）∙完整地址信息（Address Complete Message）∙地址信息（Address Message）∙地址信号的发送（Address Signaling）∙指定的呼叫模式（Addressed Call Mode）∙ADE：自适应的设计工程（Adaptive Design Engineering）∙临近的信号发送点（Adjacent Signaling Points）∙ADM：加减多路复用器（Add-Drop Multiplexer）∙管理模块（Administrative Module）∙管理干线群（Administrative Trunk Group）∙ADPCM：适配性差异脉冲编码调制（Adaptive Differential Pulse-Code Modulation）∙高级交换分机（Advanced Branch Exchange）∙高空电缆（Aerial Cable）∙航空应急通讯系统项目（Aeronautical Emergency Communications System Plan）∙整合设备（Aggregation Device）∙AHT：平均处理时间（Average Handle Time）∙AIC：自动旁录中心（Automatic Intercept Center）∙AIDR：现有的数据速率（As-Is Data Rate）∙AIN：高级智能网络（Advanced Intelligent Networks）∙AIOD：自动识别向外拨号（Automatic Identified Outward Dialing）∙全球地对空通讯系统（Air-ground worldwide communications system）∙AIS：警报指示信号（Alarm Indication Signal）∙AIS：自动旁录系统（Automatic Intercept System）∙A-Law∙提示信号（Alerting signal）∙提示铃声（Alerting Tone）∙A-Link：SS7通道链接（SS7 Access link）∙所有干线繁忙（All Trunk Busy）∙同质异晶（Allomorphism）∙交替通道载波器（Alternate Access Carriers）∙交替传号反转信号（Alternate Mark Inversion Signal）∙交替通话人（Alternate party）∙AMA：通话自动计费（Automatic Message Accounting）∙环境噪声水平（Ambient Noise Level）∙美国线规（American Wire Gauge）∙美国科技（Ameritech）∙AMI：交替传号反转（Alternate Mark Inversion）∙电话放大听筒（Amplified Handset）∙ANAC Number：自动数字提示线路号（Automatic Number Announcement Circuit Number）∙模拟桥（Analog Bridge）∙模拟载波系统（Analog Carrier System）∙模拟通道（Analog Channel）∙模拟数据（Analog Data）∙模拟解码（Analog Decoding）∙模拟传真（Analog Facsimile）∙模拟循环（Analog Loop-back）∙模拟信号（Analog Signal）∙模拟交换机（Analog Switch）∙AND：自动网络拨号（Automatic Network Dialing）∙ANI：发信号码的自动是别装置（Automatic Number Identification）∙非等时同步（Anisochronous）∙ANM:应答消息（Answer Message）∙拨号回应服务（Announcement Service）∙拨号回应系统（Announcement System）∙骚扰电话处理中心（Annoyance Call Bureau）∙信号器（Annunciator）∙匿名电话阻止（Anonymous Call Rejection）∙匿名电话号码（Anonymous telephone Number）∙ANSA：交替网络服务协议（Alternate Network Service Agreement）∙回应代码（Answer Back）∙APON：异步传输模式（ATM）的无源光网络（ATM Passive Optic Network）∙APS：自动保护系统（Automatic Protection System）∙电话区号（Area Code）∙地区交换机（Area Exchange）∙算术移位（Arithmetic Shift）∙ARQ：自动重复与要求（Automatic Repeat & Request）∙清晰度百分数（Articulation Score）∙ASA：平均回答时间（Average Speed of Answer）∙ASC：公认标准委员会（Accredited Standards Committees）∙ASE：应用服务单元（Application Service Element）∙ASK：幅变调制（Amplitude Shift Keying）∙ASN：AT&T交换系统（AT&T Switched Network）∙ASP：应用软件服务供应商（Application Service Provider）∙关联模式（Associated Mode）∙异步操作（Asynchronous Operation）∙ATA：模拟终端适配器（Analog Terminal Adapter）∙ATA：等待回应的平均时间（Average Time to Answer）∙ATB：中继线路全部繁忙（All Trunks Busy）∙ATDM：异步时间分割多路技术（Asynchronous time-division multiplexing）∙ATIS：电讯工业方案联盟（Alliance for Telecom Industry Solutions）∙起始时间（Attack Time）∙试呼叫（Attempt）∙衰减常数（Attenuation Constant）∙衰减－串音的比率（Attenuation to crosstalk ratio）∙ATU-C：中央传输单元（ADSL Transmission Unit-Central Office）∙ATU-R：远程传输单元（ADSL Transmission Unit-Remote）∙占线语音提示（Audible busy tone）∙可闻振铃声（Audible Ringing Tone）∙可听音（Audible Tones）∙自动呼叫重新连接（Automatic Call Reconnect）∙自动探通术（Automatic sounding）∙自动交换系统（Automatic Switching System）∙附属服务干线群（Auxiliary Service Trunk Groups）∙可用的线路（Available Line）∙AVBO：先进的语音因忙退出（Advanced Voice Busyout）∙平均负载（Average Load）∙AXETOP▲B∙双极讯号三零替换（B3ZS）∙双极讯号六零替换（B6ZS）∙B8ZS：双极讯号八零替换（Bipolar 8 Zero Substitution）∙干扰音（Babble）∙主干线布线（Backbone Cabling）∙反向通道（Back-channel）∙后台噪音（Background Noise）∙迂回信程（Back-Haul）∙背对背连接（Back-to-back connection）∙反向信道（Backward Channel）∙反向信号（Backward signal）∙BAF：贝尔自动信息计费格式（Bellcore AMA Format）∙回波损耗平衡（Balance Return Loss）∙平衡阻抗（Balanced Impedance）∙平衡线路（Balanced Line）∙投票（Ballot）∙BAN：收费帐号（Billing Account Number）∙波段（Band）∙带阻过滤器（Band Elimination Filter）∙统一收费价格（Banded Rates）∙宽带上限（Bandwidth Cap）∙带宽压缩（Bandwidth Compression）∙带宽受限运行（Bandwidth Limited Operation）∙带宽遏流（Bandwidth throttling）∙基本服务（Basic Service）∙BCC：信息组校验符号（Block Check Character）∙BCD：二-十进制计数法（Binary Coded Decimal）∙BCH Code：多个随机错误模式校正编码（Bose, Ray-Chaudhuri, Hocquenghem Code）∙ B 通道捆绑（B-Channel Bundling）∙BDCS：宽带数字交叉连接系统（Broadband Digital Cross-Connect System）∙贝尔用户编码（Bell Customer Code）∙贝尔核心标准（Bellcore）∙BER：比特误差比率（Bit Error Rate）∙BERT：比特误差率检测（Bit Error Rate Tester）∙BHCA：忙时呼叫尝试（Busy Hour Call Attempt）∙BIB：反向指示比特（Backward Indicator Bit）∙双向开关（Bidirectional Switch）∙两端分开（Bifurcated）∙横列定向天线（Billboard Antenna）∙Bill-To-Room∙二进制戈莱码（Binary Golay Code）∙二进位信号（Binary Signal）∙二进制交换（Binary Switch）∙双相符号编码（Biphase Mark Code）∙双极性（Bipolar）∙双极编码（Bipolar Encoding）∙双极信号（Bipolar Signal）∙比特配对（Bit Pairing）∙比特损失（Bit Slip）∙比特填充（Bit Stuffing）∙同步比特（Bit Synchronous）∙比特值的完整性（Bit-count integrity）∙双三进制传输（Biternary Transmission）∙BITS：建筑物综合定时供给系统（Building Integrated Timing Supply）∙黑色噪音（Black noise）∙黑色记录（Black recording）∙盲信号传输（Blind Transmission）∙B-link：桥接链路（Bridge-link）∙被阻挡的拨号尝试（Blocked Attempt）∙通讯被阻挡（Blocking）∙通讯被阻挡的概率（Blocking probability）∙BOC：贝尔营业公司（Bell Operating Company）∙BPON：宽带无源光网络（Broadband Passive Optic Network）∙BPV：双极性违规（Bipolar Violation）∙BRAS：宽带远程接入服务器（Broadband Remote Access Server）∙BRI：基本速率接口（Basic Rate Interface）∙BR-ISDN：ISDN 基本速率（ISDN Basic Rate）∙宽带互联网接入（Broadband Internet Access）∙宽带开放接口（Broadband open access）∙宽带电话（Broadband Telephony）∙BSE：基本服务项目（Basic Service Element）∙BSN：逆序号码（Backward Sequence Number）∙英国电讯集团（BT Group）∙高峰时段（Busy Hour）∙占线信号（Busy Signal）∙忙音（Busy Tone）∙占线确认音（Busy Verification Tone）∙旁路干线群（Bypass Trunk Group）TOP▲C∙布线管理（Cabling Administration）∙CAC：连接(呼叫)准入控制（Connection (or Call) Admission Control）∙CALEA：执法中的通讯协助法案（Communications Assistance for Law Enforcement Act）∙呼叫（Call）∙电话计费系统（Call Accounting System）∙呼叫中心（Call Center）∙呼叫冲突（Call Collision）∙呼叫转移（Call Deflection）∙呼叫持续时间（Call Duration）∙呼叫过滤（Call Filters）∙呼叫转移（Call Forwarding）∙呼叫管理（Call Management）∙呼叫方（Call Originator）∙呼叫优先权（Call Priority）∙呼叫处理信号（Call Process Signals）∙呼叫设置（Call Set-up）∙呼叫设置时间（Call Setup Time）∙呼叫转移（Call Transfer）∙呼叫等待（Call Waiting）∙呼叫等待音（Call Waiting Tone）∙被呼叫方（Called Party）∙呼叫方的ID（Caller ID）∙呼叫卡（Calling Card）∙呼叫卡服务提示音（Calling Card Service Prompt Tone）∙呼叫方（Calling Party）∙CAM：无载波调幅（Carrierless Amplitude Modulation）∙CAMA：集中式自动化通话记帐制（Centralized Automatic Message Accounting）∙预占（Camp-On）∙CAP：无载波幅度/相位调制技术（Carrierless Amplitude Phase modulation）∙CAP：竞争性接入供应商（Competitive Access Provider）∙访问限制（Capping）∙Carrier∙载波频率（Carrier Frequency）∙运营商数据交换中心（Carrier Hotel）∙载波服务供应商（Carrier service provider）∙载波系统（Carrier system）∙载波（Carrier wave）∙卡森宽带规则（Carson bandwidth rule）∙CAS：随路信号（Channel Associated Signaling）∙临时呼叫（Casual Calling）∙临时呼叫用户（Casual Customer）∙事由表示号码（Cause Codes）∙CCBS：被呼叫人忙时暂停呼叫（Completion of Call to Busy Subscriber）∙CCH：每条电路每小时接续次数（Connections per Circuit Hour）∙CCITT：国际电报电话咨询委员会（Consultative Committee for International Telegraph and Telephone）∙CCITT：国际电报电话咨询委员会（International Telegraph and Telephone Consultative Committee）∙CCS：百秒呼叫（Centi Call Seconds）∙CCS：公共信道信令（Common Channel Signaling）∙CCSS7：7号公共通讯通道信令（Common Channel Signaling System 7）∙CCT：导通检测收发器（Continuity Check Transceiver）∙CDB：呼叫细节模块（Call Detail Block）∙CDE：呼叫细节元素（Call Detail Element）∙CDR：呼叫细节记录（ Call Detail Record）∙CDR：呼叫细节报告（Call Detail Reporting）∙CED：呼叫人输入的数字（Caller-entered Digits）∙CELP：代码激励线性预测（ Code Excited Linear Prediction）∙中心频率（Center Frequency）∙中间部分（Central Member）∙集中截取局指令信号（Centralized Intercept Bureau Order Tone）∙中央区（Centrex）∙CEPT：欧洲邮政和电信会议（Conference of European Posts and Telecommunications）∙CFB：来电转接占线（Call Forwarding Busy）∙CFNR：无应答呼叫前转（Call Forward No Reply）∙CFU：无条件呼叫前转（Call Forward Unconditional）∙回充（Change-back）∙转换（Change-over）∙通道（Channel）∙随路信令（Channel Associated Signaling）∙通道噪音水平（Channel Noise Level）∙信道化（Channelized）∙信道化的E1（Channelized E1）∙信道化的T1（Channelized T1）∙检测比特（Check Bit）∙校验数位（Check Digit）∙用户流失（Churn）∙CIC：运营商标识码（Carrier Identification Code）∙CIC：线路识别码（Circuit Identification Code）∙CIR：承诺信息速率（Committed Information Rate）∙线路（Circuit）∙线路噪音水平（Circuit Noise Level）∙线路交换（Circuit Switching）∙电路传送模式（Circuit Transfer Mode）∙电路标准反转多任务（Circuit-level Inverse Multiplexing）∙循环器（Circulator）∙都市广域数据中心（City Wide Digital Centrex）∙覆层（Cladding）∙四类交换机（Class 4 Switch）∙五类交换机（Class 5 Switch）∙识别音分类（Class of Service Tone）∙CLASS：自定义本地信令服务（Custom Local Area Signaling Service）∙无噪音电流（Clean Power）∙纯信道（Clear Channel）∙后向挂机信息（Clear-back Message）∙向前拆线信息（Clear-forward Message）∙CLEC：竞争性地区通信运营商（Competitive Local Exchange Carrier）∙CLID：呼叫线路识别（Calling Line Identification）∙C-link：横向链接（Cross Link）∙声音剪辑（Clipping）∙CLLI Code：共同语言、方位识别码（Common Language Location Identification Code）∙CMCU：通讯模块控制单元（Communications Module Control Unit）∙C信息（C-message）∙CMI：编码传号反转码（Coded Mark Inversion）∙CMS：呼叫管理系统（Call Management System）∙CNM：用户网络管理（Customer Network Management）∙CO：电话总局（Central Office）∙硬币收集语音（Coin Collect Tone）∙硬币面额语音（Coin Denomination Tone）∙投币电话（Coin Phone）∙硬币退出语音（Coin Return Tone）∙公用电话服务商（Common Carrier）∙通讯安全（Communication Security）∙通讯源（Communication Source）∙通讯子系统（Communication Subsystem）∙通讯系统（Communication System）∙通讯欺诈（Communications Deception）∙通讯线路（Communications Line）∙通讯模块（Communications Module）∙通讯保护（Communications Protection）∙压缩扩展（Companding）∙压缩扩展器（Compandor）∙完整的信号发送（Completed Signaling）∙压缩机（Compressor）∙集中器（Concentrator）∙传导干扰（Conducted Interference）∙会议电话模式（Conference Operation）∙确认音（Confirmation Tone）∙契约税（Contract Tariffs）∙控制面板（Control Plane）∙控制信号（Control Signal）∙受控制的改道发送（Controlled Re-routing）∙受控滑动（Controlled Slip）∙CONUS：美国连续地区（CONtiguous United States）∙卷积码（Convolutional Code）∙COT：中心局终端（Central Office Terminal）∙国家代码（Country Code）∙CPC：主叫用户类别（Calling Party Category）∙CPE：用户端设备（Customer Premises Equipment）∙CPG：呼叫状态（Call ProGress）∙CPNIE：被呼叫方号码信息原素（Called Party Number Information Element）∙CPSIE：被呼叫方非主要地址信息元素（Called Party Subaddress Information Element）∙非经允许的服务添加（Cramming）∙交叉连接（Cross Connect）∙交叉连接配置（Cross-Connection）∙交叉核查（Cross-office Check）∙CRP：用户路由节点（Customer Routing Point）∙CRV：呼叫参考值（Call Reference Value）∙CSA：电信公司营运服务区（Carrier Serving Area）∙CS-ACELP：共轭结构代数码激励线性预测（Conjugate Structure Algebraic Code Excited Linear Prediction）∙CSL：元件分层（Component Sub-Layer）∙CSU/DSU：通道服务单元/数据服务单元（Channel Service Unit/Data Service Unit）∙CSU：通道服务单元（Channel Service Unit）∙CTI：电脑电话整合技术（Computer Telephony Integration）∙CTS：清除发送（Clear-to-Sent）∙用户追查（Customer-Originated Trace）∙彻底替换（Cutover）∙直通拨号（Cut-Through Dialing）TOP▲D∙ D 信道（D Channel）∙ D 型的信道处理单元（D Type Channel Bank）∙DACS：数字接入与交叉连接系统（Digital access and cross-connect system）∙DAL：专用接入线路（Dedicated Access Line）∙暗光纤（Dark Fiber）∙数据集回应音（Data Set Answer Back Tone）∙数据过滤编码（Data Strobe Encoding）∙数据工具箱（Datakit）∙DATU：直达测试单元（Direct Access Test Unit）∙DCB：数字信道处理单元（Digital Channel Bank）∙DCE：数据终端设备（Data Circuit-terminating Equipment）∙DCF：色散补偿光纤（Dispersion Compensating Fiber）∙DCS：数字交叉连接系统（Digital Cross Connect System）∙DDD：直拨长途电话（Direct Distance Dialing）∙DDI：直接拨入（Direct Dialing In）∙解码（Decode）∙解码器（Decoder）∙专用线路（Dedicated Line）∙起止信号畸变程度（Degree of start-stop distortion）∙延迟拨号信令（Delay Dial Signaling）∙延迟失真（Delay Distortion）∙延迟均衡器（Delay Equalizer）∙增量调制（Delta Modulation）∙解调（Demodulation）∙解调器（Demodulator）∙投币音（Deposit Coin Tone）∙绕拨（Dial Around）∙杰克拨号音（Dial Jack Tone）∙长途拨号线路（Dial Long Line）∙非正常拨号音（Dial Off-Normal Tone）∙拨号盘脉冲（Dial Pulse）∙号盘脉冲（Dial Pulsing）∙拨号音（Dial Tone）∙拨号音迟延（Dial Tone Delay）∙拨号音标识器（Dial Tone Marker）∙走针轮系（Dial Train）∙拨号对等性（Dialing Parity）∙拨号正常传输信号（Dial-Normal Transmission Signal）∙拨号（Dialup）∙拨号连接（Dial-up Access）∙拨号线路（Dial-up Line）∙膜片（Diaphragm）∙DIB：目录信息库（Directory Information Base）∙DID/DNIS：直接向内拨号/拨号识别服务（Direct Inward Dialing/Dialed Number Identification Service）∙DID：直接向内拨号（Direct-Inward-Dial）∙差分信号（Differential Signaling）∙数字式（Digital）∙数字载波系统（Digital Carrier System）∙数字传输（Digital Transmission）∙数字化（Digitize）∙直接交互连接干线群（Direct Interlata Connecting Trunk Groups）∙直接逐行扫描系统（Direct Progressive System）∙定向增益（Directive Gain）∙电话查询服务（Directory Assistance）∙电话查询服务中继群（Directory Assistance Trunk Group）∙索引电话号码（Directory Number）∙不均等（Disparity）∙差量受限操作（Dispersion-limited operation）∙不同的呼叫等待语音（Distinctive Call Waiting Tone）∙分配光缆（Distribution Cable）∙多通道传输（Diversity）∙DLC：数位回路载波器（Digital Loop Carrier）∙D-Link：对角线链路（Diagonal Link）∙DLTU：数字线路和中继线单元（Digital Line and Trunk Unit）∙DMS：数字多路系统（Digital Multiplex System）∙DMS-10 数字交换系统（DMS-10 Digital Switching System）∙DMS-100 数字交换系统（DMS-100 Digital Switching System）∙DMS-200 数字交换系统（DMS-200 Digital Switching System）∙DMT：不连续的多频声（Discrete Multitone）∙DNIS：被呼叫号码识别服务（Dialed Number Identification Service）∙DOC：动态过载控制（Dynamic Overload Control）∙国防部通讯系统（DOD Communication System）∙DOD：直接对外拨号（Direct Outward Dialing）∙双重蜂音信号（Double Order Tone）∙双端引线同步（Double-ended Synchronization）∙DPC：目的地端编码（Destination Point Code）∙DPCM：差异脉冲编码调制（Differential Pulse-Code Modulation）∙DPSK：微分相移键控（Differential Phase Shift Keying）∙DRE：方向性的备用设备（Directional Reservation Of Equipment）∙漂移（Drift）∙下落（Drop）∙DS2：数字信号2（Digital Signal 2）∙DSB-RC：双边带减幅载波（Double-SideBand Reduced Carrier）∙DSB-SC：双边带抑制载波（Double-SideBand Suppressed Carrier）∙DSF：色散位移光纤（Dispersion Shift Fiber）∙DSI：数字话音插空技术（Digital Speech Interpolation）∙DSN：国防交换网络（Defense Switched Network）∙DSR：数据信令速率（Data Signaling Rate）∙DS-SMF：色散移位单模块光纤（Dispersion-Shifted Single Mode Fiber）∙DSU：数据服务单元（Data Service Unit）∙DSX：数字信号交叉连接（Digital Signal Cross-connect）∙数字信号交叉连接等级1（DSX-1）∙DTMF：双音多频（Dual Tone Multifrequency）∙DTP：数据传输过程（Data Transfer Process）∙双臂存取（Dual Access）∙同时抢用线路（Dual seizure）∙双向通讯（Duplex）∙DX 信令（DX Signaling）TOP▲E∙ E Channel：回声波道（Echo Channel）∙E&M前引作信（E&M Leads Signaling）∙E&M信令（E&M Signaling）∙E&M：接收和传输（recEive and transMit）∙ E.123∙ E.164∙E-2∙E-4∙E911 服务（E911 Service）∙EBAF：扩展式贝尔核心AMA格式（Extended Bellcore AMA Format）∙回声抵消器（Echo Canceler）∙回声消除（Echo Cancellation）∙ECSA：交换载波标准协会（Exchange Carriers Standards Association）∙EDFA：掺饵光纤放大器（Erbium-Doped Fiber Amplifier）∙EDP：事件检测点（Event Detection Point）∙有效的输入噪声温度（Effective input noise temperature）∙EIA/TIA-232∙EIA/TIA-449∙EIA/TIA-586∙EIA-232∙EIA-422∙EIA-422∙EIA-423∙EIA-449∙EIA-485∙EIA-530∙EIA-530∙E-ISUP：扩展ISUP（Extended-ISUP）∙有法定资格的电讯运营商（Eligible telecommunications carrier）∙E-Link：扩展链接（Extended link）∙嵌入式操作通道（Embedded Operations Channel）∙阻塞式信令（En-block Signaling）∙脉冲结束信号（End-of-pulse Signal）∙选择结束信号（End-of-selection signal）∙EPON：以太网无源光网络（Ethernet Passive Optical Network）∙平等接入（Equal Access）∙机房（Equipment Room）∙ERC：容易辨认的编码（Easily Recognizable Code）∙厄兰（Erlang）∙厄兰单位（Erlang Unit）∙ESF：扩展的超帧（Extended Super-Frame）∙ESN：电子交换网络（Electronic Switched Network）∙ESS：电子交换系统（Electronics Switching System）∙ETB：传输块末字符（End of Transmission Block Character）∙EURESCOM：欧洲电讯开发和战略研究所（European Institute for Research and Strategic Studies in Telecommunications）∙欧洲通讯卫星组织（Eutelsat）∙偶校验（Even Parity）∙交换机（Exchange）∙通信运营商（Exchange Carrier）∙伸幅转换器（Expandor）∙EXZ：额外的零值（Excessive Zeros）TOP▲F∙FAA：可接受设施（Facility Accepted）∙基于专用交换网络服务的设备（Facilities Based Private Switched Network Services）∙设备回路（Facility Loopback）∙自备设施运营商（Facility-based Carriers）∙Fax：传真发送（Facsimile Transmission）∙FBG：光纤光栅（Fiber Bragg Grating）∙FCC：联邦通讯委员会（Federal Communications Commission）∙FDL：设备数据链接（Facility Data Link）∙功能板（Feature Boards）∙FEC：前向纠错（Forward Error Correction）∙FER：帧擦除/错误率（Frame Erasure/Error Rate）∙FEXT：远端串音（Far-End Crosstalk）∙FG：功能组（Feature Groups）∙FGA：功能组 A（Feature Group A）∙FGB：功能组B（Feature Group B）TOP▲G∙G.703∙G.704∙G.707∙G.708∙G.709∙G.711∙G.721∙G.722∙G.722.1∙G.723 or G.723.1∙G.726∙G.727∙G.728∙G.729 A/B∙G.780∙G.781∙G.782∙G.783∙G.7xx∙G.803∙G.804∙G.832∙G.DMT∙G.DMT∙G.Lite∙Gateway Redundancy 网关冗余∙Gateway Supported Prefixes∙Gateway switched exchange 网关交换机∙GETS： Government Emergency Telecommunications Service 政府应急电信服务∙GIS： Geographical Information System 地理信息系统∙Globalstar 全球星∙GloBanD∙GNE： Gateway Network Element 网关网元∙GOS： Grade Of Service 服务级别∙GPON： Gigabit PON 吉比特PON∙Ground Start 地启动∙Ground start signaling 地启动信令∙Ground-start Trunk 地启动干线∙Group 3∙Group Busy Tone 组群忙音∙GTT： Global Title Translation 全局码转换TOP▲H∙H.225∙H.225.0∙H.235∙H.245∙H.248∙H.261∙H.263∙H.264∙H.323∙H.450.3∙Hagelbarger code 黑格巴哥码∙Halftone characteristic 半音特征∙Hamming code 汉明码∙Handset 手持电话设备∙Hardwired 硬线∙H-Channel： High-speed channel 高速通道∙HCO： Hearing Carry Over 听力辅助∙HCS： Header Check Sequence 标题校验序列∙HDB3： High Density Bipolar 3 高密度双极性码3∙HDSL： High Data Bit Rate DSL 高数据比特率数字用户线路∙HDSL2： 2nd generation HDSL 第二代高数据比特率数字用户线路∙HDSL4： 4th generation HDSL 第四代高数据比特率数字用户线路∙HDT： Host Digital Terminal 主机数字终端∙Headset 头戴语音设备∙Hierarchical Network 分级网络∙High Tone 高频音∙High Water Mark 高峰计量器∙High-priority emergency service call 高优先级应急服务呼叫∙High-Usage Group 高使用群∙Hit 线路瞬间干扰∙Holding Time占线时间∙Home Tandem 家庭串联组∙Homing Arrangement 自导引设置∙Hookflash 瞬间挂机∙Hop Off∙Horizontal Wiring 横向配线∙Howler Tone 嗥鸣音∙HPF： High Pass Filter 高通滤波器∙HPPI： High Performance Parallel Interface 高性能平行接口∙Hundred Call Seconds 百秒呼叫∙Hunt亨特单位∙Hybrid balance 混合平衡∙Hybrid Coil 混合线圈∙Hybrid Fiber Coaxial Network 混合光纤同轴网络TOP▲I∙I.N.：智能网（Intelligent Network）∙IAC：初始校准控制（Initial Alignment Control）∙IAD：综合接入设备（Integrated Access Device）∙IAM：初始地址消息（Initial Address Message）∙ICPIF：计算计划损伤元素丢失/延迟因忙退出开始（Calculated Planning Impairment Factor loss/delay busyout threshold）∙IDN：综合数字网络（Integrated Digital Network）∙IEC：国际电工委员会（International Electro-technical Commission）∙ILEC：现有地区通信运营商（Incumbent Local Exchange Carrier）∙即时开启（Immediate Start）∙立即开始信令（Immediate Start Signaling ）∙IMT：多机间中继线（Inter-Machine Trunk）∙INB：安装忙碌（Install Busy）∙带内信令（In-band signaling）∙线路预占指示（Indication of Camp-On ）∙INE：智能网络元素（Intelligent Network Element）∙INE：中间网络元素（Intermediate Network Element）∙INI：网络间界面（Inter-Network Interface）∙槽内信令（In-slot Signaling）∙亮度调制（Intensity modulation ）∙交互媒介（Interactive media）∙截取回路音（Intercepting Loopback Tone）∙交互电路（Interchange circuit）∙内部通讯系统（Intercom）∙交互干线（Inter-exchange trunk）∙交互本地访问和传输区域服务（Inter-LATA Services ）∙中间交叉连接（Intermediate Cross Connects）∙内部电话线路呼叫（Interoffice Call ）∙内部线路通道（Interoffice Channel）∙符号间干扰（Intersymbol interference ）∙IntraLATA∙局内中继（Intra-Office Trunk）∙州内电话（Intrastate）∙反转多路技术（Inverse multiplexing）∙内向干线（Inward Trunk）∙IN-WATS：内向广域电话业务（Inward Wide Area Telephone Service）∙IOC：独立自主的运营公司（Independent Operating Company）∙IP集中交换（IP Centrex）∙IP-PBX∙ISC：国际交换运营商（International Switching Carrier）∙综合服务数字网络（ISDN BRI）∙ISDN D通道信令（ISDN D-Channel Signalling）∙ISDN H通道（ISDN H-Channel）∙ISDN PRI：ISDN主要速率接口（ISDN Primary Rate Interface）∙ISDN：综合服务数字网络（Integrated Services Digital Network）∙同步信号（Isochronous Signal）∙ISUA：SS7 ISUP 用户适配层（SS7 ISUP-User Adaptation Layer）∙ISUP：ISDN 用户部分（ISDN User Part）∙ITC：独立的电话公司（Independent Telephone Company）∙ITSP：互联网电话服务供应商（Internet Telephone Service Provider）∙ITU：国际电信联盟（International Telecommunication Union）∙ITU-T：国际电信联盟电信标准局（International Telecommunication Union Telecommunication Standardization Sector）∙IUA：ISDN Q.921用户适配层（ISDN Q.921-User Adaptation Layer）∙IVR：交互语音回应（Interactive Voice Response）∙IXC：长途交换运营商（Inter-Exchange Carrier）TOP▲J∙插座（Jack）∙跳线（Jumper）∙汇接点（Junction）∙连接机（Junctor）∙连接机干线群（Junctor Trunk Group）∙权限（Jurisdiction）TOP▲K∙按键脉冲（Key Pulsing）∙控制台线路（Key Station Line）∙电键系统（Key System）∙按键电话机（Key Telephone Set）∙按键电话系统（Key Telephone System）TOP▲L∙（Label）标签∙LAMA：本地自动信息计费（Local Automatic Message Accounting）∙LAPB：链路访问过程平衡（Link Access Procedure, Balanced）∙LAP-D：D信道上的链路接入程序（Link Access Procedure, D-Channel）∙LAPF：数据链路层帧方式接入协议（Link Access Procedure for Frame Mode Services）∙LAP-H：H信道链路访问规程（Link Access Procedure for H-Channel）∙LAP-M：调制解调器链路接入规程（Link Access Procedure for Modems）∙LAPS：SDH 上的链路接入规程（Link Access Procedure-SDH）∙LASS Code：局域信令服务编码（Local Area Signaling Services Code）∙LASS：局域信令服务（Local Area Signaling Service）∙最终选择路径（Last Choice Route）∙最后一英里（Last Mile）∙最后一英里技术（Last Mile Technology）∙本地访问传输区域串联（LATA Tandem）∙LATA：本地访问和传输区域（Local Access And Transport Area）∙LCV：线路编码违规（Line Code Violation）∙LDN：目录表中的号码（Listed Directory Number）∙专用线路（Leased line）∙LEC计费（LEC Billing）∙LEC卡（LEC Card）∙LEC费用（LEC Charges）∙LEC：本地电信运营商（Local Exchange Carrier）∙LERG：本地交换路径指南（Local Exchange Routing Guide）∙LH：线路查找（Line Hunting）∙占线音（Line Busy Tone）∙线路编码（Line Code）∙线路调节（Line Conditioning）∙线路设备（Line Equipment）∙寻线机（Line Finder）∙线路连接（Line Link）∙线路负荷控制（Line Load Control）∙线路噪音（Line Noise）∙线路编码（Line Number）∙线路继电器（Line Relay）∙线段（Line Segment）∙线路信令（Line Signaling）∙线路速率（Line Speed）∙线路转向（Line Turnaround）∙线性失真（Linear Distortion）∙逐段链路信令（Link-by-Link Signaling）∙LLU：本地环路开放（Local Loop Unbundling）∙LNP：本地号码转移（Local Number Portability）∙负荷（Load）∙负荷平衡（Load Balancing）∙感应加热线圈（Load Coil）∙负载线路（Loaded Lines）∙本地连接里程（Local Access Mileage）∙本地通道（Local Channel）∙本地交换回路（Local Exchange Loop）∙本地回路（Local Loop）∙持久通讯（Long-haul communications）∙纵向电压（Longitudinal voltage）∙回路（Loop）∙回路长度（Loop Length）∙回路开始信令（Loop Start Signaling）∙回路传输设备（Loop Transmission Facilities）∙回路启动干线（Loop-start Trunk）∙低音（Low Tone）∙低密度奇偶校验检查码（Low-density parity-check code）∙LPC：线性预测编码（Linear Predictive Coding）∙LPF：低通滤波器（Low-pass Filter）∙LRC：纵向冗余校验（Longitudinal Redundancy Check）∙LSSU：链路状态信号单元（Link Status Signal Unit）∙LTB：最后一个中继线忙（Last Trunk Busy）∙LTCCS：最后一个干线CCS（Last Trunk CCS）∙LTE：电路终端设备（Line Terminating Equipment）TOP▲M∙M Plane：管理层（Management Plane）∙M2PA：第二级对等适配层（MTP2 Peer-to-peer user Adaptation MTP）∙M2UA：MTP 第二级用户适配层（MTP2-User Adaptation layer ）∙M3UA：MTP第三层用户适配层（MTP3-User Adaptation layer）∙主反馈线（Main Feeder）∙主要专用交换机（Main PBX）∙手动振铃线路（Manual Ring-Down Line）∙MAP：移动应用部分（Mobile Application Part）∙MAPOS：通过SONET/SDH的多路接入协议（Multiple Access Protocol over SONET/SDH）∙海用电话（Marine Telephone）∙主机（Master）∙最大比率并合器（Maximal-ratio Combiner）∙MCU：多点会议单元（Multipoint Conferencing Unit）∙MCU：多点控制单元（Multipoint Control Unit）∙机械接头（Mechanical Splice）∙居中设置（Mediation Device）∙居中功能（Mediation function）∙MEDR：最大工程数据率（Maximum Engineering Data Rate）∙均步网络（Mesochronous network）∙微型过滤器（Microfilter）∙中间跨接法（Mid-Span）∙模态色散（Modal Dispersion）∙模态损失（Modal Loss）∙调制解调器（Modem）∙调制解调器代用器（Modem Eliminator）∙模块适配器（Modular Adapter）∙模块插孔（Modular Jack）∙MOS：平均意见得分（Mean Opinion Score）∙MSDSL：多速率对称数字环路（Multi-rate Symmetric DSL）∙MSLT Adjustment：最小扫描线时间调整器（Minimum Scan Line Time Adjustment）∙MSLT：最小扫描线时间（Minimum Scan Line Time）∙MSN：多重用户号码（Multiple Subscriber Number）∙MSO：复合业务运营商（Multiple Service Operator）∙MSO：多重系统运营商（Multiple Systems Operator）∙MSTE：复用段终端设备（Multiplex Section Terminating Equipment）∙MSU：信息信号单元（Message Signal Unit）∙MTIE：最大时间间隔误差（Maximum Time Interval Error）∙MTP：讯息转移部分（Message Transfer Part）∙MTP1：信息传输部分第一级（Message Transfer Part Level 1）∙MTP2： Message Transfer Part Level 2 信息传输部分第二级∙MTP3：信息传输部分第三级（Message Transfer Part Level 3）∙MTS：信息通讯服务（Message Telecommunications Service）∙MTS：信息长途通讯业务（Message Toll Service）∙Mu-Law (μ-law)， Mu-Law∙多信道多点分配服务（Multichannel multipoint distribution service）∙多频信令（Multifrequency Signaling）∙多级优先和占先（Multilevel precedence and preemption）∙多级信号（Multi-level Signal）∙多重线路电话（Multiline Telephone）∙多媒体（Multimedia）∙多路存取（Multiple access）∙多重自导向（Multiple Homing）∙多路复用器（Multiplexer）∙多路技术（Multiplexing）∙多点视频会议（Multipoint Videoconference）∙必须传送要求（Must-Carry）∙MVL：多重虚拟线路（Multiple Virtual Line）TOP▲N∙NAA：下一个可使用的电话局（Next Available Agent）∙无保护的呼叫（Naked Call）∙无保护的DSL（Naked DSL）∙NANP：北美电话号码分配方案（North American Numbering Plan）∙国家入网费（National Access Fee）∙国家信息基础设施（National Information Infrastructure）∙NATO音标字母（NATO phonetic alphabet）∙NCAS：非呼叫相关信令（Non-Call Associated Signaling）∙NCP：网络控制端（Network Control Point）∙NCT： Network Control and Timing 网络控制和定时∙NEBS第一级（NEBS Level 1）∙NEBS第二级（NEBS Level 2）∙NEBS第三级（NEBS Level 3）∙NEBS：网络设备构建系统（Network Equipment Building Systems）∙否定字符（Negative-acknowledge character）∙网络融合（Network Convergence）∙网络元素功能快（Network Element Function Block）∙网络指示语（Network Indicator）∙网络管理中心（Network Management Center）∙网络通讯（Network Services）∙NEXT：近端串音（Near-End Crosstalk）∙NGN：下一代网络（Next Generation Network）∙夜间服务（Night Service）∙N-ISDN：窄带ISDN（Narrowband ISDN）∙NNX∙空号音（No Such Number Tone）∙未接来电转移（No-Answer Transfer）∙类似噪音信号（Noise-Like）∙非关联模式信令（Non-associated Mode Signaling）∙非交换线路（Non-switched Line）∙NPA路由（NPA Routing）∙NPA分裂（NPA Split）∙NPA：编号计划范围（Numbering Plan Area）∙NPA-NXX 路由（NPA-NXX Routing）∙NRZ：非归零（Non-Return-to-Zero）∙NRZI：反向非归零（Nonreturn to Zero Inverted）∙NS/EP通讯（NS/EP telecommunications）∙NSP：网络服务供应商（Network Service Provider）∙NT-1：网络终端1（Network Termination 1）∙零调制解调器（Null Modem）∙号码检查音（Number Check Tone）∙NXXTOP▲O。

acoustic

21.1.1 Direct MethodIn this method, both generation and propagation of sound waves are directly computed by solvingthe appropriate fluid dynamics equations. Prediction of sound waves always requirestime-accurate solutions to the governing equations. Furthermore, in most practical applications ofthe direct method, one has to employ governing equations that are capable of modeling viscousand turbulence effects, such as unsteady Navier-Stokes equations (i.e., DNS), RANS equations,and filtered equations used in DES and LES.The direct method is thus computationally difficult and expensive inasmuch as it requires highly accurate numerics, very fine computational meshes all the way to receivers, and acoustically nonreflecting boundary conditions. The computational cost becomes prohibitive when sound is tobe predicted in the far field (e.g., hundreds of chord-lengths in the case of an airfoil). The direct method becomes feasible when receivers are in the near field (e.g., cabin noise). In many such situations involving near-field sound, sounds (or pseudo-sounds for that matter) are predominantly due to local hydrodynamic pressure which can be predicted with a reasonable cost and accuracy.Since sound propagation is directly resolved in this method, one normally needs to solve the compressible form of the governing equations (e.g., compressible RANS equations, compressible form of filtered equations for LES). Only in situations where the flow is low subsonic and the receivers in the near field sense primarily local hydrodynamic pressure fluctuations (i.e., pseudo sound) can incompressible flow formulations be used. But this incompressible treatment will alsonot allow to simulate resonance and feedback phenomena.21.1.2 Integral Method Based on Acoustic AnalogyFor predictions of mid- to far-field noise, the methods based on Lighthill's acoustic analogy [ 209] offer viable alternatives to the direct method. In this approach, the near-field flow obtained from appropriate governing equations such as unsteady RANS equations, DES, or LES are used topredict the sound with the aid of analytically derived integral solutions to wave equations. The acoustic analogy essentially decouples the propagation of sound from its generation, allowing oneto separate the flow solution process from the acoustics analysis.FLUENT offers a method based on the Ffowcs Williams and Hawkings (FW-H) equation and its integral solutions [ 105]. The FW-H formulation adopts the most general form of Lighthill's acoustic analogy, and is capable of predicting sound generated by equivalent acoustic sources suchas monopoles, dipoles, and quadrupoles. FLUENT adopts a time-domain integral formulation wherein time histories of sound pressure, or acoustic signals, at prescribed receiver locations are directly computed by evaluating a few surface integrals.Time-accurate solutions of the flow-field variables, such as pressure, velocity components, and density on source (emission) surfaces, are required to evaluate the surface integrals. Time-accurate solutions can be obtained from unsteady Reynolds-averaged Navier-Stokes (URANS) equations,large eddy simulation (LES), or detached eddy simulation (DES) as appropriate for the flow at hand and the features that you want to capture (e.g., vortex shedding). The source surfaces can be placed not only on impermeable walls, but also on interior (permeable) surfaces, which enables you to account for the contributions from the quadrupoles enclosed by the source surfaces. Both broadband and tonal noise can be predicted depending on the nature of the flow (noise source) being considered, turbulence model employed, and the time scale of the flow resolved in the flow calculation.The FW-H acoustics model in FLUENT allows you to select multiple source surfaces and receivers. It also permits you either to save the source data for a future use, or to carry out an "on the fly'' acoustic calculation simultaneously as the transient flow calculation proceeds, or both. Sound pressure signals thus obtained can be processed using the fast Fourier transform (FFT) and associated postprocessing capabilities to compute and plot such acoustic quantities as the overall sound pressure level (SPL) and power spectra.One important limitation of FLUENT's FW-H model is that it is applicable only to predicting the propagation of sound toward free space. Thus, while the model can be legitimately used to predict far-field noise due to external aerodynamic flows, such as the flows around ground vehicles and aircrafts, it cannot be used for predicting the noise propagation inside ducts or wall-enclosed space.21.1.3 Broadband Noise Source ModelsIn many practical applications involving turbulent flows, noise does not have any distinct tones, and the sound energy is continuously distributed over a broad range of frequencies. In those situations involving broadband noise, statistical turbulence quantities readily computable from RANS equations can be utilized, in conjunction with semi-empirical correlations and Lighthill's acoustic analogy, to shed some light on the source of broadband noise.FLUENT offers several such source models that enable you to quantify the local contribution (per unit surface area or volume) to the total acoustic power generated by the flow. They include the following:∙Proudman's formula∙jet noise source model∙boundary layer noise source model∙source terms in the linearized Euler equations∙source terms in Lilley's equationConsidering that one would ultimately want to come up with some measures to mitigate the noise generated by the flow in question, the source models can be employed to extract useful diagnostics on the noise source to determine which portion of the flow is primarily responsible for the noise generation. Note, however, that these source models do not predict the sound at receivers.Unlike the direct method and the FW-H integral method, the broadband noise source models donot require transient solutions to any governing fluid dynamics equations. All the source models need is what typical RANS models would provide, such as the mean velocity field, turbulentkinetic energy ( ) and the dissipation rate ( ). Therefore, the use of broadband noise source models requires the least computational resources.21.2 Acoustics Model TheoryThis section describes the theoretical background for the Ffowcs Williams and Hawkings model and the broadband noise source models21.2.1 The Ffowcs Williams and Hawkings ModelThe Ffowcs Williams and Hawkings (FW-H) equation is essentially an inhomogeneous wave equation that can be derived by manipulating the continuity equation and the Navier-Stokes equations. The FW-H [ 43, 105] equation can be written as:(21.2-1) where=fluid velocity component in the direction=fluid velocity component normal to the surface=surface velocity components in the direction= surface velocity component normal to the surface= Dirac delta function= Heaviside functionis the sound pressure at the far field (). denotes a mathematicalsurface introduced to "embed'' the exterior flow problem () in an unbounded space,which facilitates the use of generalized function theory and the free-space Green function to obtainthe solution. The surface () corresponds to the source (emission) surface, and can bemade coincident with a body (impermeable) surface or a permeable surface off the body surface.is the unit normal vector pointing toward the exterior region (),is the far-fieldsound speed, andis the Lighthill stress tensor, defined as(21.2-2)is the compressive stress tensor. For a Stokesian fluid, this is given by(21.2-3)The free-stream quantities are denoted by the subscript. The solution to Equation 21.2-1 isobtained using the free-space Green function ( ). The complete solution consists of surface integrals and volume integrals. The surface integrals represent the contributions from monopole and dipole acoustic sources and partially from quadrupole sources, whereas the volumeintegrals represent quadrupole (volume) sources in the region outside the source surface. The contribution of the volume integrals becomes small when the flow is low subsonic and the source surface encloses the source region. In FLUENT , the volume integrals are dropped. Thus, we have(21.2-4) where(21.2-5)(21.2-6) where(21.2-7)(21.2-8)When the integration surface coincides with an impenetrable wall, the two terms on the right in Equation 21.2-4,and , are often referred to as thickness and loading terms, respectively, in light of their physical meanings. The square brackets in Equations 21.2-5 and 21.2-6 denote that the kernels of the integrals are computed at the corresponding retarded times,, defined as follows, given the observer time, , and the distance to the observer, ,(21.2-9)The various subscripted quantities appearing in Equations 21.2-5 and 21.2-6 are the innerproducts of a vector and a unit vector implied by the subscript. For instance,and, whereand denote the unit vectors inthe radiation and wall-normal directions, respectively. The dot over a variable denotes source-time differentiation of that variable.Please note the following remarks regarding the applicability of this integral solution:∙The FW-H formulation in FLUENT can handle rotating surfaces as well as stationary surfaces.∙It is not required that the surface coincide with body surfaces or walls. The formulation permits source surfaces to be permeable, and therefore can be placed in theinterior of the flow.∙When a permeable source surface (either interior or nonconformal sliding interface) is placed at a certain distance off the body surface, the integral solutions given byEquations 21.2-5 and 21.2-6 include the contributions from the quadrupole sourceswithin the region enclosed by the source surface. When using a permeable source surface, the mesh resolution needs to be fine enough to resolve the transient flow structures inside the volume enclosed by the permeable surface.21.2.2 Broadband Noise Source ModelsProudman's FormulaProudman [ 293], using Lighthill's acoustic analogy, derived a formula for acoustic power generated by isotropic turbulence without mean flow. More recently, Lilley [ 210] rederived the formula by accounting for the retarded time difference which was neglected in Proudman's original derivation. Both derivations yield acoustic power due to unit volume of isotropicturbulence (in W/m ) as(21.2-10)where and are the turbulence velocity and length scales, respectively, and is the speed ofsound. in Equation 21.2-10 is a model constant. In terms of and , Equation 21.2-10 canbe rewritten as(21.2-11)(21.2-12)The rescaled constant, , is set to 0.1 in FLUENT based on the calibration of Sarkar and Hussaini [ 316] using direct numerical simulation of isotropic turbulence.FLUENT can also report the acoustic power in dB, which is computed from(21.2-13)whereis the reference acoustic power ( by default).The Proudman's formula gives an approximate measure of the local contribution to total acoustic power per unit volume in a given turbulence field. Proper caution, however, should be taken when interpreting the results in view of the assumptions made in the derivation, such as high Reynolds number, small Mach number, isotropy of turbulence, and zero mean motion.The Jet Noise Source ModelThis source model for axisymmetric jets is based on the works of Goldstein [ 125] who modified the model originally proposed by Ribner [ 305] to better account for anisotropy of turbulence in axisymmetric turbulent jets.In Goldstein's model, the total acoustic power emitted by the unit volume of a turbulent jet is computed from(21.2-14)whereandare the radial and angular coordinates of the receiver location, and isthe directional acoustic intensity per unit volume of a jet defined by(21.2-15)in Equation 21.2-15 is the modified convection factor defined by(21.2-16)(21.2-17)(21.2-18)The remaining parameters are defined as(21.2-19)(21.2-20)(21.2-21)(21.2-22)(21.2-23)(21.2-24)whereand are computed differently depending on the turbulence model chosen for thecomputation. When the RSM is selected, they are computed from the corresponding normal stresses. For all other two-equation turbulence models, they are obtained from(21.2-25)(21.2-26) FLUENTreports the acoustic power both in the dimensional units ( ) and in dB computed from(21.2-27)whereis the reference acoustic power ( by default).The Boundary Layer Noise Source ModelFar-field sound generated by turbulent boundary layer flow over a solid body at low Mach numbers is often of practical interest. The Curle's integral [ 71] based on acoustic analogy can be used to approximate the local contribution from the body surface to the total acoustic power. To that end, one can start with the Curle's integral(21.2-28)wheredenotes the emission time (), and the integration surface.Using this, the sound intensity in the far field can then be approximated by(21.2-29) whereis the correlation area,, andis the angle between andthe wall-normal direction .The total acoustic power emitted from the entire body surface can be computed from(21.2-30)(21.2-31)which can be interpreted as the local contribution per unit surface area of the body surface to the total acoustic power. The mean-square time derivative of the surface pressure and the correlation area are further approximated in terms of turbulent quantities like turbulent kinetic energy, dissipation rate, and wall shear.FLUENT reports the acoustic surface power defined by Equation 21.2-31 both in physical( ) and dB units.Source Terms in the Linearized Euler EquationsThe linearized Euler equations (LEE) can be derived from the Navier-Stokes equations starting from decompositions of the flow variables into mean, turbulent, and acoustic components, and by assuming that the acoustic components are much smaller than the mean and turbulent components. The resulting linearized Euler equations for the acoustic velocity components can be written as(21.2-32)where the subscript " '' refers to the corresponding acoustic components, and the prime superscript refers to the turbulent components.The right side of Equation 21.2-32 can be considered as effective source terms responsible for sound generation. Among them, the first three terms involving turbulence are the maincontributors. The first two terms denoted by are often referred to as "shear-noise" sourceterms, since they involve the mean shear. The third term denoted by is often called the "self-noise" source term, for it involves turbulent velocity components only.The turbulent velocity field needed to compute the LEE source terms is obtained using the method of stochastic noise generation and radiation (SNGR) [ 30]. In this method, the turbulent velocityfield and its derivatives are computed from a sum of Fourier modes.(21.2-33)where,,are the amplitude, phase, and directional (unit) vector of the Fouriermode associated with the wave-number vector Note that the source terms in the LEE arevector quantities, having two or three components depending on the dimension of the problem at hand.Source Terms in Lilley's EquationLilley's equation is a third-order wave equation that can be derived by combining the conservation of mass and momentum of compressible fluids. When the viscous terms are omitted, it can be written in the following form:(21.2-34)where .Lilley's equation can be linearized about the underlying steady flow as(21.2-35)where is the turbulent velocity component.Substituting Equation 21.2-35 into the source term of Equation 21.2-34, we have(21.2-36)The resulting source terms in Equation 21.2-36 are evaluated using the mean velocity field and the turbulent (fluctuating) velocity components synthesized by the SNGR method. As with the LEE source terms, the source terms in Equation 21.2-36 are grouped depending on whether the mean velocity gradients are involved ( shear noise or self noise), and reported separately in FLUENT.21.3 Using the Ffowcs Williams and Hawkings Acoustics ModelThe procedure for computing sound using the FW-H acoustics model in FLUENT consists largely of two steps. In the first step, a time-accurate flow solution is generated, from which time histories of the relevant variables (e.g., pressure, velocity, and density) on the selected source surfaces are obtained. In the second step, sound pressure signals at the user-specified receiver locations are computed using the source data collected during the first step.Note that you can also use the FW-H model for a steady-state simulation in the case where your model has a single rotating reference frame. Here, the loading noise due to the motion of the noise sources is computed using the FW-H integrals (see Equations 21.2-5 and21.2-6), except that the term involving the time derivative of surface pressure ( inEquation 21.2-6) is set to zero.In computing sound pressure using the FW-H integral solution, FLUENT uses a so-called "forward-time projection'' to account for the time delay between the emission time (the time at which the sound is emitted from the source) and the reception time (the time at which the sound arrives at the receiver location). The forward-time projection approach enables you to compute sound at the same time "on the fly'' as the transient flow solution progresses, without having to save the source data.In this section, the procedure for setting up and using the FW-H acoustics model is outlined first, followed by detailed descriptions of each of the steps involved. Remember that only the steps that are pertinent to acoustics modeling are discussed here. For information about the inputs related to other models that you are using in conjunction with the FW-H acoustics model, see the appropriate sections for those models.The general procedure for carrying out an FW-H acoustics calculation in FLUENT is as follows:1. Calculate a converged flow solution. For a transient case, run the transient solution until you obtain a "statistically steady-state" solution as described below.2. Enable the FW-H acoustics model and set the associated model parameters.DefineModelsAcoustics...3. Specify the source surface(s) and choose the options associated with acquisition and saving of the source data. For a steady-state case, specify the rotating surface zone(s) as the source surface(s).4. Specify the receiver location(s).5. Continue the transient solution for a sufficiently long period of time and save the source data (transient cases only).SolveIterate...6. Compute and save the sound pressure signals.SolveAcoustic Signals...7. Postprocess the sound pressure signals.PlotFFT...Before you start the acoustics calculation for a transient case, a FLUENT transient solution should have been run to a point where the transient flow field has become "statisticallysteady''. In practice, this means that the unsteady flow field under consideration, including all the major flow variables, has become fully developed in such a way that its statistics do not change with time. Monitoring the major flow variables at selected points in the domain is helpful for determining if this condition has been met.As discussed earlier, URANS, DES, and LES are all legitimate candidates for transient flow calculations. For stationary source surfaces, the frequency of the aerodynamically generated sound heard at the receivers is largely determined by the time scale or frequency of the underlying flow. Therefore, one way to determine the time-step size for the transient computation is to make it small enough to resolve the smallest characteristic time scale of the flow at hand that can be reproduced by the mesh and turbulence adopted in your model.Once you have obtained a statistically stationary flow-field solution, you are ready to acquire the source data.21.3.1 Enabling the FW-H Acoustics ModelTo enable the FW-H acoustics model, select Ffowcs-Williams & Hawkings in the Acoustics Model panel (Figure 21.3.1).Define Models Acoustics...When you turn on Ffowcs-Williams & Hawkings, the panel will expand to show the relevant fields for user inputs.Figure 21.3.1: The Acoustics Model PanelSetting Model ConstantsUnder Model Constants in the Acoustics Model panel, specify the relevant acoustic parameters and constants used by the model.Far-Field Density (for example, in Equation 21.2-1) is the far-field fluid density.Far-Field Sound Speed (for example, in Equation 21.2-1) is the sound speed in the farfield (= ).Reference Acoustic Pressure (for example, in Equation 28.10-11) is used to calculate the sound pressure level in dB (see Section 28.10.4). The default reference acoustic pressure isPa.Number of Time Steps Per Revolution is available only for steady-state cases that have a single rotating reference frame. Here you will specify the number of equivalent time steps that it will take for the rotating zone to complete one revolution.Number of Revolutions is available only for steady-state cases that have a single rotating reference frame. Here you will specify the number of revolutions that will be simulated in the model.Source Correlation Length is required when sound is to be computed using a 2D flow result. The FW-H integrals will be evaluated over this length in the depth-wise direction using the identical source data.The default values are appropriate for sound propagating in air at atmospheric pressure and temperature.Computing Sound "on the Fly''The FW-H acoustics model in FLUENT allows you to perform simultaneous calculation of the sound pressure signals at the prescribed receivers without having to write the source data to files, which can save a significant amount of disk space on your machine. To enable this "on-the-fly'' calculation of sound, turn on the Compute Acoustic Signals Simultaneously option in the Acoustics Model panel.Because the noise computation takes a negligible percentage of memory and computational time compared to a transient flow calculation, this option can be used by itself or along with the process of source data file export and sound calculation. For the latter, computingsignals "on the fly'' allows you to see when the signals have become statistically steady so you can know when to stop the simulation.When the Compute Acoustic Signals Simultaneously option is enabled, the FLUENT console window will print a message at the end of each time step indicating that the sound pressure signals have been computed (e.g., Computing sound signals at x receiver locations ..., where x is the number of receivers you specified). Enabling this option instructs FLUENT to compute sound pressure signals at the end of each time step, which will slightly increase the computation time.Note that this option is available only when the FW-H acoustics model has been enabled.See below for details about exporting source data without enabling the FW-H model. Writing Source Data FilesAlthough the "on-the-fly'' capability is a convenient feature, you will want to save the source data as well, because the acquisition of source data during a transient flow-field calculation is the most time-consuming part of acoustics computations, and you most likely will not want to discard it. Bysaving the source data, you can always reuse it to compute the sound pressure signals at new or additional receiver locations.To enable saving the source data to files, turn on the Export Acoustic Source Data option in the Acoustics Model panel. Once this option is selected, the relevant source data at all face elements of the selected source surfaces will be written into the files you specify. The source data vary depending on the solver option you have chosen and whether the source surface is a wall or not. Table 21.3.1 shows the flow variables saved as the source data.See Section 21.3.2 for details on how to specify parameters for exporting source data.Exporting Source Data Without Enabling the FW-H ModelYou can also export sound source data for use with SYSNOISE without having to enable the Ffowcs Williams and Hawkings (FW-H) model. You will still need to specify source surfaces (see Section 21.3.2), as .index and .asd files are required by SYSNOISE. In addition, you can choose fluid zones as emission sources if you want to export quadrupole sources. To enable the selection of fluid zones as sources, use thedefine models acoustics export-volumetric-sources?text command and change the selection to yes.SYSNOISE also requires centroid data for source zones that are being exported.For fan noise calculations, once you have specified the source zones in the Acoustic Sources panel and you have selected Export Acoustic Source Data from the Acoustic Model panel, you can export geometry in cylindrical coordinates by using thedefine models acoustics cylindrical-export?text command and changing the selection to yes. By default, FLUENT exports source zones for SYSNOISE in Cartesian coordinates.You can then export the centroid data to a .data file using the following text command:define models acoustics write-centroid-infoSince you will not be using the FW-H model to compute signals, you will not need to specify any acoustic model parameters or receiver locations. Also, you will not be able to turn on the Compute Acoustic Signals Simultaneously option in the Acoustics Model panel, and Acoustic Signals... will not be available in the Solve menu.21.3.2 Specifying Source SurfacesIn the Acoustics Model panel, click the Define Sources... button to open the Acoustic Sources panel (Figure 21.3.2). Here you will specify the source surface(s) to be used in the acoustics calculation and the inputs associated with saving source data to files.Figure 21.3.2: The Acoustic Sources PanelUnder Source Zones, you can select multiple emission (source) surfaces and the surface Type that you can select is not limited to a wall. You can also choose interior surfaces and sliding interfaces (both stationary and rotating) as source surfaces.The ability to choose multiple source surfaces is useful for investigating the contributions from individual source surfaces. The results based on the use of multiple source surfaces are valid as long as there are negligible acoustic interactions among the surfaces. Thus, some caution needs to be taken when selecting multiple source surfaces.In cases where multiple source surfaces are selected, no source surface may enclose any of the other source surfaces. Otherwise, the sound pressure calculated based on the source surfaces will not be accurate, as the contribution from the enclosed (inner) source surfaces is over predicted, since the FW-H model is unable to account for the shading of the sound from the inner source surfaces by the enclosure surface.If you specify any interior surfaces as source surfaces, the interior surface must be generated in advance (e.g., in GAMBIT) in such a way that the two cell zones adjacent to the surface have different cell zone IDs. Furthermore, you must correctly specify which of the two zones is occupied by the quadrupole sources (interior cell zone). This will allow FLUENT to determine the direction in which the sound will propagate. When you first attempt to select a legitimate interior surface (i.e., an interior surface having two different cell zones on both sides) as a source surface, the Interior Cell Zone Selection panel (Figure 21.3.3) will appear. You will then needto select the interior cell zones from the two zones listed under the Interior Cell Zone.Figure 21.3.4 shows an example of an interior source surface.Figure 21.3.3: The Interior Cell Zone Selection PanelLike general interior surfaces, if the source surfaces selected are sliding interfaces, a panel similar to Figure 21.3.3 will appear that will show the two adjacent cell zones and you will be asked to specify the zone which has the sound sources.When a permeable surface (either interior or sliding interface) is chosen as the sourcesurface, other wall surfaces inside the volume enclosed by the permeable surface thatgenerate sound should not be chosen for the acoustics calculation. For example, when。

运放噪声估计

10Hz (a) eni2 (b) Ini2RG2 (c) 4KTRG (d) Eni2 100Hz 1kHz 10kHz 100kHz
The Noise Model
Figure 1 is a typical noise model depicting the noise voltage and noise current sources that are added together in the form of root mean square to give the total equivalent input voltage noise (RMS), therefore:
R1 R3 - + R 2 R G = ------------------- R 1 + R 3
4KTR1 4KTR3
6. On linear scale graph paper enter each of the values for Eni2 versus frequency. In most cases, sufﬁcient accuracy can be obtained simply by joining the points on the graph with straight line segments. For the bandwidth of interest, calculate the area under the curve by adding the areas of trapezoidal segments. This procedure assumes a perfectly square bandpass condition; to allow for the more normal -6dB/octave bandpass skirts, multiply the upper (-3dB) frequency by 1.57 to obtain the effective bandwidth of the circuit, before computing the area. The total area obtained is equivalent to the square of the total input noise over the given bandwidth. 7. Take the square root of the area found above and multiply by the gain (G) of the circuit to ﬁnd the total Output RMS noise. 8. Take the square root of the area found above and multiply by the gain (G) of the circuit to ﬁnd the total Output RMS noise.

hfss里的broadband pulse表达式

hfss里的broadband pulse表达式在HFSS中，使用broadband pulse可以在时域中产生宽频脉冲，这对于仿真和分析高频电路和天线系统中的信号传输非常有用。

broadband pulse表示为一个时间域的函数，可以通过表达式来定义。

首先，我们需要定义脉冲的起始时间、结束时间和持续时间。

这些参数分别表示为t_start、t_end和t_duration。

起始时间和结束时间决定了脉冲的时间长度，而持续时间决定了脉冲的宽度。

接下来，我们可以使用高斯函数来定义脉冲的形状。

高斯函数可以表示为：f(t) = exp(-(t-t_center)^2/(2*sigma^2))其中，t_center是脉冲的中心时间，sigma是高斯函数的标准差。

脉冲的中心时间可以定义为(t_start + t_end)/2，标准差可以根据脉冲的宽度来选择。

然后，我们可以将高斯函数与一个频率响应函数相乘，以定义脉冲的频谱。

频率响应函数表示为：H(f) = exp(-j*2*pi*f*t_shift)其中，f是频率，t_shift是时间延迟。

时间延迟可以帮助我们调整脉冲的相位。

最后，我们可以通过将高斯函数和频率响应函数进行Fourier 变换，从时域转换到频域，并使用逆Fourier变换将其转换回时域。

这样，我们就可以得到用于定义broadband pulse的表达式。

在HFSS中，可以使用VBScript来定义broadband pulse的表达式。

具体代码如下所示：Dim oDefinitionSet oDefinition = oDesign.DefinitionoDefinition.ClearAllSources' Define broadband pulse parametersDim t_start, t_end, t_duration, t_center, sigma, t_shiftt_start = 0 ' Start timet_end = 10e-9 ' End timet_duration = t_end - t_start ' Pulse durationt_center = (t_start + t_end) / 2 ' Pulse center timesigma = t_duration / 6 ' Gaussian standard deviationt_shift = 0 ' Time delay' Define frequency rangeDim f_start, f_endf_start = 1 / t_end ' Start frequencyf_end = 1 / t_start ' End frequency' Define frequency step sizeDim dfdf = (f_end - f_start) / 1000' Calculate frequency pointsDim f, freq, amp, phaseReDim freq(1000)ReDim amp(1000)ReDim phase(1000)For i = 0 to 1000f = f_start + i * df ' Frequencyfreq(i) = f ' Store frequencyamp(i) = Exp(-((f - (f_start + f_end) / 2) / (2 * (f_end - f_start) / 6)) ^ 2) ' Gaussian amplitudephase(i) = -2 * PI * f * t_shift ' Phase shiftNext' Define broadband pulseoDefinition.EditSource "Pulse('broadband_pulse')", t_start, t_end, freq, amp, phase通过上述代码，我们可以在HFSS中定义一个名为broadband_pulse的broadband pulse源，该源具有我们定义的参数。

基于leaky-lms算法对宽带噪声的控制

167
1 介绍
汽车实用技术
3 Leaky-LMS 算法性能分析
最小均方（Least Mean Square LMS）算法是由 Widrow 3.1 收敛性分析和 Hodff 在 1960 年提出来的[1][2]。该算法简单并易于实现，
（5）
对自适应系统具有良好的处理能力，所以被广泛的应用在汽车主动降噪领域。但是传统的 LMS 算法适用于周期短并且较为规律的窄带信号，对于周期长、随机性较强的宽带信号（路噪），系统的收敛速度和稳定性就不能保证了，特别是瞬
属性，该算法的收敛性和稳定性不能同时满足，当步长因子较大时，可以提高收敛速度，但是稳定性变的较差；当步长因子较小时，可以提高稳定性，但是收敛速度变的较慢。特别是对于随机性较强的宽带信号，LMS 算法的瞬态特性会较差。
Leaky-LMS 算法可以解决 LMS 算法的固有缺陷，该算法可以限制瞬态输出信号，并可以同时提高控制系统的稳定性和收敛性。
10.16638/ki.1671-7988.2019.23.058
基于 Leaky-LMS 算法对宽带噪声的控制
高坤 1,2，史晨路 1，吕晓 1，祖炳洁 2
（1.中汽研（天津）汽车工程研究院有限公司，天津 300000；2.石家庄铁道大学，河北石家庄 050043）
摘要：Leaky-LMS 算法是传统 LMS 的改进算法，该算法主要适用于随机性较强的宽带噪声，该算法解决了传统 LMS 算法收敛性、稳定性和瞬态特性的固有缺陷，同时它在系统辨识方面也有很多应用。文章详细介绍了 Leaky-LMS 算法的推导过程，并分析了该算法的收敛性和稳定性。同时通过 MATLAB 仿真，比较了传统 LMS 算法和 Leaky-LMS 算法对宽带噪声的控制效果。仿真结果表明，Leaky-LMS 算法的稳定性、收敛性和瞬态特性要优于传统 LMS 算法，有效的对汽车车内宽带噪声进行控制。关键词：泄露 LMS 算法；收敛性；稳定性；瞬态特性；MATLAB 仿真中图分类号：U462.3 文献标识码：A 文章编号：1671-7988(2019)23-167-03

运算放大器噪声

Noise – 1 噪声 – 1
TIPL 1311 TI Precision Labs – Op Amps
Presented by Ying Zhou Prepared by Art Kay and Ian Williams

Байду номын сангаас
Hello, and welcome to the TI Precision Labs discussing intrinsic op amp noise, part 1.Overall, this video series will show how to predict op amp noise with calculation and simulation, as well how to accurately measure noise. In part 1 we will define intrinsic noise, introduce the different types of noise, and discuss noise spectral density. 大家好，欢迎来到 TI Precision Labs（德州仪器高精度实验室）。本节视频将介绍 op-amp intrinsic noise（运放的固有噪声）的第一部分。在整个噪声系列中，我们会探讨如何通过计算和仿真得到运放的噪声，并学习如何准确地测量噪声。在这第一部分中，我们会给出噪声的定义，介绍不同的噪声，以及探讨噪声频谱密度。

让我们来看看噪声的一些常见的分类。
White Noise (Broadband Noise) 白噪声（宽带噪声）
4

This slide shows the time domain waveform for white noise, also known as broadband noise. The time domain waveform is what you would see if you measured noise with an oscilloscope. Notice that the horizontal axis is 1ms, full scale. Taking the reciprocal of the full-scale time gives a frequency of 1kHz. In general, broadband noise is considered to be in the middle to high frequency range; that is, frequencies greater than 1kHz. In the next slide we’ll consider lower frequency noise sources. Also notice the statistical distribution to the right hand side of the slide. The distribution is Gaussian, with a mean value of 0V and the skirts of the distribution at approximately ±40mV. The distribution indicates that the probability of measuring noise near 0V is high, where as the probability of measuring noise near the skirts of the distribution is relatively low. Later we will see how the distribution can be used to estimate the peak-to-peak value of the noise signal. 这里显示的是 white noise（白噪声）的时域波形，即人们所知的 broadband noise（宽带噪声）。时域波形是你在使用示波器时观察到的波形。注意到横坐标的满量程是 1ms。满量程 1ms 的倒数则就是 1kHz 的频率。一般来说，宽带噪声是指从中频到高频的范围，即大于 1kHz 的频率。之后我们会涉及低频噪声源。在这里，右图显示的是一个统计分布图。这是一个高斯分布，平均值为 0V，

fluent气动噪声算例-Broadband Noise Modeling

Tutorial:Broadband Noise ModelingPurposeThe purpose of this tutorial is to provide guidelines and recommendations for the basic setup and solution procedure for solving an acousticsﬁeld generated from a sedan car using the broadband noise model.The problem is initially solved for steady state,and then the broadband acoustic model is included in the calculation to perform postprocessing.PrerequisitesThis tutorial assumes that you are familiar with the user interface,basic setup and solution procedures in FLUENT.This tutorial does not cover mechanics of using the broadband noise model,but focuses on setting up the problem for a sedan car and performing postprocessing.It also assumes that you have a basic understanding of aeroacoustic physics.If you have not used FLUENT before,it would be helpful toﬁrst review FLUENT6.2User’s Guide and FLUENT6.2Tutorial Guide.Problem DescriptionThe problem involves a sedan car model as shown in Figure1.The car is traveling at70 miles per hour.You will study only the acousticsﬁeld generated by the motion of the car to highlight the noise source on the sedan body,therefore the mirrors and the wheels of the car are ignored.Figure1:The Sedan CarBroadband Noise ModelingPreparation1.Copy the meshﬁle,sedan-acoustics.msh from the inputﬁle into your working di-rectory.2.Start the3D version of FLUENT.Setup and SolutionStep1:Grid1.Read the meshﬁle,sedan-acoustics.msh.File−→Read−→Case...2.Check the grid.Grid−→Check...3.Keep default scale for the grid.Grid−→Scale...4.Display the grid.Display−→Grid...Figure2:Grid DisplayBroadband Noise Modeling Step2:Models1.Keep the default solver settings.Deﬁne−→Models−→Solver...2.Enable the standard k-epsilon turbulence model.Deﬁne−→Models−→Viscous...Step3:MaterialsDeﬁne−→Materials...1.Keep the default selection of air in the Materials panel.Step4:Operating ConditionsDeﬁne−→Operating Conditions...1.Keep the default operating conditions.Step5:Boundary ConditionsDeﬁne−→Boundary Conditions...1.Set the boundary conditions for velocity inlet(inlet).(a)Under Zone,select inlet.The Type will be reported as velocity-inlet.(b)Click Set...to open the Velocity Inlet panel.Broadband Noise Modelingi.Specify a value of31for Velocity Magnitude.ii.Select Intensity and Length Scale in the Turbulence Speciﬁcation Method drop-down list.iii.Specify a value of2and0.35for Turbulence Intensity and Turbulence Length Scale respectively.2.Set the boundary conditions for pressure outlet(outlet)as shown in the panel.3.Keep the default boundary conditions for other walls.Broadband Noise Modeling Step6:Solution1.Retain the default under-relaxation factors and discretization schemes.Solve−→Controls−→Solution...2.Enable the plotting of residuals during the calculation(Figure3).Solve−→Monitors−→Residual...3.Initialize the solution.Solve−→Initialize−→Initialize...(a)Select inlet in the Compute From drop-down list and click Init.4.Write the caseﬁle(sedan.cas.gz).5.Start the calculation by requesting70iterations.Solve−→Iterate...6.Write the dataﬁle(sedan.dat.gz).Broadband Noise ModelingFigure3:Scaled ResidualsStep7:Enable the Broadband Acoustic ModelDeﬁne−→Models−→Acoustics...1.Under Model,select Broadband Noise Sources.(a)Specify a value4e-10for Reference Acoustic Power(w).(b)Set the Number of Realizations to50.Broadband Noise Modeling(c)Retain the default values for the rest of the model constants and click OK toclose the panel.Step8:Postprocessing1.Display theﬁlled contours of Acoustics Power Level(dB)on the surfaces of the sedancar,i.e.,front,rear,and cabinet(Figure4).Display−→Contours...(a)Under Options,select Filled.(b)Select Acoustics...and Acoustic Power Level(dB)from the Contours of drop-downlists.(c)Under Surfaces,select front,rear,and cabinet.(d)Click Display.2.Similarly,display theﬁlled contours of Surface Acoustics Power Level(dB)(Figure5),and Lilley’s Total Noise Source(Figure6)on the surfaces of the sedan car.Broadband Noise ModelingFigure4:Contours of Acoustic Power LevelFigure5:Contours of Surface Acoustics Power LevelBroadband Noise ModelingFigure6:Contours of Lilley’s Total Noise SourceSummaryThis tutorial demonstrated the use of FLUENT’s broadband noise acoustic model to solve an acousticsﬁeld generated from a sedan car.You have learned how to set up the relevant parameters and postprocess the noise signals to highlight the source of noise on the sedan car body.。

ADS1271中文资料

proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could101001k10k100kFrequency(Hz)High−Resolution ModeShorted Input1,048,576Points101001k10k100kFrequency(Hz)PACKAGING INFORMATIONOrderable Device Status (1)Package Type Package Drawing Pins Package Qty Eco Plan (2)Lead/Ball FinishMSL Peak Temp (3)ADS1271IPW ACTIVE TSSOP PW 1694None CU Level-2-240C-1YEAR ADS1271IPWRACTIVETSSOPPW162500NoneCULevel-2-240C-1YEAR(1)The marketing status values are defined as follows:ACTIVE:Product device recommended for new designs.LIFEBUY:TI has announced that the device will be discontinued,and a lifetime-buy period is in effect.NRND:Not recommended for new designs.Device is in production to support existing customers,but TI does not recommend using this part in a new design.PREVIEW:Device has been announced but is not in production.Samples may or may not be available.OBSOLETE:TI has discontinued the production of the device.(2)Eco Plan -May not be currently available -please check /productcontent for the latest availability information and additional product content details.None:Not yet available Lead (Pb-Free).Pb-Free (RoHS):TI's terms "Lead-Free"or "Pb-Free"mean semiconductor products that are compatible with the current RoHS requirements for all 6substances,including the requirement that lead not exceed 0.1%by weight in homogeneous materials.Where designed to be soldered at high temperatures,TI Pb-Free products are suitable for use in specified lead-free processes.Green (RoHS &no Sb/Br):TI defines "Green"to mean "Pb-Free"and in addition,uses package materials that do not contain halogens,including bromine (Br)or antimony (Sb)above 0.1%of total product weight.(3)MSL,Peak Temp.--The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications,and peak solder temperature.Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided.TI bases its knowledge and belief on information provided by third parties,and makes no representation or warranty as to the accuracy of such information.Efforts are underway to better integrate information from third parties.TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysison incoming materials and chemicals.TI and TI suppliers consider certain information to be proprietary,and thus CAS numbers and other limited information may not be available for release.In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s)at issue in this document sold by TI to Customer on an annual basis.PACKAGE OPTION ADDENDUM27-Dec-2004Addendum-Page 1元器件交易网元器件交易网IMPORTANT NOTICETexas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications,enhancements, improvements, and other changes to its products and services at any time and to discontinueany product or service without notice. Customers should obtain the latest relevant information before placingorders and should verify that such information is current and complete. All products are sold subject to TI’s termsand conditions of sale supplied at the time of order acknowledgment.TI warrants performance of its hardware products to the specifications applicable at the time of sale inaccordance with TI’s standard warranty. T esting and other quality control techniques are used to the extent TIdeems necessary to support this warranty. Except where mandated by government requirements, testing of allparameters of each product is not necessarily performed.TI assumes no liability for applications assistance or customer product design. Customers are responsible fortheir products and applications using TI components. T o minimize the risks associated with customer productsand applications, customers should provide adequate design and operating safeguards.TI does not warrant or represent that any license, either express or implied, is granted under any TI patent right,copyright, mask work right, or other TI intellectual property right relating to any combination, machine, or processin which TI products or services are used. Information published by TI regarding third-party products or servicesdoes not constitute a license from TI to use such products or services or a warranty or endorsement thereof.Use of such information may require a license from a third party under the patents or other intellectual propertyof the third party, or a license from TI under the patents or other intellectual property of TI.Reproduction of information in TI data books or data sheets is permissible only if reproduction is withoutalteration and is accompanied by all associated warranties, conditions, limitations, and notices. Reproductionof this information with alteration is an unfair and deceptive business practice. TI is not responsible or liable forsuch altered documentation.Resale of TI products or services with statements different from or beyond the parameters stated by TI for thatproduct or service voids all express and any implied warranties for the associated TI product or service andis an unfair and deceptive business practice. TI is not responsible or liable for any such statements.Following are URLs where you can obtain information on other Texas Instruments products and applicationsolutions:Products ApplicationsAmplifiers Audio /audioData Converters Automotive /automotiveDSP Broadband /broadbandInterface Digital Control /digitalcontrolLogic Military /militaryPower Mgmt Optical Networking /opticalnetworkMicrocontrollers Security /securityTelephony /telephonyVideo & Imaging /videoWireless /wirelessMailing Address:Texas InstrumentsPost Office Box 655303 Dallas, Texas 75265Copyright 2004, Texas Instruments Incorporated。

汽车NVH介绍普及稿

•Mount
•Intake
•Powerplant
•动力总成
Powerplant
•Engine
•Transmission
•Exhaust
•Driveline
PPT文档演模板
汽车NVH介绍普及稿
•发动机噪声源
• 燃烧噪声
活塞载荷气缸盖载荷曲轴轴承载荷
• 机械振动与噪声
曲轴系统突轮轴系统链,齿轮,皮带非燃烧引起的冲击附件
•P/T Sound Quality
•0 Hz
•100 Hz
•250 Hz
•800
Hz
•5000 Hz
PPT文档演模板
汽车NVH介绍普及稿
•源-通道-接受体模型
•源
– 动力系统 –风 – 路面 – 其他
• 通道
– 底盘 – 车身 – 内饰 – 其他
• 接受体
– 耳朵 –手 –脚 – 座椅
•多通道分析
Miscellaneous noise from the engine compartment 200-2000 Hz sound, usually a tone and its harmonics
PPT文档演模板
汽车NVH介绍普及稿
•NVH – 车速 – 发动机转速的关系
•动力系统(P/T) NVH
NVH影响到售后服务约1/5的售后服务与NVH有关
PPT文档演模板
汽车NVH介绍普及稿
决定NVH的因素
•政府法规 •公司的需要和技术能力
PPT文档演模板
•顾客的要求
•竞争车
汽车NVH介绍普及稿
•NVH对不同的声音和振动有特定的描述
•Boom: •Clunk: •Drone, Moan: •Groan: •Growl: •Harshness: •Rattle: •Rumble: •Shake: •Shudder: •Squeal: •Tick and Hash: •Whine:

相噪与抖动的转换

相噪与抖动的转换MT-008TUTORIAL Converting Oscillator Phase Noise to Time Jitterby Walt KesterINTRODUCTIONA low aperture jitter specification of an ADC is critical to achieving high levels of signal-to-noise ratios (SNR). (See References 1, 2, and 3). ADCs are available with aperture jitter specifications as low as 60-fs rms (AD9445 14-bits @ 125 MSPS and AD9446 16-bits @ 100 MSPS). Extremely low jitter sampling clocks must therefore be utilized so that the ADC performance is not degraded, because the total jitter is the root-sum-square of the internal converter aperture jitter and the external sampling clock jitter. However, oscillators used for sampling clock generation are more often specified in terms of phase noise rather than time jitter. The purpose of this discussion is to develop a simple method for converting oscillator phase noise into time jitter.PHASE NOISE DEFINEDFirst, a few definitions are in order. Figure 1 shows a typical output frequency spectrum of a non-ideal oscillator (i.e., one that has jitter in the time domain, corresponding to phase noise in the frequency domain). The spectrum shows the noise power in a 1-Hz bandwidth as a function of frequency. Phase noise is defined as the ratio of the noise in a 1-Hz bandwidth at a specified frequency offset, f m, to the oscillator signal amplitude at frequency f O.o mFigure 1: Oscillator Power Spectrum Due to Phase NoiseThe sampling process is basically a multiplication of the sampling clock and the analog input signal. This is multiplication in the time domain, which is equivalent to convolution in the frequency domain. Therefore, the spectrum of the sampling clock oscillator is convolved with the input and shows up on the FFT output of a pure sinewave input signal (see Figure 2).f of sf oIDEAL SINEWAVEINPUT SAMPLING CLOCK WITH PHASE NOISEFFT OUTPUT FOR IDEAL ADC WITH N →∞(MEASURED FROM DC TO f s /2)SNR = 20log 1012πf o t j f fFigure 2: Effect of Sampling Clock Phase Noise Ideal Digitized SinewaveThe "close-in" phase noise will "smear" the fundamental signal into a number of frequency bins, thereby reducing the overall spectral resolution. The "broadband" phase noise will cause a degradation in the overall SNR as predicted by Eq. 1 (Reference 1 and 2):π=j 10t f 21log 20SNR .Eq. 1It is customary to characterize an oscillator in terms of its single-sideband phase noise as shown in Figure 3, where the phase noise in dBc/Hz is plotted as a function of frequency offset, f m , with the frequency axis on a log scale. Note the actual curve is approximated by a number of regions, each having a slope of 1/f x , where x = 0 corresponds to the "white" phase noise region (slope = 0 dB/decade), and x = 1 corresponds to the "flicker" phase noise region (slope = –20 dB/decade). There are also regions where x = 2, 3, 4, and these regions occur progressively closer to the carrier frequency.PHASE NOISE (dBc/Hz)FREQUENCY OFFSET, f m , (LOG SCALE)Figure 3: Oscillator Phase Noise in dBc/Hz vs. Frequency OffsetNote that the phase noise curve is somewhat analogous to the input voltage noise spectral density of an amplifier. Like amplifier voltage noise, low 1/f corner frequencies are highly desirable in an oscillator.We have seen that oscillators are typically specified in terms of phase noise, but in order to relate phase noise to ADC performance, the phase noise must be converted into jitter. In order to make the graph relevant to modern ADC applications, the oscillator frequency (sampling frequency) is chosen to be 100 MHz for discussion purposes, and a typical graph is shown in Figure 4. Notice that the phase noise curve is approximated by a number of individual line segments, and the end points of each segment are defined by data points.FREQUENCY OFFSET (Hz)PHASENOISE(dBc/Hz)f mFigure 4: Calculating Jitter from Phase NoiseCONVERTING PHASE NOISE TO JITTERThe first step in calculating the equivalent rms jitter is to obtain the integrated phase noise power over the frequency range of interest, i.e., the area of the curve, A. The curve is broken into a number of individual areas (A1, A2, A3, A4), each defined by two data points. Generally speaking, the upper frequency range for the integration should be twice the sampling frequency, assuming there is no filtering between the oscillator and the ADC input. This approximates the bandwidth of the ADC sampling clock input.Selecting the lower frequency for the integration also requires some judgment. In theory, it should be as low as possible to get the true rms jitter. In practice, however, the oscillator specifications generally will not be given for offset frequencies less than 10 Hz, or so—however, this will certainly give accurate enough results in the calculations. A lower frequency of integration of 100 Hz is reasonable in most cases, if that specification is available. Otherwise, use either the 1-kHz or 10-kHz data point.One should also consider that the "close-in" phase noise affects the spectral resolution of the system, while the broadband noise affects the overall system SNR. Probably the wisest approach is to integrate each area separately as explained below and examine the magnitude of the jitter contribution of each area. The low frequency contributions may be negligible compared to the broadband contribution if a crystal oscillator is used. Other types of oscillators may have significant jitter contributions in the low frequency area, and a decision must be made regarding their importance to the overall system frequency resolution.The integration of each individual area yields individual power ratios. The individual power ratios are then summed and converted back into dBc. Once the integrated phase noise power is known, the rms phase jitter in radians is given by the equation (see References 3-7 for further details, derivations, etc.),10/A 102)radians (Jitter Phase RMS ?=,Eq. 2and dividing by 2πf O converts the jitter in radians to jitter in seconds:O10/A f 2102)onds (sec Jitter Phase RMS π?=.Eq. 3It should be noted that computer programs and spreadsheets are available online to perform the integration by segments and calculate the rms jitter, thereby greatly simplifying the process (References 8, 9).Figure 5 shows a sample calculation which assumes only broadband phase noise. The broadband phase noise chosen of –150 dBc/Hz represents a reasonably good signal generator specification, so the jitter number obtained represents a practical situation. The phase noise of –150 dBc/Hz (expressed as a ratio) is multiplied by the bandwidth of integration (200 MHz) to obtain the integrated phase noise of –67 dBc. Note that this multiplication is equivalent to adding thequantity 10 log 10[200 MHz – 0.01 MHz] to the phase noise in dBc/Hz. In practice, the lower frequency limit of 0.01 MHz can be dropped from the calculation, as it does not affect the final result significantly. A total rms jitter of approximately 1 ps is obtained using Eq. 3.10k100k1M10M100M1GFREQUENCY OFFSET (Hz)PHASE NOISE (dBc/Hz)f m–150RMS PHASE JITTER (radians) ≈2?10 = 6.32×10–4radiansA/10RMS JITTER (seconds) =RMS PHASE JITTER (radians)2 πf OA = –150dBc + 10 log 10200×106–0.01×106= –150dBc + 83dB = –67dBc= 1psFigure 5: Sample Jitter Calculation Assuming Broadband Phase NoiseCrystal oscillators generally offer the lowest possible phase noise and jitter, and some examples are shown for comparison in Figure 6. All the oscillators shown have a typical 1/f corner frequency of 20 kHz, and the phase noise therefore represents the white phase noise level. The two Wenzel oscillators are fixed-frequency and represent excellent performance (Reference 9). It is difficult to achieve this level of performance with variable frequency signal generators, as shown by the –150 dBc specification for a relatively high quality generator.Wenzel ULN Series*–174dBc/Hz @ 10kHz+Wenzel Sprinter Series,–165dBc/Hz @ 10kHz+ High Quality Signal Generator –150dBc/Hz @ 10kHz+ z Thermal noise floor of resistive source in a matched system @ +25°C = –174dBm/Hz z 0dBm = 1mW = 632mV p-p into 50Ωz * An oscillator with an output of +13dBm (2.82V p-p) into 50Ωwith a phase noise of –174dBc/Hz has a noise floor of+13dBm –174dBc = –161dBm, 13dB above the thermal noise floor (Wenzel ULN and Sprinter Series Specifications and Pricing Used with Permission of Wenzel Associates)Figure 6: 100-MHz Oscillator Broadband Phase Noise Floor Comparisons (Wenzel ULNand Sprinter Series Specifications and Pricing used with Permission of WenzelAssociates)At this point, it should be noted that there is a theoretical limit to the noise floor of an oscillator determined by the thermal noise of a matched source: –174 dBm/Hz at +25°C. Therefore, an oscillator with a +13-dBm output into 50 Ω (2.82-V p-p) with a phase noise of –174 dBc/Hz has a noise floor of –174 dBc + 13 dBm = –161 dBm. This is the case for the Wenzel ULN series as shown in Figure 6.Figure 7 shows the jitter calculations from the two Wenzel crystal oscillators. In each case, the data points were taken directly for the manufacturer's data sheet. Because of the low 1/f corner frequency, the majority of the jitter is due to the "white" phase noise area. The calculated values of 64 fs (ULN-Series) and 180 fs represent extremely low jitter. For informational purposes, the individual jitter contributions of each area have been labeled separately. The total jitter is the root-sum-square of the individual jitter contributors.1001k 10k 100k 1M 10MPHASENOISE (dBc/Hz)1001k10k100k1M10MPHASENOISE (dBc/Hz)100M100MFigure 7: Jitter Calculations for Low Noise 100-MHz Crystal Oscillators(Phase Noise Data used with Permission of Wenzel Associates)In system designs requiring low jitter sampling clocks, the costs of low noise dedicated crystal oscillators is generally prohibitive. An alternative solution is to use a phase-locked-loop (PLL) in conjunction with a voltage-controlled oscillator to "clean up" a noisy system clock as shown in Figure 8. There are many good references on PLL design (see References 10-13, for example), and we will not pursue that topic further, other than to state that using a narrow bandwidth loop filter in conjunction with a voltage-controlled crystal oscillator (VCXO) typically gives the lowest phase noise. As shown in Figure 8, the PLL tends to reduce the "close-in" phase noise while at the same time, reducing the overall phase noise floor. Further reduction in the white noise floor can be obtained by following the PLL output with an appropriate bandpass filter.f sFigure 8: Using a Phase-Locked Loop (PLL) and BandpassFilter to Condition a Noisy Clock SourceThe effect of enclosing a free-running VCO within a PLL is shown in Figure 9. Notice that the "close-in" phase noise is reduced significantly by the action of the PLL.Figure 9: Phase Noise for a Free-Running VCO and a PLL-Connected VCOAnalog Devices offers a wide portfolio of frequency synthesis products, including DDS systems, N, and fractional-N PLLs. For example, the ADF4360 family are fully integrated PLLs complete with an internal VCO. With a 10-kHz bandwidth loop filter, the phase noise of the ADF4360-1 2.25-GHz PLL is shown in Figure 10, and the line-segment approximation and jitter calculations shown in Figure 11. Note that the rms jitter is only 1.57 ps, even with a non-crystal VCO.PHASE NOISE (dBc/Hz)Figure 10: Phase Noise for ADF4360-1 2.25-GHz PLLwith Loop Filter BW = 10 kHz1001k10k100k1M 10M 100M 1G 4.5GPHASE NOISE (dBc/Hz)FREQUENCY OFFSET (Hz)Figure 11: Line Segment Approximation to ADF4360-1, 2.25-GHzPLL Phase Noise Showing JitterHistorically, PLL design relied heavily on textbooks and application notes to assist in the design of the loop filter, etc. Now, with Analog Devices free downloadable ADIsimPLL? software, PLL design is much easier. To start, choose a circuit by entering the desired output frequency range, and select a PLL, VCO, and a crystal reference. Once the loop filter configuration has been selected, the circuit can be analyzed and optimized for phase noise, phase margin, gain, spur levels, lock time, etc., in both the frequency and time domain. The program also performs the rms jitter calculation based on the PLL phase noise, thereby allowing the evaluation of the final PLL output as a sampling clock.SUMMARYSampling clock jitter can be disastrous to the SNR performance of high performance ADCs. terms of their phase noise. This article has shown how to convert phase noise into jitter so that the SNR degradation can be easily calculated.Although not as good as relatively expensive stand alone crystal oscillators, modern PLLs using crystal VCOs (along with suitable filtering) can achieve jitter performance suitable for all but the most demanding requirements.The entire problem of clock distribution has become much more critical because of low jitter requirements. Analog Devices is now offering a line of clock distribution ICs to serve these needs (/doc/17197931eefdc8d376ee3245.html /clocks).REFERENCES1.Brad Brannon, "Aperture Uncertainty and ADC System Performance," Application Note AN-501, AnalogDevices, download at /doc/17197931eefdc8d376ee3245.html .2.Bar-Giora Goldberg, "The Effects of Clock Jitter on Data Conversion Devices," RF Design, August 2002,pp. 26-32, /doc/17197931eefdc8d376ee3245.html .3.Ulrich L. Rohde, Digital PLL Frequency Synthesizers, Theory and Design, Prentice-Hall, 1983, ISBN 0-13-214239-2, all of Chapter 2 and pp. 411-418 for computer analysis.4.Joseph V. Adler, "Clock-Source Jitter: A Clear Understanding Aids Oscillator Selection," EDN, February18, 1999, pp. 79-86, /doc/17197931eefdc8d376ee3245.html .5.Neil Roberts, "Phase Noise and Jitter – A Primer for Digital Designers," EEdesign, July 14, 2003,/doc/17197931eefdc8d376ee3245.html .6.Boris Drakhlis, "Calculate Oscillator Jitter by using Phase-Noise Analysis Part 1," Microwaves and RF,January 2001, p. 82, /doc/17197931eefdc8d376ee3245.html .7.Boris Drakhlis, "Calculate Oscillator Jitter by using Phase-Noise Analysis Part 2," Microwaves and RF,February 2001, p. 109, /doc/17197931eefdc8d376ee3245.html .8.Raltron Electronics Corporation, 10651 Northwest 19th Street, Miami, Florida 33172, Tel: (305) 593-6033, /doc/17197931eefdc8d376ee3245.html . (see "Convert SSB Phase Noise to Jitter" under "Engineering Design Tools").9.Wenzel Associates, Inc., 2215 Kramer Lane, Austin, Texas 78758, Tel: (512) 835-2038,/doc/17197931eefdc8d376ee3245.html (see "Allan Variance from Phase Noise" under "Spreadsheets").10.Mike Curtin and Paul O'Brien, "Phase-Locked Loops for High-Frequency Receivers and Transmitters, Part1, Analog Dialogue 33-3, 1999, /doc/17197931eefdc8d376ee3245.html .11.Mike Curtin and Paul O'Brien, "Phase-Locked Loops for High-Frequency Receivers and Transmitters, Part2, Analog Dialogue 33-5, 1999, /doc/17197931eefdc8d376ee3245.html .12.R. E. Best, Phase-Locked Loops: Theory, Design and Applications, Fourth Edition, McGraw-Hill, 1999,ISBN 0071349030.13. F. M. Gardner, Phaselock Techniques, Second Edition, John Wiley, 1979, ISBN 0471042943.Copyright 2009, Analog Devices, Inc. All rights reserved. Analog Devices assumes no responsibility for customer product design or the use or application of customers’ products or for any infringements of patents or rights of others which may result from Analog Devices assistance. All trademarks and logos are property of their respective holders. Information furnished by Analog Devices applications and development tools engineers is believed to be accurate and reliable, however no responsibility is assumed by Analog Devices regarding technical accuracy and topicality of the content provided in Analog Devices Tutorials.。

GLOBAL POSITIONING SYSTEM USING BROADBAND NOISE RE

专利名称：GLOBAL POSITIONING SYSTEM USINGBROADBAND NOISE REDUCTION发明人：Paul L. FEINTUCH,Ronald A.BORRELL,Anthony SAGLEMBENI,Joshua G.SLATER申请号：US11839616申请日：20070816公开号：US20080211715A1公开日：20080904专利内容由知识产权出版社提供专利附图：摘要：The present invention is directed to a global positioning system (GPS) havingimproved signal-to-noise ratio for reducing the required signal level for which GPS may be implemented in electronic devices or cell phones. The acquisition function of the GPS receiver is improved by inserting new signal processing for broadband noise reduction that allows subsequent improved estimation of critical time shift and frequency shift parameters needed for GPS acquisition at lower received signal levels. The decoding of the navigation bits from the satellite transmissions is improved to provide ephemeris data needed for computing navigation solutions at lower received signal levels, by examining the output of the same new signal processing for broadband noise reduction for spikes in the power of the output time series that correspond to navigation bit flips.申请人：Paul L. FEINTUCH,Ronald A. BORRELL,Anthony SAGLEMBENI,Joshua G. SLATER 地址：San Dimas CA US,Corona CA US,Ladera Ranch CA US,Orange CA US国籍：US,US,US,US更多信息请下载全文后查看。

ULTRA-BROADBAND LOW-NOISE GAIN-FLATTENED RARE-EART

专利名称：ULTRA-BROADBAND LOW-NOISE GAIN-FLATTENED RARE-EARTH-DOPED FIBREAMPLIFIER发明人：FLEMING, Simon, Charles申请号：EP99937996.9申请日：19990303公开号：EP1060420A1公开日：20001220专利内容由知识产权出版社提供摘要：PROBLEM TO BE SOLVED: To easily raise the cooling capacity to rolls by arranging a refrigerator which can cool a cooling liquid used for cooling the roll to the room temp. or lower. SOLUTION: Rotary joints 8 connectable with cooling liquid flowing passages 7 in the inner part of the roll 1, is fitted to the shaft end parts of one pair of horizontal and parallel rolls 1 and the refrigenator 6 is connected with the rotary joints through a cooling liquid supplying passage 9 and a cooling liquid draining passage 10. Further, a thermo-meter 11, such as a radiation thermo-meter, toward the surface of the roll 1 and a thermo-meter 12, such as a thermocouple, at the outlet part of the refrigerator 6 in the cooling liquid supplying passage 9, are fitted and a control device 16 which transmits a temp. adjusting signal 15 for adjusting the cooling capacity to the refrigerator 6 by inputting the temp. detecting signal 13, 14 from at least one side of the thermo-meter 11, 12 is arranged. The cooling liquid cooled in the refrigerator 6 is carried into the flowing passage 7 through the supplying passage 9 and the rotary joint 8 to cool the roll 1. The cooling liquid whose temp. rises by cooling the roll 1, is returned back to the refrigerator 6 through the rotary joint 8 and the draining passage 10 and cooled toroom temp. or lower.申请人：THE UNIVERSITY OF SYDNEY地址：Parramatta Road Sydney,New South Wales 2006 AU 国籍：AU代理机构：Kuhnen, Rainer Andreas, Dipl.-Ing.更多信息请下载全文后查看。

1、下载文档前请自行甄别文档内容的完整性，平台不提供额外的编辑、内容补充、找答案等附加服务。
2、"仅部分预览"的文档,不可在线预览部分如存在完整性等问题,可反馈申请退款(可完整预览的文档不适用该条件!)。
3、如文档侵犯您的权益，请联系客服反馈,我们会尽快为您处理(人工客服工作时间：9:00-18:30)。