LM170E01中文资料
LM317中文资料大全
LM317中文资料大全LM117/LM317简介LM117/LM317是美国国家半导体公司的三端可调正稳压器集成电路。
我国和世界各大集成电路生产商均有同类产品可供选用，是使用极为广泛的一类串联集成稳压器。
LM117/LM317 的输出电压范围是1.2V 至37V，负载电流最大为1.5A。
它的使用非常简单，仅需两个外接电阻来设置输出电压。
此外它的线性调整率和负载调整率也比标准的固定稳压器好。
LM117/LM317 内置有过载保护、安全区保护等多种保护电路。
通常LM117/LM317 不需要外接电容，除非输入滤波电容到LM117/LM317 输入端的连线超过6 英寸（约15 厘米）。
使用输出电容能改变瞬态响应。
调整端使用滤波电容能得到比标准三端稳压器高的多的纹波抑制比。
LM117/LM317 能够有许多特殊的用法。
比如把调整端悬浮到一个较高的电压上，可以用来调节高达数百伏的电压，只要输入输出压差不超过LM117/LM317 的极限就行。
当然还要避免输出端短路。
还可以把调整端接到一个可编程电压上，实现可编程的电源输出。
LM117负电压输出LM317正电压输出LM317特性简介可调整输出电压低到1.2V保证1.5A 输出电流典型线性调整率0.01%页脚内容1典型负载调整率0.1%80dB纹波抑制比输出短路保护过流、过热保护调整管安全工作区保护标准三端晶体管封装电压范围LM117/LM317 1.25V 至37V 连续可调图1 LM317典型应用电路页脚内容2图2 LM317外形引脚图片D2PAK封典型的TO-3封装TO-220封装ISOWATT220封装装LM117LM117KLM217 LM217K LM217T LM217D2T页脚内容3LM317电气参数：页脚内容4页脚内容5页脚内容6SVR电源电压抑制Tj=25℃f=120HzCADJ =0 6665 -80 -dBCADJ=10uF80 -6680 -dBLM317如何应用计算图3决定LM317输出电压的是电阻R1,R2的比值,假设R2是一个固定电阻.因为输出端的电位高,电流经R1, R 2流入接地点. LM317的控制端消耗非常少的电流,可忽略不计.所以, 控制端的电位是I x R2,又因为LM317控制端, 输出端接脚间的电位差为1.25 V,所以Out(输出）的电压是:接下来,计算I: out与adj接脚间的电位差为1.25 V,电阻R1.电流I是: 1.25/R1。
LM331中文资料_中文手册_芯片中文资料_芯片中文手册
电压-频率变换器LM331LM331是美国NS公司生产的性能价格比较高的集成芯片。
LM331可用作精密的频率电压（F/V）转换器、A/D转换器、线性频率调制解调、长时间积分器以及其他相关的器件。
LM331为双列直插式8脚芯片，其引脚如图3所示。
LM331内部有（1）输入比较电路、（2）定时比较电路、（3）R-S触发电路、（4）复零晶体管、（5）输出驱动管、（6）能隙基准电路、（7）精密电流源电路、（8）电流开关、（9）输出保护点路等部分。
输出管采用集电极开路形式，因此可以通过选择逻辑电流和外接电阻，灵活改变输出脉冲的逻辑电平，从而适应TTL、DTL和CMOS 等不同的逻辑电路。
此外，LM331可采用单/双电源供电，电压范围为4～40V，输出也高达40V。
引脚1（PIN1）为电流源输出端，在f０（PIN3）输出逻辑低电平时，电流源ＩＲ输出对电容ＣＬ充电。
引脚2（PIN2）为增益调整，改变ＲＳ的值可调节电路转换增益的大小。
引脚3（PIN3）为频率输出端，为逻辑低电平，脉冲宽度由Ｒt和Ｃt决定。
引脚4（PIN4）为电源地。
引脚5（PIN5）为定时比较器正相输入端。
引脚6（PIN6）为输入比较器反相输入端。
引脚7（PIN7）为输入比较器正相输入端。
引脚8（PIN8）为电源正端。
LM331频率电压转换器V/F变换和F/V变换采用集成块LM331，LM331是美国NS公司生产的性能价格比较高的集成芯片，可用作精密频率电压转换器用。
LM331采用了新的温度补偿能隙基准电路，在整个工作温度范围内和低到4.0V电源电压下都有极高的精度。
同时它动态范围宽，可达100dB；线性度好，最大非线性失真小于0.01％，工作频率低到0.1Hz时尚有较好的线性；变换精度高，数字分辨率可达12位；外接电路简单，只需接入几个外部元件就可方便构成V/F或F/V等变换电路，并且容易保证转换精度。
图2是由LM331组成的电压频率变换电路，LM331内部由输入比较器、定时比较器、R－S触发器、输出驱动、复零晶体管、能隙基准电路和电流开关等部分组成。
派克液压中文样本
液压注意 – 用户方责任 错误或不当地选择或使用本样本或有关资料阐述的产品，可能会导致人生伤亡及财产损失！ 本样本以及其它由派克汉尼汾公司及其子公司、销售公司与授权分销商所提供的资料，仅供用户专业技术人员在对产品和系统的选型进行深入调查考证时参考。
用户应全面分析自身设备的运行工况、适用的工业标准，并仔细查阅现行的样本，以详细地了解产品及系统的相关信息，通过自己的分析和试验，对产品及系统的独立的最终选择负责，确保能满足自身设备的所有性能、耐用性、维修型、安全性以及预警功能等要求。
对于派克或其子公司或授权分销商而言，应负责按用户提供的技术资料和规范，选择和提供适当的元件或系统，而用户则应负责确定这些技术资料和规范对其设备的所有运行工况和能合理预见的使用工况是否充分和准确。
目录目录页次概述 1 订货代号 2 技术参数 4 变量控制器 5 控制选项 “C”, 压力限定(恒压)变量控制器 5 控制选项 “L”, 负载传感及压力限定变量控制器 6 控制选项 “AM”, 带遥控口的标准型先导式压力限定变量控制器 7 控制选项 “AN”, 带ISO 4401 NG06先导阀安装界面的先导式压力限定变量控制器 8 控制选项 “AE”及“AF”, 带电磁比例调节的先导式压力限定变量控制器 9 控制选项 “AMT”, “ALT”及“LOT”, 带最高压力限定的扭矩限定(恒功率)变量控制器 10 P1性能特性 11典型流量特性 11 典型总效率特性 13 典型轴输入功率特性 15 典型噪声特性 18 典型轴承寿命 20 PD性能特性 22典型流量特性 22 典型总效率特性 24 典型轴输入功率特性 26 典型噪声特性 29 典型轴承寿命 31 安装尺寸 33 P1/PD 018 33 P1/PD 028 36 P1/PD 045 40 P1/PD 060 44 P1/PD 075 49 P1/PD 100 54 P1/PD 140 59 变量控制器安装尺寸 65 可提供的扩展的液压产品 75派克汉尼汾备记派克汉尼汾概述简介, 优点派克汉尼汾简介 • 开式回路用轴向柱塞式变量液压泵 • 中压，连续工作压力280 bar • 高驱动转速型，适用于行走机械； 低噪声型，适用于工业应用 • 静音及高效的控制效能 优点 • 总结构尺寸紧凑 • 低噪声• 流量脉动小，进一步降低噪声• 采用弹性密封，不使用密封垫，从而避免外泄漏的产生• 总效率高，功耗小，减小发热• 采用带无泄漏调节装的简单变量控制器 • 符合SAE 及ISO 标准的安装法兰及油口 • 采用圆锥滚柱轴承，使用寿命长 • 全功率后驱动能力• 后部或侧面油口配置可选• 泄油口的配置对水平安装及驱动轴向上垂直安装均适用• 带有最大及最小排量调节选项 • 具有壳体至吸口单向阀选项，可延长轴封寿命 • 使用、维修方便 脉动容腔技术下列图表所示为侧向油口配置P1/PD 18, 28及45泵采用 “脉动容腔” 技术的效果，脉动容腔可降低泵出口处的压力脉动幅值40-60%，这样，无需增加成本来加装噪声缓冲元件，便可大大降低液压系统的整体噪声，P1系列 PD 系列出口压力p / bar平均压力脉动 / b a rP1 045出口压力脉动2600 rpm 无脉动容腔2600 rpm 带脉动容腔订货代号18 ml, 28 ml, 45ml派克汉尼汾P 类型 01 驱动轴 转向R 5密封材料E 油口配置0 壳体-吸口 单向阀 0 排量调节 018 排量 S 安装法兰 及油口 S 轴封 M 应用范围A 设计系列0 通轴驱动选项 C0控制选项0附加控制选项 00油漆 00修改代号系列 P D * 仅适用于045排量, “S”型安装法兰及油口00 标准型, 无修改M2 按要求修改 代号修改代号 * 适用于028及045排量 ** 仅适用于045排量 代号设计系列 A 现行设计系列5 氟碳橡胶 (FPM) 代号密封材料 A 82-2 SAE A M33x2 M27x2 BSPP 1/4”, 3/8” 101-2 SAE B M42x2 M27x2 BSPP 1/4”, 1/2” 101-2SAE B M48x2M33x2Ø38/25DN51/25BSPP 1/4”, 1/2”B ISO M33x2 M27x2 BSPP 1/4”,3/8”ISO M42x2 M27x2 BSPP 1/4”, 1/2” ISO M48x2M33x2Ø38/25DN51/25BSPP 1/4”, 1/2”代号 018排量 028排量 045排量 安装法兰及油口 安装 法兰 螺纹 油口 辅助 油口 安装 法兰 螺纹 油口 辅助 油口 安装法兰螺纹油口法兰 油口辅助 油口 S 82-2 SAE A SAE 16/12 SAE 4/6 101-2 SAE B SAE 20/12 SAE 4/8 101-2SAE B SAE 24/16Ø38/2561系列SAE 4/10M ISO M33x2 M27x2 M12x1.5 M16x1.5 ISO M42x2 M27x2 M12x1.5 M22x1.5 ISO M48x2M33x2Ø38/25DN51/25M12x1.5M22x1.5代号 018驱动轴 028驱动轴 045驱动轴 01 SAE A 11T 花键SAE B-B 15T 花键 SAE B-B 15T 花键02 SAE 19-1平键Ø0.75” SAE B-B 平键Ø1” SAE B-B 平键Ø1” 08— SAE B 13T 花键 SAE B 13T 花键 04 ISO/DIN 平键, Ø20ISO/DIN 平键, Ø25ISO/DIN 平键, Ø25 06 SAE A 9T 花键— — PD 工业液压用 代号 系列P1 行走机械用 代号 排量 018 18 ml/rev (1.10 in 3/rev) 028 28 ml/rev (1.71 in 3/rev) 045 45 ml/rev (2.75 in 3/rev) 代号 类型 P 开式回路用变量柱塞泵 U*通用 代号应用范围 S 工业液压 (PD) M 行走机械 (P1) R 顺时针 (右转)L 逆时针 (左转)代号 转向 代号 轴封 S 单唇轴封 * 并不具有控制功能，仅在运输时予以防护，详情见第7页的控制说明。
LME49710_07中文资料
Key Specifications
■ Power Supply Voltage Range
±2.5V to ±17V
■ THD+N (AV = 1, VOUT = 3VRMS, fIN = 1kHz)
RL = 2kΩ
0.00003% (typ)
RL = 600Ω
0.00003% (typ)
■ Input Noise Density
2.5nV/√Hz (typ)
■ Slew Rate
±20V/μs (typ)
■ Gain Bandwidth Product
55MHz (typ)
■ Open Loop Gain (RL = 600Ω)
140dB (typ)
■ Input Bias Current
7nA (typ)
■ Input Offset Voltage
Metal Can
20210402
Order Number LME49710HA See NS Package Number — H08C
20210405
3
元器件交易网
LME49710
Absolute Maximum Ratings (Notes 1, 2)
0.05mV (typ)
■ DC Gain Linearity Error
0.000009%
Features
■ Easily drives 600Ω loads ■ Optimized for superior audio signal fidelity ■ Output short circuit protection ■ PSRR and CMRR exceed 120dB (typ) ■ SOIC, DIP, TO-99 metal can packages
LM1117_中文PDF资料
1A LDO稳压器电路概述是一款正电压输出的低压降三端线性稳压电路，在1A输出电流下的压降为1.2V。
分为两个版本，固定电压输出版本和可调电压输出版本。
固定输出电压1.5V、1.8V、2.5V、3.3V、5.0V和可调版本的电压精度为1％；固定电压为1.2V的产品输出电压精度为2％。
内部集成过热保护和限流电路，适用于各类电子产品。
特点*固定输出电压为1.5V、1.8V、2.5V、3.3V、5.0V和可调版本的电压精度为1％*固定电压为1.2V的输出电压精度为2％*低漏失电压：1A输出电流时仅为 1.2V*限流功能*过热切断*温度范围：-40°C~ 125°C应用*膝上型电脑，掌上电脑和笔记本电脑*电池充电器* SCSI-II主动终端*移动电话*无绳电话*电池供电系统*便携式设备* SMPS波斯特稳压器产品规格分类（温度范围：-40°C~ 125°C）X1117X1117X1117X1117X1117换页UX产品名称封装打印名称材料包装X 1117H - ADJ无铅编带X 1117H- 1.8TRX 1117H - 1.8编带X 1117H- 3.3TRX 1117H - 3.3编带SOT-223-3LX 1117H- ADJTR编带编带编带无铅无铅X 1117H- ADJTRX 1117H- 1.8TRX 1117H- 3.3TRX 1117H - ADJX 1117H - 3.3X 1117H - 1.8无卤无卤无卤内部框图3带隙限流TSD21AVOUTADJ/GNDVIN固定版本:F1和F2连接A断开可调版本: A连接， F1和 F2断开F1F2X1117换页极限参数参数符号范围单位输入工作电压引脚温度 (焊接5秒)TLead260°C工作结温范围TJ150°C储存温度TSTG-65 ~ +150°C功耗PD内部限制 (注1)mWESD能力 (最小值)ESD2000V注1：最大允许功耗是最大工作结温TJ (max)，结对空热阻θJA和环境温度Tamb的函数。
LM1117-1.2中文资料
0≤ IOUT≤1A, 6.5V≤VIN≤12V
4.900 5.000 5.10
TSOUT
Rline
VINMIN ≤VIN≤ 12V, VOUT=Fixed/Adj,
IOUT=10mA
0.3
%
3
7 mV
Rload Vdrop
Iq
10mA≤IOUT≤ 1A,VOUT=Fixed/Adj IOUT=100mA IOUT=500mA IOUT=1A 4.25V≤VIN≤ 6.5V
推荐工作条件
参数 输入电压 工作结温范围
符号 VIN TJ
范围 15
-40 ~ +125
单位 V °C
电气特性(除非特别指定，否则黑色字体所示的参数，Tamb=25°C，正常工作结温范围 -40°C ~125°C。)
参数 基准电压
输出电压
符号
测试条件
最小值 典型值 最大值 单位
AMS1117-ADJ,
0.5
%
Tamb=125°C, 1000Hrs
0. 3
%
% of VOUT, 10Hz≤f≤10kHz
0.003
%
SOT-223-3
120
TO-252-2
100
θJA TO-263-3
60
°C/W
SOT-89-3
165
TO-220-3
60
ADVANCED MONOLITHIC SYSTEMS (translate by BONA 0755-82800289)
88029102910ams111730iout10mavin45vtj25c0iout1a44vvin10v297029403000300030303060ams111733iout10mavin5vtj25c0iout1a475vvin10v326732353300330033333365输出电压voutams111750iout10mavin7vtj25ciout1a65vvin12v4950490050005000505510输出电压温度稳定性tsout03线性调整rlinevinminvin12vvoutfixedadjiout10mamv负载调整rload10maiout1avoutfixedadj12mv漏失电压vdropiout100maiout500maiout1a100105110120125130静态电流iq425vvin65v10ma纹波抑制比psrrfripple120hzvinvout3vvripple1vpp6075db可调管脚电流iadj60120iout1a14vvinvout10v02温度稳定性05长期稳定性tamb125c1000hrsvout10hzf10khz0003sot2233120to2522100to263360sot893165热阻系数无散热片jato220360advancedmonolithicsystemswwwamssemitechcombona075582800289ams1117管脚排列图管脚描述管脚号管脚名称输入工作电压
LM1117 中文PDF资料
Vdrop IOUT=500mA
1.05 1.25 V
IOUT=1A
1.10 1.30
Iq 4.25V≤VIN≤ 6.5V
5
10 mA
fRIPPLE=120Hz, (VIN-VOUT)=3V,
PSRR
60 75
dB
VRIPPLE=1VPP
Iadj
60 120 µA
0≤ IOUT≤1A, 1.4V ≤VIN-VOUT≤10V
单位:毫米
SOT-89-3L
单位:毫米
封装外形图
TO-220-3L
X1117
单位:毫米
TO-263-3L
单位:毫米
封装外形图
TO-252-2L
X1117
单位:毫米
6.50±0.25 5.30±0.20
2.30±0.10 0.5±0.1
0.2
5 µA
TSD
150
°C
Ilimit
1.2 1.4 1.5 A
参数 温度稳定性 长期稳定性 RMS输出噪声
热阻系数 (无散热片)
管脚排列图
符号
测试条件
Tamb=125°C, 1000Hrs % of VOUT, 10Hz≤f≤10kHz SOT-223-3L TO-252-2L θJA TO-263-3L SOT-89-3L TO-220-3L
1.2 1.224 V 1.2 1.248
X1117 -1.5,
IOUT=10mΑ, VIN=3.5V ,TJ=25°C 10mA≤IOUT≤1A, 3.0V≤VIN≤10V
1.485 1.500 1.515 V 1.470 1.500 1.530
X1117-1.8,
LM4701资料
LM4701Overture ™Audio Power Amplifier Series 30W Audio Power Amplifier with Mute and Standby ModesGeneral DescriptionThe LM4701is an audio power amplifier capable of deliver-ing typically 30W of continuous average output power into an 8Ωload with less than 0.1%(THD +N).The LM4701has an independent smooth transition fade-in/out mute and a power conserving standby mode which can be controlled by external logic.The performance of the LM4701,utilizing its Self Peak In-stantaneous Temperature (˚Ke)(SPiKe ™)Protection Cir-cuitry,places it in a class above discrete and hybrid amplifi-ers by providing an inherently,dynamically protected Safe Operating Area (SOA).SPiKe Protection means that these parts are completely safeguarded at the output against over-voltage,undervoltage,overloads,including thermal runaway and instantaneous temperature peaks.Key Specificationsn THD+N at 1kHz at continuous average output power of 25W into 8Ω:0.1%(max)n THD+N from 20Hz to 20kHz at 30W of continuous average output power into 8Ω:0.08%(typ)n Standby current: 2.1mA (typ)Featuresn SPiKe Protectionn Minimal amount of external components necessary n Quiet fade-in/out mute function n Power conserving standby-mode nNon-Isolated 9-lead TO-220packageApplicationsn TVsn Component stereo n Compact stereoTypical Application Connection DiagramSPiKe ™Protection and Overture ™are trademarks of National Semiconductor Corporation.DS100835-1*Optional components dependent upon specific design requirements.Refer to the External Components Description section for a component functional description.FIGURE 1.Typical Audio Amplifier Application CircuitPlastic PackageDS100835-2Top ViewOrder Number LM4701T See NS Package Number TA9AFor Staggered Lead Non-Isolated PackageOnly a 9-Pin PackageMarch 1998LM4701Overture Audio Power Amplifier Series 30W Audio Power Amplifier with Mute and Standby Modes©1999National Semiconductor Corporation Absolute Maximum Ratings(Notes5,4) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications.Supply Voltage|V CC|+|V EE|(No Signal)66V Supply Voltage|V CC|+|V EE|(with Input and Load)64V Common Mode Input Voltage(V CC or V EE)and|V CC|+|V EE|≤60V Differential Input Voltage60V Output Current Internally Limited Power Dissipation(Note6)62.5W ESD Susceptibility(Note7)2000V Junction Temperature(Note8)150˚C Thermal ResistanceθJC 1.8˚C/W θJA43˚C/W Soldering InformationTF Package(10sec.)260˚C Storage Temperature−40˚C≤T A≤+150˚C Operating Ratings(Notes4,5)Temperature RangeT MIN≤T A≤T MAX−20˚C≤T A≤+85˚C Supply Voltage|V CC|+|V EE|(Note1)20V to64VElectrical Characteristics(Notes4,5)The following specifications are for V CC=+28V,V EE=−28V with R L=8Ω,unless otherwise specified.Limits ap-ply for T A=25˚C.Symbol Parameter Conditions LM4701Units(Limits)Typical Limit(Note9)(Note10)|V CC|+|V EE|Power Supply Voltage GND−V EE≥9V1820V(min) (Note11)64V(max) P O Output Power THD+N=0.1%(max),f=1kHz(Note3)(Continuous Average)R L=8Ω,|V CC|=|V EE|=28V3025W/ch(min)R L=4Ω,|V CC|=|V EE|=20V(Note13)2215W/ch(min) THD+N Total Harmonic Distortion30W/ch,R L=8Ω,0.08% Plus Noise20Hz≤f≤20kHz,A V=26dBSR(Note3)Slew Rate V IN=1.414Vrms,t rise=2ns1812V/µs(min) I TOTAL Total Quiescent Power V CM=0V,V O=0V,I O=0mA(Note2)Supply Current Standby:Off2540mA(max)Standby:On 2.1mA Standby PinV IL Standby Low Input Voltage Not in Standby Mode0.8V(max) V IH Standby High Input Voltage In Standby Mode 2.0 2.5V(min) Mute PinV IL Mute Low Input Voltage Output Not Muted0.8V(max) V IH Mute High Input Voltage Output Muted 2.0 2.5V(min) A M Mute Attenuation V PIN8=2.5V11580dB(min) V OS(Note2)Input Offset Voltage V CM=0V,I O=0mA 2.015mV(max) I B Input Bias Current V CM=0V,I O=0mA0.20.5µA(max) I OS Input Offset Current V CM=0V,I O=0mA0.0020.2µA(max) I O Output Current Limit|V CC|=|V EE|=10V,t ON=10ms, 3.5 2.9A PK(min)V O=0VV OD Output Dropout Voltage|V CC−V O|,V CC=20V,I O=+100mA 1.8 2.3V(max) (Note2)(Note12)|V O−V EE|,V EE=−20V,I O=−100mA 2.5 3.2V(max) PSRR Power Supply Rejection Ratio V CC=30V to10V,V EE=−30V,11585dB(min) (Note2)V CM=0V,I O=0mAV CC=30V,V EE=−30V to−10V11085dB(min)V CM=0V,I O=0mA2Electrical Characteristics(Continued)(Notes4,5)The following specifications are for V CC=+28V,V EE=−28V with R L=8Ω,unless otherwise specified.Limits ap-ply for T A=25˚C.Symbol Parameter Conditions LM4701Units(Limits)Typical Limit(Note9)(Note10)CMRR(Note 2)Common Mode Rejection Ratio V CC=35V to10V,V EE=−10V to−35V,11080dB(min)V CM=10V to−10V,I O=0mAA VOL(Note2)Open Loop Voltage Gain R L=2kΩ,∆V O=30V11090dB(min) GBWP Gain-Bandwidth Product f O=100kHz,V IN=50mVrms7.55MHz(min) e IN Input Noise IHF—A Weighting Filter 2.08µV(max) (Note3)R IN=600Ω(Input Referred)SNR Signal-to-Noise Ratio P O=1W,A-Weighted,98dBMeasured at1kHz,R S=25ΩP O=25W,A-Weighted108dBMeasured at1kHz,R S=25ΩNote1:Operation is guaranteed up to64V,however,distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations are not taken into account.Refer to the Application Information section for a complete explanation.Note2:DC Electrical Test;Refer to Test Circuit#1.Note3:AC Electrical Test;Refer to Test Circuit#2.Note4:All voltages are measured with respect to the GND(pin7),unless otherwise specified.Note5:Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.Operating Ratings indicate conditions for which the device is func-tional,but do not guarantee specific performance limits.Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guar-antee specific performance limits.This assumes that the device is within the Operating Ratings.Specifications are not guaranteed for parameters where no limit is given,however,the typical value is a good indication of device performance.Note6:For operating at case temperatures above25˚C,the device must be derated based on a150˚C maximum junction temperature and a thermal resistance of θJC=1.8˚C/W(junction to case).Refer to the section,Determining the Correct Heat Sink,in the Application Information section.Note7:Human body model,100pF discharged through a1.5kΩresistor.Note8:The operating junction temperature maximum is150˚C,however,the instantaneous Safe Operating Area temperature is250˚C.Note9:Typicals are measured at25˚C and represent the parametric norm.Note10:Limits are guarantees that all parts are tested in production to meet the stated values.Note11:V EE must have at least−9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled.In addition,the voltage dif-ferential between V CC and V EE must be greater than14V.Note12:The output dropout voltage,V OD,is the supply voltage minus the clipping voltage.Refer to the Clipping Voltage vs.Supply Voltage graph in the Typical Per-formance Characteristics section.Note13:For a4Ωload,and with±20V supplies,the LM4701can deliver typically22Watts of continuous average power per channel with less than0.1%(THD+N).With supplies above±20V,the LM4701cannot deliver more than22watts into4Ωdue to current limiting of the output transistors.Thus,increasing the power supply above±20V will only increase the internal power dissipation,not the possible output power.Increased power dissipation will require a larger heat sink as explained in the Application Information section.3Test Circuit #1(Note 2)(DC Electrical Test Circuit)Test Circuit #2(Note 3)(AC Electrical Test Circuit)Bridged Amplifier Application CircuitDS100835-3DS100835-4DS100835-5FIGURE 2.Bridged Amplifier Application Circuit 4Single Supply Application CircuitAuxillary Amplifier Application CircuitDS100835-6FIGURE 3.Single Supply Amplifier Application CircuitDS100835-7FIGURE 4.Auxillary Amplifier Application Circuit5Equivalent Schematic(Excluding Active Protection Circuitry)DS100835-8 6External Components DescriptionComponents Functonal Description1R B Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the load upon power down of the system due to the low input impedance of the circuitry when theundervoltage circuitry is off.This phenomenon occurs when the supply voltages are below1.5V.2R I Inverting input resistance to provide AC gain in conjunction with R F.Also creates a highpass filter with C I at f C=1/(2πR I C I).3R F Feedback resistance to provide AC gain in conjunction with R I.4C I(Note14)Feedback capacitor which ensures unity gain at DC.5C S Provides power supply filtering and bypassing.Refer to the Supply Bypassing application section for proper placement and selection of bypass capacitors.6R V(Note14)Acts as a volume control by setting the input voltage level.7R IN(Note14)Sets the amplifier’s input terminals DC bias point when C IN is present in the circuit.Also works with C IN to create a highpass filter at f C=1/(2πR IN C IN).Refer to Figure4.8C IN(Note14)Input capacitor which blocks the input signal’s DC offsets from being passed onto the amplifier’s inputs.9R SN(Note14)Works with C SN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The pole is set at f C=1/(2πR SN C SN).Refer to Figure4.10C SN(Note14)Works with R SN to stabilize the output stage by creating a pole that reduces high frequency instabilities.11L(Note14)Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of the series resonant circuit.Also provides a low impedance at low frequencies to shortout R and pass audio signals to the load.Refer to Figure4.12R(Note14)13R A Provides DC voltage biasing for the transistor Q1in single supply operation.14C A Provides bias filtering for single supply operation.15R INP(Note14)Limits the voltage difference between the amplifier’s inputs for single supply operation.Refer to the Clicks and Pops application section for a more detailed explanation of the function of R INP.16R BI Provides input bias current for single supply operation.Refer to the Clicks and Pops application section for a more detailed explanation of the function of R BI.17R E Establishes a fixed DC current for the transistor Q1in single supply operation.This resistor stabilizes the half-supply point along with C A.Note14:Optional components dependent upon specific design requirements.7Typical Performance CharacteristicsTHD +N vs FrequencyDS100835-10THD +N vs FrequencyDS100835-11THD +N vs FrequencyDS100835-12THD +N vs Output Power DS100835-13THD +N vs Output Power DS100835-14THD +N vs Output PowerDS100835-15THD +N vs Output Power DS100835-16THD +N vs Output Power DS100835-17THD +N vs Output PowerDS100835-18Clipping Voltage vs Supply VoltageDS100835-19Clipping Voltage vs Supply VoltageDS100835-20Clipping Voltage vs Supply VoltageDS100835-21 8Typical Performance Characteristics(Continued)Power Dissipation vsOutput PowerDS100835-22Power Dissipation vsOuput PowerDS100835-23Power Dissipation vsOutput PowerDS100835-24Output Power vsLoad ResistanceDS100835-25Output Power vsSupply VoltageDS100835-26Output Mute vsMute Pin VoltageDS100835-27Pulse ResponseDS100835-28Large Signal ResponseDS100835-29Output Mute vsMute Pin VoltageDS100835-30 9Typical Performance Characteristics(Continued)Power Supply Rejection RatioDS100835-31Common-Mode Rejection RatioDS100835-32Open LoopFrequency ResponseDS100835-33Safe Area DS100835-34Spike Protection ResponseDS100835-35Supply Current vs Supply VoltageDS100835-36Pulse Thermal ResistanceDS100835-37Pulse Thermal ResistanceDS100835-38Supply Current vs Output VoltageDS100835-39 10Typical Performance Characteristics(Continued)Application InformationMUTE MODEBy placing a logic-high voltage on the mute pin,the signal going into the amplifiers will be muted.If the mute pin is left floating or connected to a logic-low level,the amplifier will be in a non-muted state.Refer to the Typical Performance Characteristics section for curves concerning Mute Attenu-ation vs Mute Pin Voltage.STANDBY MODEThe standby mode of the LM4701allows the user to drasti-cally reduce power consumption when the amplifier is idle. By placing a logic-high voltage on the standby pin,the ampli-fier will go into Standby Mode.In this mode,the current drawn from the V CC supply is typically less than10µA total for both amplifiers.The current drawn from the V EE supply is typically2.1mA.Clearly,there is a significant reduction in idle power consumption when using the standby mode.Re-fer to the Typical Performance Characteristics section for curves showing Supply Current vs Standby Pin Voltage for both supplies.UNDER-VOLTAGE PROTECTIONUpon system power-up,the under-voltage protection cir-cuitry allows the power supplies and their corresponding ca-pacitors to come up close to their full values before turning on the LM4701such that no DC output spikes occur.Upon turn-off,the output of the LM4701is brought to ground be-fore the power supplies such that no transients occur at power-down.OVER-VOLTAGE PROTECTIONThe LM4701contains over-voltage protection circuitry that limits the output current to approximately3.5Apk while also providing voltage clamping,though not through internal clamping diodes.The clamping effect is quite the same, however,the output transistors are designed to work alter-nately by sinking large current spikes.SPiKe PROTECTIONThe LM4701is protected from instantaneous peak-temperature stressing of the power transistor array. The Safe Operating Area graph in the Typical Performance Characteristics section shows the area of device operation where SPiKe Protection Circuitry is not enabled.The wave-form to the right of the SOA graph exemplifies how the dy-namic protection will cause waveform distortion when en-abled.THERMAL PROTECTIONThe LM4701has a sophisticated thermal protection scheme to prevent long-term thermal stress of the device.When the temperature on the die reaches165˚C,the LM4701shuts down.It starts operating again when the die temperature drops to about155˚C,but if the temperature again begins to rise,shutdown will occur again at165˚C.Therefore,the de-vice is allowed to heat up to a relatively high temperature ifPulse Power LimitDS100835-40Pulse Power LimitDS100835-41Supply Current vsCase TemperatureDS100835-42Standby Current(I CC)vs Standby Pin VoltageDS100835-43Supply Current(I EE)vsStandby Pin VoltageDS100835-44Input Bias Current vsCase TemperatureDS100835-45 11Application Information(Continued)the fault condition is temporary,but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion be-tween the thermal shutdown temperature limits of165˚C and 155˚C.This greatly reduces the stress imposed on the IC by thermal cycling,which in turn improves its reliability under sustained fault conditions.Since the die temperature is directly dependent upon the heat sink used,the heat sink should be chosen such that thermal shutdown will not be reached during normal ing the best heat sink possible within the cost and space constraints of the system will improve the long-term reliability of any power semiconductor device,as discussed in the Determining the Correct Heat Sink Section. DETERMINING MAXIMUM POWER DISSIPATIONPower dissipation within the integrated circuit package is a very important parameter requiring a thorough understand-ing if optimum power output is to be obtained.An incorrect maximum power dissipation calculation may result in inad-equate heat sinking causing thermal shutdown and thus lim-iting the output power.Equation(1)exemplifies the theoretical maximum power dis-sipation point of each amplifier where V CC is the total supply voltage.P DMAX=V CC2/2π2R L(1) Thus by knowing the total supply voltage and rated output load,the maximum power dissipation point can be calcu-lated.Refer to the graphs of Power Dissipation vs Output Power in the Typical Performance Characteristics section which show the actual full range of power dissipation not just the maximum theoretical point that results from equation(1). DETERMINING THE CORRECT HEAT SINKThe choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level such that the thermal protection circuitry does not operate under normal circumstances.The thermal resistance from the die(junction)to the outside air(ambient)is a combination of three thermal resistances,θJC,θCS andθSA.The thermal resistance,θJC(junction to case),of the LM4701is2˚C/ing Thermalloy Therma-cote thermal compound,the thermal resistance,θCS(case to sink),is about0.2˚C/W.Since convection heat flow(power dissipation)is analogous to current flow,thermal resistance is analogous to electrical resistance,and temperature drops are analogous to voltage drops,the power dissipation out of the LM4701is equal to the following:P DMAX=(T JMAX−T AMB)/θJA(2) where T JMAX=150˚C,T AMB is the system ambient tempera-ture andθJA=θJC+θCS+θSA.Once the maximum package power dissipation has been calculated using equation(1),the maximum thermal resis-tance,θSA,(in˚C/W)for a heat sink can be calculated.This calculation is made using equation(3)which is derived by solving forθSA in equation(2).θSA=[(T JMAX−T AMB)−P DMAX(θJC+θCS)]/P DMAX(3) Again it must be noted that the value ofθSA is dependent upon the system designer’s amplifier requirements.If the ambient temperature that the audio amplifier is to be working under is higher than25˚C,then the thermal resistance for the heat sink,given all other things are equal,will need to be smaller.SUPPLY BYPASSINGThe LM4701has excellent power supply rejection and does not require a regulated supply.However,to improve system performance as well as eliminate possible oscillations,the LM4701should have its supply leads bypassed with low-inductance capacitors having short leads that are lo-cated close to the package terminals.Inadequate power supply bypassing will manifest itself by a low frequency oscil-lation known as“motorboating”or by high frequency insta-bilities.These instabilities can be eliminated through multiple bypassing utilizing a large tantalum or electrolytic capacitor (10µF or larger)which is used to absorb low frequency variations and a small ceramic capacitor(0.1µF)to prevent any high frequency feedback through the power supply lines. If adequate bypassing is not provided,the current in the sup-ply leads which is a rectified component of the load current may be fed back into internal circuitry.This signal causes distortion at high frequencies requiring that the supplies be bypassed at the package terminals with an electrolytic ca-pacitor of470µF or more.BRIDGED AMPLIFIER APPLICATIONOne common power amplifier configuration is shown in Fig-ure2and is referred to as“bridged mode”operation.Bridged mode operation is different from the classical single-ended amplifier configuration where one side of the output load is connected to ground.A bridge amplifier design has a distinct advantage over the single-ended configuration,as it provides differential drive to the load,thus doubling output swing for a specified supply voltage.Consequently,theoretically four times the output power is possible as compared to a single-ended amplifier under the same conditions.This increase in attainable output power assumes that the amplifier is not current limited or clipped.A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation.For each operational amplifier in a bridge con-figuration,the internal power dissipation will increase by a factor of two over the single ended dissipation.Since there are two amplifiers used in a bridge configuration,the maxi-mum system power dissipation point will increase by a factor of four over the figure obtained by equation(1).This value of P DMAX can be used to calculate the correct size heat sink for a bridged amplifier application,assuming that both IC’s are mounted on the same heatsink.Since the inter-nal dissipation for a given power supply and load is in-creased by using bridged-mode,the heatsink’sθSA will have to decrease accordingly as shown by equation(3).Refer to the section,Determining the Correct Heat Sink,for a more detailed discussion of proper heat sinking for a given appli-cation.SINGLE-SUPPLY AMPLIFIER APPLICATIONThe typical application of the LM4701is a split supply ampli-fier.But as shown in Figure3,the LM4701can also be used in a single power supply configuration.This involves using some external components to create a half-supply bias which is used as the reference for the inputs and outputs. Thus,the signal will swing around half-supply much like it swings around ground in a split-supply application.Along with proper circuit biasing,a few other considerations must be accounted for to take advantage of all of the LM4701 functions.12Application Information(Continued)The LM4701possesses a mute and standby function with in-ternal logic gates that are half-supply referenced.Thus,to enable either the mute or standby function,the voltage at these pins must be a minimum of 2.5V above half-supply.In single-supply systems,devices such as microprocessors and simple logic circuits used to control the mute and standby functions,are usually referenced to ground,not half-supply.Thus,to use these devices to control the logic circuitry of the LM4701,a “level shifter”,like the one shown in Figure 5,must be employed.A level shifter is not needed in a split-supply configuration since ground is also half-supply.When the voltage at the Logic Input node is 0V,the 2N3904is “off”and thus resistor R C pulls up mute or standby input to the supply.This enables the mute or standby function.When the Logic Input is 5V,the 2N3904is “on”and consequently,the voltage at the collector is essentially 0V.This will disable the mute or standby function,and thus the amplifier will be in its normal mode of operation.R SHIFT ,along with C SHIFT ,cre-ates an RC time constant that reduces transients when the mute or standby functions are enabled or disabled.Addition-ally,R SHIFT limits the current supplied by the internal logic gates of the LM4701which insures device reliability.Refer to the Mute Mode and Standby Mode sections in the Applica-tion Information section for a more detailed description of these functions.CLICKS AND POPSIn the typical application of the LM4701as a split-supply au-dio power amplifier,the IC exhibits excellent “click”and “pop”performance when utilizing the mute and standby functions.In addition,the device employs Under-Voltage Protection,which eliminates unwanted power-up and power-down tran-sients.The basis for these functions are a stable and con-stant half-supply potential.In a split-supply application,ground is the stable half-supply potential.But in a single-supply application,the half-supply needs to charge up just like the supply rail,V CC .This makes the task of attaining a clickless and popless turn-on more challenging.Any uneven charging of the ampli-fier inputs will result in output clicks and pops due to the dif-ferential input topology of the LM4701.To achieve a transient free power-up and power-down,the voltage seen at the input terminals should be ideally the same.Such a signal will be common-mode in nature,and will be rejected by the LM4701.In Figure 3,the resistor R INP serves to keep the inputs at the same potential by limiting the voltage difference possible between the two nodes.This should significantly reduce any type of turn-on pop,due to an uneven charging of the amplifier inputs.This charging isbased upon a specific application loading and thus,the sys-tem designer may need to adjust these values for optimum performance.As shown in Figure 3,the resistors labeled R BI help bias up the LM4701off the half-supply node at the emitter of the 2N3904.But due to the input and output coupling capacitors in the circuit,along with the negative feedback,there are two different values of R BI ,namely 10k Ωand 200k Ω.These re-sistors bring up the inputs at the same rate resulting in a pop-less turn-on.Adjusting these resistors values slightly may re-duce pops resulting from power supplies that ramp extremely quick or exhibit overshoot during system turn-on.AUDIO POWER AMPLlFIER DESIGN Design a 25W/8ΩAudio Amplifier Given:Power Output 25WrmsLoad Impedance 8ΩInput Level 1Vrms(max)Input Impedance47k ΩBandwidth 20Hz to 20kHz ±0.25dB A designer must first determine the power supply require-ments in terms of both voltage and current needed to obtain the specified output power.V OPEAK can be determined from equation (4)and I OPEAK from equation (5).(4)(5)To determine the maximum supply voltage,the following conditions must be considered.Add the dropout voltage to the peak output swing V OPEAK ,to get the supply rail at a cur-rent of I OPEAK .The regulation of the supply determines the unloaded voltage which is usually about 15%higher.The supply voltage will also rise 10%during high line conditions.Therefore the maximum supply voltage is obtained from the following equation:Max Supplies ≈±(V OPEAK +V OD )(1+Regulation)(1.1)For 25W of output power into an 8Ωload,the required V O -PEAK is 20V.A minimum supply rail of ±25V results from add-ing V OPEAK and V OD .With regulation,the maximum supplies are ±31.7V and the required I OPEAK is 2.5A from equation (5).At this point it is a good idea to check the Power Output vs Supply Voltage to ensure that the required output power is obtainable from the device while maintaining low THD+N.In addition,the designer should verify that with the required power supply voltage and load impedance,that the required heatsink value θSA is feasible given system cost and size constraints.Once the heatsink issues have been addressed,the required gain can be determined from equation (6).(6)From equation (6),the minimum A V is A V ≥14.14.By selecting a gain of 21,and with a feedback resistor,R F =20k Ω,the value of R I follows from equation (7).R I =R F (A V −1)(7)Thus with R J =1k Ωa non-inverting gain of 21will result.Since the desired input impedance was 47k Ω,a value of 47k Ωwas selected for R IN .The final design step is to address the bandwidth requirements which must be stated as a pair of −3dB frequency points.Five times away from a −3dBDS100835-9FIGURE 5.Level Shift Circuit13Application Information(Continued)point is0.17dB down from passband response which is bet-ter than the required±0.25dB specified.This fact results in a low and high frequency pole of4Hz and100kHz respec-tively.As stated in the External Components section,R I in conjunction with C I create a high-pass filter.C I≥1/(2π*1kΩ*4Hz)=39.8µF;use39µF.The high frequency pole is determined by the product of the desired high frequency pole,f H,and the gain,A V.With a A V =21and fH=100kHz,the resulting GBWP of2.1MHz is less than the minimum GBWP of5MHz for the LM4701.This will ensure that the high frequency response of the amplifier will be no worse than0.17dB down at20kHz which is well within the bandwidth requirements of the design.14。
LM1117-adj中文资料
订购信息
封装 3 引脚 SOT-223
3 引脚 SOT-220 3 引脚 SOT-252
8 引脚 LLP
TO-263
温度范围 0℃~125℃
-40℃~125℃ 0℃~125℃
0℃~125℃
-40℃~125℃ 0℃~125℃
-40℃~125℃ 0℃~125℃
型号 LM1117MPX-ADJ LM1117MPX-1.8 LM1117MPX-2.5 LM1117MPX-2.85 LM1117MPX-3.3 LM1117MPX-5.0 LM1117IMPX-ADJ LM1117IMPX-3.3 LM1117IMPX-5.0 LM1117T-ADJ LM1117T-1.8 LM1117T-2.5 LM1117T-2.85 LM1117T-3.3 LM1117T-5.0 LM1117DTX-ADJ LM1117DTX-1.8 LM1117DTX-2.5 LM1117DTX-2.85 LM1117DTX-3.3 LM1117DTX-5.0 LM1117IDTX-ADJ LM1117IDTX-3.3 LM1117IDTX-5.0 LM1117LTX-ADJ LM1117LTX-1.8 LM1117LTX-2.5 LM1117LTX-2.85 LM1117LTX-3.3 LM1117LTX-5.0 LM1117ILTX-ADJ LM1117ILTX-3.3 LM1117ILTX-5.0 LM1117SX-ADJ LM1117SX-2.85 LM1117SX-3.3 LM1117SX-5.0
LM1117-1.8 VIN = 3.2V, 0≤IOUT≤800mA
LM1117-2.5 VIN = 3.9V, 0≤IOUT≤800mA
LM1117-2.85 VIN = 4.25V, 0≤IOUT≤800mA
LM1117-1.2中文资料
AMS1117-3.0 AMS1117-3.0 AMS1117-3.3 AMS1117-3.3
AMS1117-5.0 AMS1117-5.0
AMS1117-ADJ AMS1117-2.85
AMS1117-1.2 AMS1117-3.0 AMS1117-1.5 AMS1117-3.3
AMS1117-1.8 AMS1117-5.0 AMS1117-2.5
输出电压
输出电压温度稳定性 线性调整 负载调整 漏失电压 静态电流 纹波抑制比 可调管脚电流 可调管脚电流变化 温度稳定性 长期稳定性 RMS输出噪声
热阻系数 (无散热片)
符号
测试条件
最小值 典型值 最大值 单位
AMS1117-2.85,
IOUT=10mA, VIN=4.85V,TJ=25°C , 2.820 2.850 .880
ADJ R1
R2 VOUT=VREF x (1+R2/R1)+IADJ x R2
VOUT 22 F
图 2. 典型可调输出电压 注：以上线路及参数仅供参考，实际的应用电路请在充分的实测基础上设定参数。
1111111111111111111111111111111111111111111111111111111111111111111111
推荐工作条件
参数 输入电压 工作结温范围
符号 VIN TJ
范围 15
-40 ~ +125
单位 V °C
电气特性(除非特别指定，否则黑色字体所示的参数，Tamb=25°C，正常工作结温范围 -40°C ~125°C。)
参数 基准电压
输出电压
符号
测试条件
最小值 典型值 最大值 单位
D110277中文资料
* Color Black available by request, please contact Customer Service.
D ±0,30 (.012) 16,00 (.630) 16,00 (.630) 16,00 (.630) 16,00 (.630) 19,00 (.748)
E
Cable Diameter
D Cannon D-Subminiature Accessories
EMI Shielded Backshell
Snap-Together Metalized Plastic
Metalized plastic backshells reduce EMI/RFI emissions. Metalized plastic provides a light weight solution. Design includes integral strain relieving cable clamp.
2
元器件交易网
D Cannon D-Subminiature Accessories
Selection Guide
Locking Hardware
Jackscrew Assembly Jackpost Assembly
see page 15
11,00 (.433)
42,01 (1.654) 8,99 (.354)
13,00 (.512)
Dimensions shown in mm (inch) Specifications and dimensions subject to change
4
元器件交易网
D ±0,30 (.120) 16,00 (.630) 16,00 (.630) 16,00 (.630) 16,00 (.630) 19,00 (.748)
LM1117中文资料
3.267 3.168
3.300 3.333 V 3.300 3.432 V
ｍ
LM1117I-5.0 IOUT = 10mA,VIN-VOUT =7V, TJ = 25℃
4.950
5.000
5.050
ｒV
∆VOUT 线性调整率
0≤IOUT≤800mA, 6.5V≤VIN≤12V LM1117I-ADJ
续上表
符号
参数
条件
最小值 典型值 最大值 单位
∆VOUT 线性调整率
LM1117-2.85 IOUT = 0m来自, 4.25V≤VIN≤10V
1
6 mV
LM1117-3.3
IOUT = 0mA, 4.75V≤VIN≤15V LM1117-5.0
1
6 mV
．
∆VOUT 负载调整率
IOUT = 0mA, 6.5V≤VIN≤15V LM1117-ADJ
N05A N06A
编带和卷轴 编带和卷轴
ｍ
-40℃~125℃
LM1117IMPX-ADJ LM1117IMPX-3.3
N03B N05B
编带和卷轴 编带和卷轴
ｒ
0℃~125℃
LM1117IMPX-5.0 LM1117T-ADJ
N06B LM1117T-ADJ
ａ 编带和卷轴
直条
LM1117T-1.8 LM1117T-2.5
3.267 3.235
3.300 3.333 V 3.300 3.365 V
LM1117-5.0
∆VOUT 线性调整率
IOUT = 10mA,VIN-VOUT =7V, TJ = 25℃ 0≤IOUT≤800mA, 6.5V≤VIN≤12V LM1117-ADJ IOUT = 10mA, 1.5V≤VIN-VOUT≤13.75V
LM1000说明书
统用（1A 可供 4 台语音系统用）。
5
安装步骤
LM1000 用戶手冊
1、依 PABX 系统的号码安排及单机使用说明， 参照
接线图依序： （1） 将整流变压器插到市电插座上。 （2） 将 整 流 变 压 器 的 输 出 接 至 语 音 系 统 的
“DC12V”电源输入，同时注意状态指示 灯会闪亮，表示可正常工作。 （3） 安装多个语音系统者，请参考第 3 页，使 用 8 芯复接线复接。 （4） 接上分机线。
LM1000 用戶手冊
PABX 资料
语音系统能匹配大多数的 PABX 可即插即 用（转接键时间 0.13 秒以上，转接及取回外 线只使用转接键而不需功能码者）。但有些 PABX 转接或取回外线的操作方式较为特殊， 请先按照 PABX 的单机操作方式设定资料。
智能型语音系统除了一般自动总机之 转接功能外，又增加了一个公共语音信箱 以供留言，内藏语音及自动测试功能，让 装机更为简易。信号音智能模式及弹性号 码计划，让交换机连线更容易完成。
3、查询系统的段落/片语号码是从[接待语]拨至该段落所 按的号码，如下图所示。
接待语
M_1
-------------
M_9
M_11 -------
M_19
M_91 -------- M_99
14
LM1000 用戶手冊
4、查询语音播放完毕后，语音系统会播放[查询后引导语]。 5、段落号首 Y：1=查询段落 1，1×，1××，1×××
4
其它说明：
LM1000 用戶手冊
1、内置电池仅用于保持语音及资料，可外接
12V/7AH 电池作为不停电设备。
2、若外接电池极性错误，语音系统电源会自
动切断，请将整流变压器拔起，确认电
DW01中文资料
一、主要特性静态电流待机电流（检测到过放之后）过充检测精度（Topt=25℃）过充检测精度（Topt=0 到50℃）过放检测精度过放检测电压过流保护过充延迟（VDD=4.4V）过放延迟（VDD=2.2V 带有内置电容）封装典型值：4.0uA典型值：0.2uA±50mV±60mV±100mV2.0V 到3.0V，每步0.005V 0.04V 到0.32V，每步0.04V 110mS22mS（最小值）SOT23-6/6-pin二、基本描述DW01 是一款单节可充电锂电池保护集成电路，具有过充、过放、过流及短路保护功能。
IC 内部包含：三个电压检测电路、一个基准电路、一个延迟电路、一个短路保护电路和一个逻辑电路。
当充电电压逐渐增大超过过充检测电路的阈值VDET1 时，Cout Pin 的输出电压即过充检测电路的输出电压VD1 会变到低电位，也就是充电器负端的电位。
在进入过充保护状态后，当VDD 电压降低到VREL1 下方或者当电池组脱离充电器而接一个负载，且VDD 介于VDET1 与VREL1 之间时VD1 可以复位，即Cout Pin 输出变为高电位。
当放电电压低于过放检测电路的阈值VDET2 时，经过一段固定的延迟时间，Dout Pin 的输出即过放检测电路的输出VD2 会变为低电位。
这时，若给电池充电，当电池电压上升到过放检测电路的阈值电压之上时，VD2 恢复，Dout 的输出电压变为高电平。
当有过流情况出现时，内部过流检测电路会检测到，经过一段固定的延迟时间后，VD3 和Dout 变为低电平，放电回路被切断。
这时，若将电池组从负载系统中分开，VD3 会恢复使Dout 变为高电平。
当有外部短路电流时，短路保护电路会立即使Dout变为低电位，当外部短路电流消失后，Dout 会转换为高电位。
在检测到过放之后，会通过关闭一些内部电路使电源电流非常低。
IC 过充检测电路的延迟时间可以通过连接外部电容进行设置。
APTGT100A170T中文资料
VR=1700V
50% duty cycle
IF = 100A IF = 100A VR = 900V
di/dt =800A/µs
100 1.8 1.9 385 490 28 46
2.2
V ns µC
May, 2005
APT website –
2-5
APTGT100A170T – Rev 0
元器件交易网
APTGT100A170T
Temperature sensor NTC (T0406 on for more information).
Min
Typ 9 0.36 0.3 370 40 650 180 400 50 800 300 32 31
Max
Unit nF
ns
ns
mJ
Reverse diode ratings and characteristics
Symbol Characteristic VRRM IRM IF(A V) VF trr Qrr
T: Thermistor temperature
Thermal and package characteristics
Symbol Characteristic RthJC VISOL TJ TSTG TC Torque Wt Junction to Case Operating junction temperature range Storage Temperature Range Operating Case Temperature Mounting torque Package Weight IGBT Diode
Package outline (dimensions in mm)
LM1117 中文PDF资料
VOUT 22 F
R2 VOUT=VREF x (1+R2/R1)+IADJ x R2
图 2. 典型可调输出电压
注：以上线路及参数仅供参考，实际的应用电路请在充分的实测基础上设定参数。
输出电压变化(%)
典型电气特性曲线
温度稳定性
负载瞬态反应（VOUT=5 V）
可调管脚电流 ( A)
参数符号测试条件最小值典型值最大值单位x1117adj基准电压vrefiout10mavinvout2vtj25c123812501262v10maiout1a14vvinvout10v122512501270x111712iout10mvin32vtj25c1176121224v10maiout1a30vvin10v1152121248x111715iout10mvin35vtj25c148515001515v10maiout1a30vvin10v147015001530x111718iout10mavin38vtj25c178218001818v0iout1a32vvin10v176418001836输出电压voutx111725iout10mavin45vtj25c247525002525v0iout1a39vvin10v245025002550x111733iout10mavin5vtj25c326733003333v0iout1a475vvin10v323533003365x111750iout10mavin7vtj25c49505000505v0iout1a65vvin12v49005000510输出电压温度稳定性tsout03vinminvin12v线性调整rline37mvvoutfixedadjiout10ma负载调整rload10maiout1avoutfixedadj612mviout100ma100120漏失电压vdropiout500ma105125viout1a110130静态电流iq425vvin65v510mafripple120hzvinvout3v纹波抑制比psrr6075dbvripple1vpp可调管脚电流iadj60120a可调管脚电流变化0iout1a14vvinvout10v025a温保点tsd150c限流点ilimit121415ax1117参数符号测试条件最小值典型值最大值单位温度稳定性05长期稳定性tamb125c1000hrs0
LM2710MT-ADJ资料
LM2710Step-up PWM DC/DC Converter Integrated with 5BuffersGeneral DescriptionThe LM2710is a compact bias solution for TFT displays.It has a current mode PWM step-up DC/DC converter with a 1.4A,0.17Ωinternal switch.Capable of generating 8V at 300mA from a Lithium Ion battery,the LM2710is ideal for generating bias voltages for large screen LCD panels.The LM2710can be operated at switching frequencies of 600kHz or 1.25MHz,allowing for easy filtering and low noise.An external compensation pin gives the user flexibility in setting frequency compensation,which makes possible the use of small,low ESR ceramic capacitors at the output.The LM2710uses a patented internal circuitry to limit startup inrush current of the boost switching regulator without the use of an external softstart capacitor.An external softstart pin enables the user to tailor the softstart to a specific application.The LM2710contains a Vcom buffer and 4Gamma buffers capable of supplying 50mA source and sink.The TSSOP-20package ensures a low profile overall solu-tion.Featuresn 1.4A,0.17Ω,internal power switch n V IN operating range:2.2V to 7.5Vn 600kHz/1.25MHz selectable frequency step-up DC/DC convertern 20pin TSSOP packagen Inrush current limiting circuitry n External softstart override n Vcom buffern 4Gamma buffersApplicationsn LCD Bias Supplies n Handheld Devices n Portable ApplicationsnCellular Phones/Digital CamerasTypical Application Circuit20043431February 2004LM2710TFT Step-up PWM DC/DC Converter Integrated with 5Buffers©2004National Semiconductor Corporation Connection DiagramTop View20043404TSSOP 20packageT JMAX=125˚C,θJA =120˚C/W (Note 1)Pin DescriptionPin Name Function1V SW Power switch input.2V IN Switching Regulator Power input.3SHDN Shutdown pin,active low.4FSLCT Frequency Select pin.FSLCT =V IN for 1.25MHz,FSLCT =AGND or floating for 600kHz.5Vs+Vcom and Gamma Buffer input supply.6Vcom-in Vcom Buffer input.7GMA1-in Gamma Buffer input.8GMA2-in Gamma Buffer input.9GMA3-in Gamma Buffer input.10GMA4-in Gamma Buffer input.11GMA4-out Gamma Buffer output.12GMA3-out Gamma Buffer output.13GMA2-out Gamma Buffer output.14GMA1-out Gamma Buffer output.15Vcom-out Vcom Buffer output.16SS Soft start pin.17V C Boost Compensation Network Connection.18FB Output Voltage Feedback input.19AGND Vcom and Gamma Buffer ground,Analog ground connection for Regulator.20GNDSwitch Power Ground.L M 2710 2Pin FunctionsV SW (Pin 1):This is the drain of the internal NMOS power switch.Minimize the metal trace area connected to this pin to minimize EMI.V IN (Pin 2):Input Supply Pin.Bypass this pin with a capacitor as close to the device as possible.The capacitor should connect between V IN and GND.SHDN(Pin 3):Shutdown Pin.The shutdown pin signal is active low.A voltage of less than 0.3V disables the device.A voltage greater than 0.85V enables the device.FSLCT(Pin 4):Frequency Select Pin.Connecting FSLCT to AGND selects a 600kHz operating frequency for the switch-ing regulator.Connecting FSLCT to V IN selects a 1.25MHz operating frequency.If FSLCT is left floating,the switching frequency defaults to 600kHz.Vs+(Pin 5):Supply pin for the Vcom buffer and the four Gamma buffers.Bypass this pin with a capacitor as close to the device as possible.The capacitor should connect be-tween Vs+and GND.Vcom-in(Pin 6):Vcom Buffer input pin.GMA1-in(Pin 7):Gamma Buffer input pin.GMA2-in(Pin 8):Gamma Buffer input pin.GMA3-in(Pin 9):Gamma Buffer input pin.GMA4-in(Pin 10):Gamma Buffer input pin.GMA4-out(Pin 11):Gamma Buffer output pin.GMA3-out(Pin 12):Gamma Buffer output pin.GMA2-out(Pin13):Gamma Buffer output pin.GMA1-out(Pin 14):Gamma Buffer output pin.Vcom-out(Pin 15):Vcom Buffer output pin.SS(Pin 16):Softstart pin.Connect capacitor to SS pin and AGND to slowly ramp inductor current on startup.V C (Pin 17):Compensation Network for Boost switching regulator.Connect resistor/capacitor network between V C pin and AGND for boost switching regulator AC compensa-tion.FB(Pin 18):Feedback pin.Set the output voltage by select-ing values of R1and R2using:Connect the ground of the feedback network to the AGND plane,which can be tied directly to the GND pin.AGND(Pin 19):Analog ground pin.Ground connection for the Vcom buffer,Gamma buffers and the boost switching regulator.AGND must be tied directly to GND at the pins.GND(Pin 20):Power ground pin.Ground connection for the NMOS power device of the boost switching regulator.GND must be tied directly to AGND at the pins.Ordering InformationOrder Number Package Type NSC Package DrawingSupplied AsLM2710MT-ADJ TSSOP-20MTC2073Units,RailLM2710MTX-ADJTSSOP-20MTC202500Units,Tape and ReelLM27103Block Diagrams2004340320043451L M 2710 4Absolute Maximum Ratings (Note 2)If Military/Aerospace specified devices are required,please contact the National Semiconductor Sales Office/Distributors for availability and specifications.V IN-0.3V to 7.5V V SW Voltage -0.3V to 18V FB Voltage -0.3V to 7V V C Voltage 0.965V to 1.565VSHDN Voltage -0.3V to V IN FSLCT Voltage AGND to V IN Supply Voltage,Vs+-0.3V to 12V Buffer Input Voltage Rail-to-Rail Buffer Output VoltageRail-to-RailESD Ratings (Note 3)Human Body Model 2kV Machine Model200VOperating ConditionsOperating Temperature −40˚C to +125˚C Storage Temperature −65˚C to +150˚CSupply Voltage,V IN 2.2V to 7.5VV SW Voltage17V Supply Vcom Buffer,Vs+4V to 12V Supply Gamma Buffer,Vs+4V to 12VElectrical CharacteristicsSpecifications in standard type face are for T J =25˚C and those with boldface type apply over the full Operating Tempera-ture Range (T J =−40˚C to +125˚C).Unless otherwise specified,V IN =2.2V and Vs+=8V,Rox =50Ω,Cox =1nF.Switching Regulator Symbol ParameterConditionsMin (Note 4)Typ (Note 5)Max (Note 4)UnitsI QQuiescent CurrentNot Switching,FSCLT =0V 1.62mA Not Switching,FSCLT =V IN 1.65 2.2Switching,FSCLT =0V 2.53Switching,FSCLT =V IN 3.44Shutdown mode615µA V FBFeedback Voltage 1.239 1.265 1.291V %V FB /∆V IN Feedback Voltage Line Regulation0.030.05%/V I CL Switch Current Limit (Note 6)V IN =2.5V,V OUT =8V 1.4A R DSON Switch R DSON (Note 7)V IN =2.7V170m ΩI B FB Pin Bias Current(Note 8)3090nA V IN Input Voltage Range 2.27.5V I SS Soft Start Current 51115µA T SS Internal Soft Start Ramp TimeFSLCT =0V6.710mS g m Error Amp Transconductance ∆I =5µA60135250µmho A V Error Amp Voltage Gain 135V/V D MAX Maximum Duty Cycle 7885%f S Switching Frequency FSLCT =0V 500600700kHz FSLCT =V IN 0.91.25 1.5MHz I L Switch Leakage Current V SW =17V 0.18520µA SHDN SHDN Threshold Output High 0.850.6V Output Low 0.60.3V I SHDN Shutdown Pin Current 0V ≤SHDN ≤V IN0.51µA UVPOn Threshold 1.8 1.92V Off Threshold 1.7 1.8 1.9V Hysteresis100mV LM27105Electrical CharacteristicsSpecifications in standard type face are for T J =25˚C and those with boldface type apply over the full Operating Tempera-ture Range (T J =−40˚C to +125˚C).Unless otherwise specified,V IN =2.2V and Vs+=8V,Rox =50Ω,Cox =1nF.BUFFERS Symbol ParameterConditionsMin (Note 4)Typ (Note 5)Max (Note 4)Units V OS Input offset voltage 2.510mV ∆V os /∆T Offset Voltage Drift 8µV/˚C I B Input Bias Current170800nA CMVR Input Common-mode Voltage Range0.05Vs+-0.05V Z IN Input Impedance 400k ΩC IN Input Capacitance 1pFI OUTContinuous Output CurrentVs+=8V,Source 415971mAVs+=8V,Sink −65−53−36Vs+=12V,Source 507185Vs+=12V,Sink−75−61−42V OUT SwingR L =10k,Vo min.0.075V R L =10k,Vo max.7.88R L =2k,Vo min.0.075R L =2k,Vo max.7.865A VCL Voltage Gain R L =2k ΩR L =10k Ω0.9950.99850.9980.9999V/V NL Gain Linearity R L =2k Ω,Buffer input=0.5to (Vs+-0.5V)0.01%Vs+Supply Voltage 412V PSRR Power Supply Rejection RatioVs+=4to 12V 90316µV/V Is+Supply Current/Amplifier Vo =Vs+/2,No Load 12mA SR Slew Rate 10V/µs BW Bandwidth -3dB,R L =10k Ω,C L =10pf 6MHz φ0Phase Margin50Deg˚Note 1:The maximum allowable power dissipation is a function of the maximum junction temperature,T J (MAX),the junction-to-ambient thermal resistance,θJA ,and the ambient temperature,T A .See the Electrical Characteristics table for the thermal resistance of various layouts.The maximum allowable power dissipation at any ambient temperature is calculated using:P D (MAX)=(T J(MAX)−T A )/θJA .Exceeding the maximum allowable power dissipation will cause excessive die temperature,and the regulator will go into thermal shutdown.Note 2:Absolute maximum ratings are limits beyond which damage to the device may occur.Operating Ratings are conditions for which the device is intended to be functional,but device parameter specifications may not be guaranteed.For guaranteed specifications and test conditions,see the Electrical Characteristics.Note 3:The human body model is a 100pF capacitor discharged through a 1.5k Ωresistor into each pin.The machine model is a 200pF capacitor discharged directly into each pin.Note 4:All limits guaranteed at room temperature (standard typeface)and at temperature extremes (bold typeface).All room temperature limits are 100%production tested or guaranteed through statistical analysis.All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC)methods.All limits are used to calculate Average Outgoing Quality Level (AOQL).Note 5:Typical numbers are at 25˚C and represent the most likely norm.Note 6:Duty cycle affects current limit due to ramp generator.See Switch Current Limit vs.V IN and Switch Current Limit vs.Temperature graphs in the Typical Performance Characteristics section.Note 7:See Typical Performance Characteristics section for Tri-Temperature data for R DSON vs.V IN .Note 8:Bias current flows into FB pin.L M 2710 6Typical Performance CharacteristicsEfficiency vs.Load Current (V OUT =8V,f S =600kHz)Efficiency vs.Load Current (V OUT =8V,f S =1.25MHz)2004342620043425Efficiency vs.Load Current (V OUT =10V,f S =1.25MHz)Switch Current Limit vs.Temperature(V OUT =8V)2004346020043420Switch Current Limit vs.V INR DSON vs.V IN (I SW =1A)2004342220043427LM27107Typical Performance Characteristics(Continued)I Q vs.V IN(600kHz,not switching)I Q vs.V IN(600kHz,switching)2004342120043429I Q vs.V IN(1.25MHz,not switching)I Q vs.V IN(1.25MHz,switching)2004342120043419I Q vs.V IN (In shutdown)Frequency vs.V IN(600kHz)2004341820043423L M 2710 8Typical Performance Characteristics(Continued)Frequency vs.V IN(1.25MHz)Feedback Pin Current vs.Temperature2004342420043457C SS Pin Current vs.V IN Load Transient Response2004345820043476 V OUT=8V,V IN=3V,F=1.25MHz1)Load,80mA to260mA to80mA2)I L,500mA/div,DC3)V OUT,100mV/div,ACT=100µs/divLoad Transient Response Load Transient Response20043483 V OUT=8V,V IN=3V,F=600kHz1)Load,80mA to260mA to80mA2)I L,500mA/div,DC3)V OUT,200mV/div,ACT=100µs/div20043475V OUT=10V,V IN=5V,F=1.25MHz1)Load,195mA to385mA to195mA2)I L,500mA/div,DC3)V OUT,500mV/div,ACT=100µs/divLM27109Typical Performance Characteristics(Continued)Internal Soft Start Internal Soft Start20043479V OUT=8V,V IN=3V,R LOAD=27Ω,C SS=none,F=600kHz1)SHDN,1V/div,DC2)I L,500mA/div,DC3)V OUT,5V/div,DCT=1ms/div20043477V OUT=8V,V IN=3V,R LOAD=27Ω,C SS=none,F=1.25MHz1)SHDN,1V/div,DC2)I L,500mA/div,DC3)V OUT,5V/div,DCT=1ms/divExternal Soft StartInput Offset Voltage mon Mode Voltage(3units)20043478V OUT=8V,V IN=3V,R LOAD=27Ω,C SS=330nF,F=1.25MHz1)SHDN,1V/div,DC2)I L,500mA/div,DC3)V OUT,5V/div,DCT=4ms/div20043461Input Offset Voltage mon Mode Voltage(Over Temperature)Input Bias Current mon Mode Voltage2004346220043463LM27110Typical Performance Characteristics(Continued)Output Voltage vs.Output Current(sinking)Output Voltage vs.Output Current(sourcing)2004346420043465Supply Current mon Mode VoltageLarge Signal Step Response (50Ω,1nF pensation)2004346620043467Large Signal Step Response (no pensation)Positive Slew Rate vs.Capacitive Load2004346820043469LM271011Typical Performance Characteristics(Continued)Negative Slew Rate vs.Capacitive LoadPhase Margin vs.Capacitive Load2004347020043471Unity Gain Frequency vs.Capacitive Load CMRR vs.Frequency2004347220043473PSRR vs.Frequency20043474L M 2710 12OperationCONTINUOUS CONDUCTION MODEThe LM2710is a current-mode,PWM boost regulator.Aboost regulator steps the input voltage up to a higher outputvoltage.In continuous conduction mode(when the inductorcurrent never reaches zero at steady state),the boost regu-lator operates in two cycles.In the first cycle of operation,shown in Figure1(a),thetransistor is closed and the diode is reverse biased.Energyis collected in the inductor and the load current is supplied byC OUT.The second cycle is shown in Figure1(b).During this cycle,the transistor is open and the diode is forward biased.Theenergy stored in the inductor is transferred to the load andoutput capacitor.The ratio of these two cycles determines the output voltage.The output voltage is defined approximately as:where D is the duty cycle of the switch,D and D'will berequired for design calculationsSETTING THE OUTPUT VOLTAGEThe output voltage is set using the feedback pin and aresistor divider connected to the output as shown in thetypical operating circuit.The feedback pin voltage is1.265V,so the ratio of the feedback resistors sets the output voltageaccording to the following equation:SOFT-START CAPACITORThe LM2710has patented internal circuitry that is used tolimit the inductor inrush current on start-up.This inrushcurrent limiting circuitry serves as a soft-start.However,many applications may require much more soft-start thanwhat is available with the internal circuitry.The external SSpin is used to tailor the soft-start for a specific application.A11µA current charges the external soft-start capacitor,Css.The soft-start time can be estimated as:Tss=Css*0.6V/11µAThe minimum soft-start time is set by the internal soft-startcircuitry,typically7ms for600kHz operation and approxi-mately half that for1.25MHz operation.Only longer soft-starttimes may be implemented using the SS pin and a capacitorC SS.If a shorter time is designed for using the above equa-tion,the internal soft-start circuitry will override it.Due to the unique nature of the dual internal/external soft-start,care was taken in the design to ensure temperaturestable operation.As you can see with the Iss data in theElectrical Characterisitcs table and the graph"Soft-StartCurrent vs.V IN"in the Typical Performance Characterisitcssection,the soft start curent has a temperature coefficientand would lead one to believe there would be significantvariation with temperature.Though the current has a tem-perature coefficient the actual programmed external softstart time does not show this extreme of a temperaturevariation.As you can see in the following transient plots:20043402FIGURE1.Simplified Boost Converter Diagram(a)First Cycle of Operation(b)Second Cycle Of OperationLM271013Operation(Continued)V OUT =8V,V IN =2.5V,R L =27Ω,C SS =330nF,T =4ms/div,F =1.25MHz.Trace:1)SHDN,1V/div,DC Coupled 2)I L ,0.5A/div,DC Coupled 3)V OUT ,5V/div,DC Coupled20043480T A =−20˚C20043481T A =27˚C20043482T A =85˚CWhen programming the softstart time externally,simply use the equation given in the Soft-Start Capacitor section above.This equation uses the typical room temperature value of the soft start current,11µA,to set the soft start time.INTRODUCTION TO COMPENSATIONThe LM2710is a current mode PWM boost converter.The signal flow of this control scheme has two feedback loops,one that senses switch current and one that senses output voltage.To keep a current programmed control converter stable above duty cycles of 50%,the inductor must meet certain criteria.The inductor,along with input and output voltage,will determine the slope of the current through the inductor (see Figure 2(a)).If the slope of the inductor current is too great,the circuit will be unstable above duty cycles of 50%.A 10µH inductor is recommended for most 600kHz applica-tions,while a 4.7µH inductor may be used for most 1.25MHz applications.If the duty cycle is approaching the maximum of 85%,it may be necessary to increase the inductance by as much as 2X.See Inductor and Diode Selection for more detailed inductor sizing.The LM2710provides a compensation pin (V C )to customize the voltage loop feedback.It is recommended that a series combination of R C and C C be used for the compensation network,as shown in the typical application circuit.For any given application,there exists a unique combination of R C and C C that will optimize the performance of the LM2710circuit in terms of its transient response.The series combi-nation of R C and C C introduces a pole-zero pair according to the following equations:where R O is the output impedance of the error amplifier,approximately 1M Ω.For most applications,performance can be optimized by choosing values within the range 5k Ω≤R C ≤60k Ω(R C can be up to 200k Ωif C C2is used,see High Output Capacitor ESR Compensation )and 680pF ≤C C ≤20043405FIGURE 2.(a)Inductor current.(b)Diode current.L M 2710 14Operation(Continued)4.7nF.Refer to the Applications Information section for rec-ommended values for specific circuits and conditions.Refer to the Compensation section for other design PENSATION FOR BOOST DC/DCThis section will present a general design procedure to help insure a stable and operational circuit.The designs in this datasheet are optimized for particular requirements.If differ-ent conversions are required,some of the components may need to be changed to ensure stability.Below is a set of general guidelines in designing a stable circuit for continu-ous conduction operation,in most all cases this will provide for stability during discontinuous operation as well.The power components and their effects will be determined first,then the compensation components will be chosen to pro-duce stability.INDUCTOR AND DIODE SELECTIONAlthough the inductor sizes mentioned earlier are fine for most applications,a more exact value can be calculated.To ensure stability at duty cycles above 50%,the inductor must have some minimum value determined by the minimum input voltage and the maximum output voltage.This equa-tion is:where fs is the switching frequency,D is the duty cycle,and R DSON is the ON resistance of the internal switch taken from the graph "R DSON vs.V IN "in the Typical Performance Char-acteristics section.This equation is only good for duty cycles greater than 50%(D >0.5),for duty cycles less than 50%the recommended values may be used.The corresponding in-ductor current ripple as shown in Figure 2(a)is given by:The inductor ripple current is important for a few reasons.One reason is because the peak switch current will be the average inductor current (input current or I LOAD /D’)plus ∆i L .As a side note,discontinuous operation occurs when the inductor current falls to zero during a switching cycle,or ∆i L is greater than the average inductor current.Therefore,con-tinuous conduction mode occurs when ∆i L is less than the average inductor current.Care must be taken to make sure that the switch will not reach its current limit during normal operation.The inductor must also be sized accordingly.It should have a saturation current rating higher than the peak inductor current expected.The output voltage ripple is also affected by the total ripple current.The output diode for a boost regulator must be chosen correctly depending on the output voltage and the output current.The typical current waveform for the diode in con-tinuous conduction mode is shown in Figure 2(b).The diode must be rated for a reverse voltage equal to or greater than the output voltage used.The average current rating must be greater than the maximum load current expected,and the peak current rating must be greater than the peak inductor current.During short circuit testing,or if short circuit condi-tions are possible in the application,the diode current rating must exceed the switch current ing Schottky diodes with lower forward voltage drop will decrease power dissipa-tion and increase efficiency.DC GAIN AND OPEN-LOOP GAINSince the control stage of the converter forms a complete feedback loop with the power components,it forms a closed-loop system that must be stabilized to avoid positive feed-back and instability.A value for open-loop DC gain will be required,from which you can calculate,or place,poles and zeros to determine the crossover frequency and the phase margin.A high phase margin (greater than 45˚)is desired for the best stability and transient response.For the purpose of stabilizing the LM2710,choosing a crossover point well be-low where the right half plane zero is located will ensure sufficient phase margin.A discussion of the right half plane zero and checking the crossover using the DC gain will follow.INPUT AND OUTPUT CAPACITOR SELECTIONThe switching action of a boost regulator causes a triangular voltage waveform at the input.A capacitor is required to reduce the input ripple and noise for proper operation of the regulator.The size used is dependant on the application and board layout.If the regulator will be loaded uniformly,with very little load changes,and at lower current outputs,the input capacitor size can often be reduced.The size can also be reduced if the input of the regulator is very close to the source output.The size will generally need to be larger for applications where the regulator is supplying nearly the maximum rated output or if large load steps are expected.A minimum value of 10µF should be used for the less stressful conditions while a 22µF to 47µF capacitor may be required for higher power and dynamic rger values and/or lower ESR may be needed if the application requires very low ripple on the input source voltage.The choice of output capacitors is also somewhat arbitrary and depends on the design requirements for output voltage ripple.It is recommended that low ESR (Equivalent Series Resistance,denoted R ESR )capacitors be used such as ceramic,polymer electrolytic,or low ESR tantalum.Higher ESR capacitors may be used but will require more compen-sation which will be explained later on in the section.The ESR is also important because it determines the peak to peak output voltage ripple according to the approximate equation:∆V OUT )2∆i L R ESR (in Volts)A minimum value of 10µF is recommended and may be increased to a larger value.After choosing the output capaci-tor you can determine a pole-zero pair introduced into the control loop by the following equations:Where R L is the minimum load resistance corresponding to the maximum load current.The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small.If low ESR capacitors are used it can be neglected.If higher ESR capacitors are used see the High Output Capacitor ESR Compensation section.LM271015Operation(Continued)RIGHT HALF PLANE ZEROA current mode control boost regulator has an inherent righthalf plane zero(RHP zero).This zero has the effect of a zeroin the gain plot,causing an imposed+20dB/decade on therolloff,but has the effect of a pole in the phase,subtractinganother90˚in the phase plot.This can cause undesirableeffects if the control loop is influenced by this zero.To ensurethe RHP zero does not cause instability issues,the controlloop should be designed to have a bandwidth of less than1⁄2the frequency of the RHP zero.This zero occurs at a fre-quency of:where I LOAD is the maximum load current.SELECTING THE COMPENSATION COMPONENTSThe first step in selecting the compensation components R Cand C C is to set a dominant low frequency pole in the controlloop.Simply choose values for R C and C C within the rangesgiven in the Introduction to Compensation section to set thispole in the area of10Hz to500Hz.The frequency of the polecreated is determined by the equation:where R O is the output impedance of the error amplifier,approximately1MΩ.Since R C is generally much less thanR O,it does not have much effect on the above equation andcan be neglected until a value is chosen to set the zero f ZC.f ZC is created to cancel out the pole created by the outputcapacitor,f P1.The output capacitor pole will shift with differ-ent load currents as shown by the equation,so setting thezero is not exact.Determine the range of f P1over the ex-pected loads and then set the zero f ZC to a point approxi-mately in the middle.The frequency of this zero is deter-mined by:Now R C can be chosen with the selected value for C C.Check to make sure that the pole f PC is still in the10Hz to500Hz range,change each value slightly if needed to ensureboth component values are in the recommended range.Afterchecking the design at the end of this section,these valuescan be changed a little more to optimize performance ifdesired.This is best done in the lab on a bench,checking theload step response with different values until the ringing andovershoot on the output voltage at the edge of the load stepsis minimal.This should produce a stable,high performancecircuit.For improved transient response,higher values of R Cshould be chosen.This will improve the overall bandwidthwhich makes the regulator respond more quickly to tran-sients.If more detail is required,or the most optimal perfor-mance is desired,refer to a more in depth discussion ofcompensating current mode DC/DC switching regulators.HIGH OUTPUT CAPACITOR ESR COMPENSATIONWhen using an output capacitor with a high ESR value,orjust to improve the overall phase margin of the control loop,another pole may be introduced to cancel the zero createdby the ESR.This is accomplished by adding another capaci-tor,C C2,directly from the compensation pin V C to ground,inparallel with the series combination of R C and C C.The poleshould be placed at the same frequency as f Z1,the ESRzero.The equation for this pole follows:To ensure this equation is valid,and that C C2can be usedwithout negatively impacting the effects of R C and C C,f PC2must be greater than10f ZC.CHECKING THE DESIGNThe final step is to check the design.This is to ensure abandwidth of1⁄2or less of the frequency of the RHP zero.This is done by calculating the open-loop DC gain,A DC.Afterthis value is known,you can calculate the crossover visuallyby placing a−20dB/decade slope at each pole,and a+20dB/decade slope for each zero.The point at which the gain plotcrosses unity gain,or0dB,is the crossover frequency.If thecrossover frequency is less than1⁄2the RHP zero,the phasemargin should be high enough for stability.The phase mar-gin can also be improved by adding C C2as discussed earlierin the section.The equation for A DC is given below withadditional equations required for the calculation:mc)0.072fs(in V/s)where R L is the minimum load resistance,V IN is the maxi-mum input voltage,g m is the error amplifier transconduc-tance found in the Electrical Characteristics table,and R D-SON is the value chosen from the graph"R DSON vs.V IN"inthe Typical Performance Characteristics section.BUFFER(Vcom and GMAx)COMPENSATIONThe architecture used for the buffers in the LM2710requiresexternal compensation on the output.Depending on theequivalent capacitive load of the TFT-LCD panel,externalcomponents at the buffer outputs may or may not be neces-sary.If the capacitance presented by the load is equal to orgreater than5nF no external components are needed as theTFT-LCD panel will act as compensation itself.Distributedresistive and capacitive loads enhance stability and increase LM27116。
TPS40001EVM-001资料
User’s Guide
TPS40001 Based Converter Delivers 10-A Output
User’s Guide
元TICE Texas Instruments (TI) provides the enclosed product(s) under the following conditions: This evaluation kit being sold by TI is intended for use for ENGINEERING DEVELOPMENT OR EVALUATION PURPOSES ONLY and is not considered by TI to be fit for commercial use. As such, the goods being provided may not be complete in terms of required design-, marketing-, and/or manufacturing-related protective considerations, including product safety measures typically found in the end product incorporating the goods. As a prototype, this product does not fall within the scope of the European Union directive on electromagnetic compatibility and therefore may not meet the technical requirements of the directive. Should this evaluation kit not meet the specifications indicated in the EVM User’s Guide, the kit may be returned within 30 days from the date of delivery for a full refund. THE FOREGOING WARRANTY IS THE EXCLUSIVE WARRANTY MADE BY SELLER TO BUYER AND IS IN LIEU OF ALL OTHER WARRANTIES, EXPRESSED, IMPLIED, OR STATUTORY, INCLUDING ANY WARRANTY OF MERCHANTABILITY OR FITNESS FOR ANY PARTICULAR PURPOSE. The user assumes all responsibility and liability for proper and safe handling of the goods. Further, the user indemnifies TI from all claims arising from the handling or use of the goods. Please be aware that the products received may not be regulatory compliant or agency certified (FCC, UL, CE, etc.). Due to the open construction of the product, it is the user’s responsibility to take any and all appropriate precautions with regard to electrostatic discharge. EXCEPT TO THE EXTENT OF THE INDEMNITY SET FORTH ABOVE, NEITHER PARTY SHALL BE LIABLE TO THE OTHER FOR ANY INDIRECT, SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES. TI currently deals with a variety of customers for products, and therefore our arrangement with the user is not exclusive. TI assumes no liability for applications assistance, customer product design, software performance, or infringement of patents or services described herein. Please read the EVM User’s Guide and, specifically, the EVM Warnings and Restrictions notice in the EVM User’s Guide prior to handling the product. This notice contains important safety information about temperatures and voltages. For further safety concerns, please contact the TI application engineer. Persons handling the product must have electronics training and observe good laboratory practice standards. No license is granted under any patent right or other intellectual property right of TI covering or relating to any machine, process, or combination in which such TI products or services might be or are used.
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Ver 1.1
Feb. 24, 2003
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元器件交易网
LM170E01 Liquid Crystal Display
Product Specification
RECORD OF REVISIONS
Revision No Ver 1.0 Ver 1.1 Date Feb. 14. 2003 Feb. 24. 2003 Page 4/28 6/28 18/28 Final draft General feature : Power consumption 3-1 Electrical characteristics Power supply input current Power consumption 4. Optical characteristics Gray scale Description
Ver 1.1
Feb. 24, 2003
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元器件交易网
LM170E01 Liquid Crystal Display
Product Specification 1. General Description
The LM170E01-A4 is a Color Active Matrix Liquid Crystal Display with an integral Cold Cathode Fluorescent Lamp(CCFL) backlight system. The matrix employs a-Si Thin Film Transistor as the active element. It is a transmissive type display operating in the normally white mode. This TFT-LCD has a 17.0 inch diagonal measured active display area with SXGA resolution(1024 vertical by 1280 horizontal pixel array) Each pixel is divided into Red, Green and Blue sub-pixels or dots which are arranged in vertical stripes. Gray scale or the brightness of the sub-pixel color is determined with a 8-bit gray scale signal for each dot, thus, presenting a palette of more than 16.2M colors with FRC(Frame Rate Control). The LM170E01-A4 has been designed to apply the interface method that enables low power, high speed,low EMI. FPD Link or compatible must be used as a LVDS(Low Voltage Differential Signaling) chip. The LM170E01-A4 is intended to support applications where thin thickness,wide viewing angle, low power are critical factors and graphic displays are important. In combination with the vertical arrangement of the sub-pixels, the LM170E01-A4 characteristics provide an excellent flat panel display for office automation products such as monitors.
Ver 1.1
Feb. 24, 2003
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元器件交易网
LM170E01 Liquid Crystal Display
Product Specification 3. Electrical specifications 3-1. Electrical characteristics
Product Specification
CONTENTS
NO. 1 2 3 3-1 3-2 3-3 3-4 3-5 3-6 3-7 4 5 6 7 7-1 7-2 8 8-1 8-2 9 COVER CONTENTS RECORD OF REVISIONS GENERAL DESCRIPTION ABSOLUTE MAXIMUM RATINGS ELECTRICAL SPECIFICATIONS ELECTRICAL CHARACTERISTICS INTERFACE CONNECTIONS SIGNAL TIMING SPECIFICATIONS SIGNAL TIMING WAVEFORMS COLOR INPUT DATA REFERANCE POWER SEQUENCE VCC POWER DIP CONDITION OPTICAL SPECIFICATIONS MECHANICAL CHARACTERISTICS RELIABILITY INTERNATIONAL STANDARDS SAFETY EMC PACKING DESIGNATION OF LOT MARK PACKING FORM PRECAUTIONS ITEM Page 1 2 3 4 5 6 6 8 12 13 14 15 16 17 21 24 25 25 25 26 26 26 27
The LM170E01-A4 requires two power inputs. One is employed to power the LCD electronics and to drive the TFT array and liquid crystal. Another which powers the CCFL, is typically generated by an inverter. The inverter is an external unit to the LCD. Table 2. Electrical Characteristics Parameter MODULE : Power Supply Input Voltage Permissive Power Input Ripple Power Supply Input Current Differential Impedance Power Consumption Rush Current LAMP for each CCFL: Operating Voltage Operating Current Established Starting Voltage at 25 °C at 0 °C Operating Frequency Discharge Stabilization Time Power Consumption Life Time Symbol Values Min. 4.5 90 Typ. 5.0 0.43 100 2.15 2.0 Max. 5.5 0.1 0.50 110 2.50 3.0 Units Notes
元器件交易网
LM170E01 Liquid Crystal Display
Product Specification
Ver 1.1
Feb. 24, 2003
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元器件交易网
LM170E01 Liquid Crystal Display
Table 1. Absolute Maximum Ratings Parameter Power Supply Input Voltage Operating Temperature Storage Temperature Operating Ambient Humidity Storage Humidity Symbol VCC TOP TST HOP HST Values Min. －0.3 0 －20 10 10 Max. ＋5.5 ＋50 ＋60 ＋90 ＋90 Units V dc ℃ ℃ %RH %RH Notes At 25℃ 1 1 1 1
IRUSH VBL IBL VBS f BL TS PBL
VCC VRF ICC Zm PC
90% 60 60% Humidity [(%)RH] 50 Wet Bulb Temperature [C] 40 40% 30 20 10 0 10% 10 20 30 40 50 60 70 80 Storage
Operation
-20
0
Dry Bulb Temperature [C]
Figure 2. Temperature and relative humidity
G1024
TFT-LCD
+5.0V VCC
Power Circuit Block CN2,3 CN4,5
Backlight Assembly(4CCFL)
Figure 1. Block diagram
General Features
Active screen size Outline Dimension Pixel Pitch Pixel Format Display Colors Luminance, white Power Consumption Weight Display operating mode Surface treatments Ver 1.1 17.0 inch (43.27cm) diagonal 358.5(H) x 296.5(V) x 17.0(D) mm(Typ.) 0.264 mm x 0.264 mm 1280 horiz. by 1024 vert. Pixels. RGB stripe arrangement 16.2M colors 250 cd/m2(Typ. Center 1 point) 19.05 Watts(Typ.) 1890g (Typ.) Transmissive mode, normally white Hard coating (3H), Anti-glare treatment of the front polarizer Feb. 24, 2003 4 / 28