LTC电源管理芯片

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电源管理芯片型号

电源管理芯片型号电源管理芯片是一种用于控制和管理电源供应的集成电路，常用于电子设备和计算机系统中。

它能够监测电源电压、电流和温度等参数，以确保电子设备或计算机系统正常工作，并保护设备免受过电压、过电流和过温等不良条件的损害。

电源管理芯片的型号有很多种，下面简单介绍几种常见的型号。

1. MAX77650：这是一款高性能、集成度很高的电源管理芯片。

它具有多种功能，包括锂电池充放电管理、电源管理和系统监测等。

它采用低功耗设计，能够延长电池寿命，同时提供多种省电模式。

2. TPS54160：这是一款高效率、同步降压型电源管理芯片。

它适用于工业和通讯设备，能够提供稳定的电源输出。

它的主要特点是高效率和低纹波，能够满足电子设备对稳定电源的要求。

3. LT3652：这是一款微型化、高效率的电源管理芯片。

它适用于锂电池充电和电源管理。

它采用了开关电源技术，能够提供高效率的电源转换，同时集成了多种保护机制，能够确保电子设备的安全使用。

4. LTC6804：这是一款用于电池管理的芯片。

它可以对电池进行均衡充放电，并能够监测电池的电压、温度和容量等参数。

它采用高精度的ADC技术，能够提供准确的电池状态监测。

5. BQ25895：这是一款专用于充电管理的芯片。

它支持快速充电和逆变充电模式，能够根据不同设备的需求，选择合适的充电模式。

同时，它还具有多种保护机制，能够保护设备免受过充、过放和短路等不良条件的损害。

以上仅是部分电源管理芯片的型号介绍，每一款型号都有自己的特点和应用场合。

随着电子设备的不断发展，电源管理芯片的功能和性能也在不断提高，以满足电子设备对高效、稳定和安全电源供应的需求。

LTC4054丝印LTH7锂电充电管理IC

LTC4054丝印LTH7锂电充电管理IC————————————————————————————————作者：————————————————————————————————日期：2V1.31）典型应用电路注：R1 电阻建议不要省列与 C 1 构成 R C 滤波防止过充电压。

如果 R 1 电阻不接 C 1 使用 10UF 以上电容。

典型运用电路仅供参考，其它以实际运用为准。

概述HX4054A 是一款单节锂离子电池恒流/恒压线性充电器，简单的外部应用电路非常适合便携式设备应用，适合 USB 电源和适配器电源工作，内部采用防倒充电路，不需要外部隔离二极管。

热反馈可对充电电流进行自动调节，以便在大功率操作或高环境温度条件下对芯片温度加以限制。

HX4054A 充电截止电压为 4.2V, 充电电流可通过外部电阻进行设置。

当充电电流降至设定值的 1/10 时，HX4054A 将自动结束充电过程。

当输入电压被移掉后，HX4054A 自动进入低电流待机状态，将待机电流降至 1uA 以下。

HX4054A 在有输入电源时也可置于停机模式，从而将工作电流降至 30uA 。

兼容TP4054 XT4054 LN2054 SE9016 ME4054 XT4052 EUP8054 AS3020 XZ3085A BL8572 HX6001 QX4054 HYM4054 SUN4054 FK5406 AP5056 SLM4054 LP4054 APL3202 OPC8020 OHL5100 LM2054 UCT3054 TQ7051 特点∙ 锂电池正负极反接保护（没充电的情况下 ∙ 最大充电电流：500mA∙ 无需 MOSFET 、检测电阻器和隔离二极管 ∙ 智能热调节功能可实现充电速率最大化 ∙ 智能再充电功能 ∙ 预充电压：4.2V±1% ∙ 3C/10 充电终止 ∙ 待机电流 30uA∙ BAT 超低自耗电 1uA ∙ 2.9V 涓流充电阈值∙ 单独的充电、结束指示灯控制信号 ∙封装形式：SOT23-5应用∙ 手机、PDA 、MP3/MP4 ∙ 蓝牙耳机、GPS ∙ 充电座∙数码相机、Mini 音响等便携式设备V1.32管脚定购信息SOT23-5L封装定购型号包装形式产品正印SOT23-5LHX4054A极限参数（注 1）符号参数额定值单位 VCC 输入电源电压 -0.3~7 V PROG PROG 脚电压 -0.3~0.3 V BAT BAT 脚电压 -0.3~7 V CHRG CHRG 脚电压-0.3~7 V T BAT_SHTBAT 脚短路持续时间连续 - I BAT BAT 脚电流 600 mA I PROG PROG 脚电流 600 uA T OP 工作环境温度 -40~85 ℃ T STG储存温度 -65~125 ℃ ESDHBM2000 V MM200V注 1：最大极限值是指超出该工作范围芯片可能会损坏。

ltc4413原理

ltc4413原理LTC4413是一款用于电源管理的芯片，其原理基于电源选择和电源切换功能。

本文将对LTC4413的工作原理进行详细介绍。

LTC4413芯片是一种电源选择器，用于选择两个不同的电源之间的优先级。

它可以自动监测电源的电压和电流，并根据预设的优先级选择最佳的电源。

这款芯片主要用于电池备份系统、电池充电系统以及其他需要实现电源切换和管理的应用。

LTC4413芯片具有多种保护功能，如过压保护、欠压保护和过流保护等。

当其中一个电源的电压超过或低于预设阈值时，芯片会自动切换到另一个电源，以保护电路和设备的安全运行。

LTC4413芯片的工作原理如下：首先，根据预设的优先级，芯片会自动选择其中一个电源作为主要电源。

然后，它会监测主要电源的电压和电流，以及备用电源的电压。

如果主要电源的电压正常，则芯片会将其输出到负载上。

如果主要电源的电压异常（如过高或过低），芯片会自动切换到备用电源，并将其输出到负载上。

LTC4413芯片还具有电源切换的功能。

当主要电源失效或被拔掉时，芯片会自动切换到备用电源。

这种切换过程是无缝的，可以确保负载的电源持续供应。

当主要电源恢复正常时，芯片会再次自动切换回主要电源，以保持系统的稳定运行。

除了电源选择和切换功能外，LTC4413芯片还具有多种保护功能。

例如，当主要电源的电压超过设定值时，芯片会自动切断电源，以防止电路和设备的损坏。

当备用电源的电流超过设定值时，芯片也会自动切断电源，以保护电路和设备的安全运行。

LTC4413是一款功能强大的电源管理芯片，它可以实现电源选择、切换和保护等多种功能。

通过自动监测电源的电压和电流，并根据预设的优先级选择最佳的电源，该芯片可以确保负载的电源持续供应，同时保护电路和设备的安全运行。

在电池备份系统、电池充电系统以及其他需要实现电源切换和管理的应用中，LTC4413芯片具有广泛的应用前景。

锂电池管理芯片LTC1960

1234SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS t ONPI Gate B1I/B2I/DCI Turn-On Time V GS < –3V, C LOAD = 3nF (Note 5)300µs t OFFPI Gate B1I/B2I/DCI Turn-Off Time V GS > –1V, C LOAD = 3nF (Note 5)10µs V PONI Input Gate Clamp Voltage I LOAD = 1µAGB1I Highest (V BAT1 or V SCP) – V GB1I 4.75 6.77.5VGB2I Highest (V BAT2 or V SCP) – V GB2I 4.75 6.77.5VGDCI Highest (V DCIN or V SCP) – V GDCI 4.75 6.77.5V V POFFI Input Gate Off Voltage I LOAD = –25µAGB1I Highest (V BAT1 or V SCP) – V GB1I0.180.25VGB2I Highest (V BAT2 or V SCP) – V GB2I0.180.25VGDCI Highest (V DCIN or V SCP) – V GDCI0.180.25V Logic I/OI IH/I IL SSB/SCK/MOSI Input High/Low Current●–11µA V IL SSB/MOSI/SCK Input Low Voltage●0.8V V IH SSB/MOSI/SCK Input High Voltage●2V V OL MISO Output Low Voltage I OL = 1.3mA●0.4V I OFF MISO Output Off-State Leakage Current V MISO = 5V●2µA SPI Timing (See Timing Diagram)T WD Watch Dog Timer● 1.2 2.5 4.5sec t SSH SSB High Time680ns t CYC SCK Period C LOAD = 200pF R PULLUP = 4.7k on MISO●2µs t SH SCK High Time680ns t SL SCK Low Time680ns t LD Enable Lead Time200ns t LG Enable Lag Time200ns t su Input Data Set-Up Time●100ns t H Input Data Hold Time●100ns t A Access Time (From Hi-Z to Data Active on MISO)●125ns t dis Disable Time (Hold Time to Hi-Z State on MISO)●125ns t V Output Data Valid C L = 200pF, R PULLUP = 4.7k on MISO●580ns t HO Output Data Hold●0ns t Ir SCK/MOSI/SSB Rise Time0.8V to 2V250ns t If SCK/MOSI/SSB Fall Time2V to 0.8V250ns t Of MISO Fall Time2V to 0.4V, C L = 200 pF●400nsNote 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime.Note 2. Battery voltage must be adequate to drive gates of PowerPath P-channel FET switches. This does not affect charging voltage of the battery, which can be zero volts.Note 3. See Test Circuit.Note 4. DCIN, BAT1, BAT2 are held at 12V and GDCI, GB1I, GB2I are forced to 10.5V. SCP is set at 12.0V to measure source current at GDCI,GB1I and GB2I. SCP is set at 11.9V to measure sink current at GDCI, GB1I and GB2I.Note 5. Extrapolated from testing with C L = 50pF.Note 6. VDAC offset is equal to the reference voltage, sinceV OUT = V REF(16mV • VDAC(VALUE)/2047 + 1).Note 7. The LTC1960C is guaranteed to meet specified performance from 0°C to 70°C and is designed, characterized and expected to meet specified performance at –40°C and 85°C, but is not tested at these extended temperature limits.ELECTRICAL CHARACTERISTICS The ●denotes specifications which apply over the full operating temperature range (Note 7), otherwise specifications are at T A = 25°C. V DCIN = 20V, V BAT1 = 12V, V BAT2 = 12V unless otherwise noted.51960fa6781960faInput Power RelatedSCN (Pin 4/Pin 30): PowerPath Current Sensing Negative Input. This pin should be connected directly to the “bot-tom” (output side) of the low valued resistor in series with the three PowerPath switch pairs, for detecting short-circuit current events. Also powers LTC1960 internal circuitry when all other sources are absent.SCP (Pin 5/Pin 31): PowerPath Current Sensing Positive Input. This pin should be connected directly to the “top”(switch side) of the low valued resistor in series with the three PowerPath switch pairs, for detecting short-circuit current events.GDCO (Pin 6/Pin 32): DCIN Output Switch Gate Drive.Together with GD CI, this pin drives the gate of the P-channel switch in series with the DCIN input switch.GDCI (Pin 7/Pin 33): D CIN Input Switch Gate D rive.Together with GDCO, this pin drives the gate of the P-channel switch connected to the DCIN input.GB1O (Pin 8/Pin 34): BAT1 Output Switch Gate Drive.Together with GB1I, this pin drives the gate of the P-channel switch in series with the BAT1 input switch.GB1I (Pin 9/Pin 35): BAT1 Input Switch Gate D rive.Together with GB1O, this pin drives the gate of the P-channel switch connected to the BAT1 input.GB2O (Pin 10/Pin 36): BAT2 Output Switch Gate Drive.Together with GB2I, this pin drives the gate of the P-channel switch in series with the BAT2 input switch.GB2I (Pin 11/Pin 37): BAT2 Input Switch Gate D rive.Together with GB2O, this pin drives the gate of the P-channel switch connected to the BAT2 input.CLP (Pin 24/Pin 13): This is the Positive Input to the Supply Current Limiting Amplifier CL1. The threshold is set at 100mV above the voltage at the DCIN pin. When used to limit supply current, a filter is needed to filter out the switching noise.Battery Charging RelatedV SET (Pin 13/Pin 1): The Tap Point of a Programmable Resistor Divider which Provides Battery Voltage Feedback to the Charger. A capacitor from CSN to V SET and one from V SET to GND provide necessary compensation and filter-ing for the voltage loop.I TH (Pin 14/Pin 2): This is the Control Signal of the Inner Loop of the Current Mode PWM. Higher I TH corresponds to higher charging current in normal operation. A capaci-tor of at least 0.1µF to GND filters out PWM ripple. Typical full-scale output current is 30µA. Nominal voltage range for this pin is 0V to 2.4V.I SET (Pin 15/Pin 3): A Capacitor from I SET to Ground is Required to Filter Higher Frequency Components from the Delta-Sigma IDAC.CSN (Pin 22/Pin 11): Current Amplifier CA1 Input. Con-nect this to the common output of the charger MUX switches.CSP (Pin 23/Pin 12): Current Amplifier CA1 Input. This pin and the CSN pin measure the voltage across the sense resistor, RSNS, to provide the instantaneous current signals required for both peak and average current mode operation.COMP1 (Pin 25/Pin 14): This is the Compensation Node for the Amplifier CL1. A capacitor is required from this pin to GND if input current amplifier CL1 is used. At input adapter current limit, this node rises to 1V. By forcing COMP1 low, amplifier CL1 will be defeated (no adapter current limit). COMP1 can source 10µA.BGATE (Pin 27/Pin 16): D rives the Bottom External MOSFET of the Battery Charger Buck Converter.SW (Pin 30/Pin 19): Connected to Source of Top External MOSFET Switch. Used as reference for top gate driver.BOOST (Pin 31/Pin 20): Supply to Topside Floating Driver.The bootstrap capacitor is returned to this pin. Voltage swing at this pin is from a diode drop below V CC to (DCIN + V CC ).PI FU CTIO SU U U(G/UHF)91960faTGATE (Pin 32/Pin 21): Drives the Top External MOSFET of the Battery Charger Buck Converter.SCH1 (Pin 33/Pin 22), SCH2 (Pin 36/Pin 25): Charger MUX Switch Source Returns. These two pins are con-nected to the sources of Q3/Q4 and Q9/Q10 (see Typical Application on back page of data sheet), respectively. A small pull-down current source returns these nodes to 0V when the switches are turned off.GCH1 (Pin 34/Pin 23), GCH2 (Pin 35/Pin 24): Charger MUX Switch Gate Drives. These two pins drive the gates of the back-to-back N-channel switch pairs, Q3/Q4 and Q9/Q10, between the charger output and the two batteries.External Power Supply PinsV PLUS (Pin 1/Pin 27): Supply. The V PLUS pin is connected via four internal diodes to the DCIN, SCN, BAT1, and BAT2 pins. Bypass this pin with a 1µF to 2µF capacitor.BAT1 (Pin 3/Pin 29), BAT2 (Pin 2/Pin 28): These two pins are the inputs from the two batteries for power to the LTC1960 and to provide voltage feedback to the battery charger.LOPWR (Pin 12/Pin 38): LOPWR Comparator Input from External Resistor Divider Connected from SCN to GND. If the voltage at LOPWR is lower than the LOPWR com-parator threshold, then system power has failed and power is autonomously switched to a higher voltage source, if available. See PowerPath section of LTC1960operation.DCDIV (Pin 17/Pin 5): DCDIV Comparator Input from External Resistor D ivider Connected from D CIN to GND . If the voltage at D CD IV is above the D CD IV comparator threshold, then the D C bit is set and the wall adapter power is considered to be adequate to charge the batteries. If DCDIV is taken more than 1.8Vabove V CC , then all of the power path switches are latched off until all power is removed.DCIN (Pin 29/Pin 18): Supply. External DC power source.A 1µF bypass capacitor should be connected to this pin as close as possible. No series resistance is allowed,since the adapter current limit comparator input is also this pin.Internal Power Supply PinsGND (Pin 16/Pin 4, Pin 10, Pin 26, Pin 39): Ground for Low Power Circuitry.PGND (Pin 26/Pin 15): High Current Ground Return for BGATE Driver.V CC (Pin 28/Pin 17): Internal Regulator Output. Bypass this output with at least a 2µF to 4.7µF capacitor. Do not use this regulator output to supply more than 1mA to external circuitry.Digital Interface PinsSSB (Pin 18/Pin 6): SPI Slave Select Input. Active low.TTL levels. This signal is low when clocking data to/from the LTC1960.SCK (Pin 19/Pin 7): Serial SPI Clock. TTL levels.MISO (Pin 20/Pin 8): SPI Master-In-Slave-Out Output,Open Drain. Serial data is transmitted from the LTC1960,when SSB is low, on the falling edge of SCK. TTL levels.A 4.7k pullup resistor is recommended.MOSI (Pin 21/Pin 9): SPI Master-Out-Slave-In Input.Serial data is transmitted to the LTC1960, when SSB is low, on the rising edge of SCK. TTL levels.Exposed Pad (Pin 39, UHF Package Only): Ground.Must be soldered to the PCB ground for rated thermal performance.PI FU CTIO SU U U(G/UHF)10OVERVIEWThe LTC1960 is composed of a battery charger controller, charge MUX controller, PowerPath controller, SPI inter-face, a 10-bit current DAC (IDAC) and 11-bit voltage DAC (VDAC). When coupled with a low cost microprocessor, it forms a complete battery charger/selector system for two batteries. The battery charger is programmed for voltage and current, and the charging battery is selected via the SPI interface. Charging can be accomplished only if the voltage at DCDIV indicates that sufficient voltage is avail-able from the input power source, usually an AC adapter. The charge MUX, which selects the battery to be charged, is capable of charging both batteries simultaneously by selecting both batteries for charging. The charge MUX switch drivers are configured to allow charger current to share between the two batteries and to prevent current from flowing in a reverse direction in the switch. The amount of current that each battery receives will depend upon the relative capacity of each battery and the battery voltage. This can result in significantly shorter charging times (up to 50% for Li-Ion batteries) than sequential charging of each battery. In order to continue charging, the CHARGE_BAT information must be updated more frequently than the internal watchdog timer.The PowerPath controller selects which of the pairs of PFET switches, input and output, will provide power to the system load. The selection is accomplished over the SPI interface. If the system voltage drops below the threshold set by the LOPWR resistor divider, then all of the output side PFETs are turned on quickly and power is taken from the highest voltage source available at the DCIN, BAT1 or BAT2 inputs. The input side PFETs act as diodes in this mode and power is taken from the source with the highest voltage. The input side PowerPath switch driver that is delivering power then closes its input switch to reduce the power dissipation in the PFET bulk diode. In effect, this system provides diode -like behavior from the FET switches, without the attendant high power dissipation from diodes. The microprocessor is informed of this 3-diode mode status when it polls the PowerPath status register via the SPI interface. The microprocessor can then assess which power source is capable of providing power, and program the PowerPath switches accordingly. Since high speed PowerPath switching at LOPWR trip points is handled autonomously, there is no need for real-time microproces-sor resources to accomplish this task. Simultaneous discharge of both batteries is accomplished by simply programming both batteries for discharge into the system load. The switch drivers prevent reverse cur-rent flow in the switches and automatically discharge both batteries into the load, sharing current according to the relative capacity of the batteries. Simultaneous dual dis-charge can increase battery operating time by approxi-mately 10% by reducing losses in the switches and reducing internal losses associated with high discharge rates. SPI InterfaceThe SPI interface is used to write to the internal PowerPath registers, the charger control registers, the current DAC, and the voltage DAC. The SPI is also able to read internal status registers. There are two types of SPI write com-mands. The first write command is a 1-byte command used to load PowerPath and charger control bits. The second write command is a 2-byte command used to load the DACs. The SPI read command is a 2-byte command. In order to ensure the integrity of the SPI communication, the last bit received by the SPI is echoed back over the MISO output after the next falling SCK. The data format is set up so that the master has the option of aborting a write if the returned MISO bit is not as expected.1-Byte SPI Write Format:bit 7........byte 1..........bit 0 MOSI D0 D1 D2 X A0 A1 A2 0 MISO X D0 D1 D2 X A0 A1 A2 Charger Write Address:A[2:0] = b111Charger Write Data:D2 = XD1 = CHARGE_BAT2D0 = CHARGE_BAT1PowerPath Write Address:A[2:0] = b110PowerPath Write Data:D2 = POWER_BY_DCD1 = POWER_BY_BAT2D0 = POWER_BY_BAT1OPERATIO(Refer to Block Diagram and Typical Application)111960fa121960fa2-Byte SPI Write Format:bit 7........byte 1..........bit 0 bit 7..........byte 2............bit 0MOSI D0 D1 D2 D3 D4 D5 D6 1 D7 D8 D9 D10 A0 A1 A2 0MISOX D0 D1 D2 D3 D4 D5 D6 1 D7 D8 D9 D10 A0 A1 A2IDAC Write Address:A[2:0] = b000IDAC Data Bits D9-D0:IDAC value data (MSB-LSB)IDAC Data Bit D10 :Normal mode = 0, low current mode = 1 (Dual battery charging is disabled)VDAC Write Address:A[2:0] = b001VDAC Data Bits D10-D0:VDAC value (MSB-LSB)Subsequent SPI communication is inhibited until after the addressed DAC is finished loading. It is recommended that the master transmit all zeros until MISO goes low. This handshaking procedure is illustrated in Figure 1.2-Byte SPI Read Format:bit 7........byte 1.......bit 0bit 7........byte 2............bit 0MOSI 0 0 0 0 A0 A1 A2 0 0 0 0 0 A0 A1 A2 1MISO X 0 0 0 0 A0 A1 A2 X FA LP DC PF CH X XStatus Address:A[2:0] = b010Status Read Data:LP = LOW_POWER (Low power comparator output)DC = DCDIV (DCDIV comparator output)PF = POWER_FAIL (Set if selected power supply failed to hold up system power after three tries)CH = CHARGING (One or more batteries are being charged)FA = FAULT. This bit is set for any of the following conditions:1) The LTC1960 is still in power on reset.2) The LTC1960 has detected a short circuit and has shut down power and charging.3) The system has asserted a fast off using DCDIV.Figure 1. SPI Write to VDAC of Data = b101_0101_0101Note: All other values of A[2:0] are reserved and must not be used.OPERATIOSSBSCKMOSIMISO1960 F01BYTE 1BYTE 2Figure 2. SPI Read of FA = 0, LP = 0, DC = 1, PF = 0, and CH = 1Battery Charger ControllerThe LTC1960 charger controller uses a constant off-time, current mode step-down architecture. During normal op-eration, the top MOSFET is turned on each cycle when the oscillator sets the SR latch and turned off when the main current comparator I CMP resets the SR latch. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current reverses, as indicated by cur-rent comparator IREV, or the beginning of the next cycle. The oscillator uses the equation:t OFF = 1/f OSC • (V DCIN – V CSN)/V DCINto set the bottom MOSFET on time. The peak inductor current at which ICMP resets the SR latch is controlled by the voltage on I TH. I TH is in turn controlled by several loops, depending upon the situation at hand. The average current control loop converts the voltage between CSP and CSN to a representative current. Error amp CA2 compares this current against the desired current requested by the IDAC at the I SET pin and adjusts I TH until the ID AC value is satisfied. The BAT1/BAT2 MUX provides the selected battery voltage at CHGMON, which is divided down to the V SET pin by the VDAC resistor divider and is used by error amp EA to decrease I TH if the V SET voltage is above the 0.8V reference. The amplifier CL1 monitors and limits the input current, normally from the AC adapter, to a preset level (100mV/R CL). At input current limit, CL1 will decrease the I TH voltage and thus reduce battery charging current. An overvoltage comparator, 0V, guards against transient overshoots (>7%). In this case, the top MOSFET is turned off until the overvoltage condition is cleared. This feature is useful for batteries which “load dump”themselves by opening their protection switch to perform functions such as calibration or pulse mode charging.Charging is inhibited for battery voltages below the mini-mum charging threshold, V CHMIN. Charging is not inhib-ited when the low current mode of the IDAC is selected. The top MOSFET driver is powered from a floating boot-strap capacitor C B. This capacitor is normally recharged from V CC through an external diode when the top MOSFET is turned off. A 2µF to 4.7µF capacitor across V CC to GND is required to provide a low dynamic impedance to charge the boost capacitor. It is also required for stability and power-on-reset purposes.As V IN decreases towards the selected battery voltage, the converter will attempt to turn on the top MOSFET continu-ously (“dropout’’). A dropout timer detects this condition and forces the top MOSFET to turn off, and the bottom MOSFET on, for about 200ns at 40µs intervals to recharge the bootstrap capacitor.Charge MUX SwitchesThe equivalent circuit of a charge MUX switch driver is shown in Figure 3. If the charger controller is not enabled, the charge MUX drivers will drive the gate and source of the series connected MOSFETs to a low voltage and the switch is off. When the charger controller is on, the charge MUX driver will keep the MOSFETs off until the voltage at CSN rises at least 35mV above the battery voltage. GCH1 is then driven with an error amplifier EAC until the voltage between BAT1 and CSN satisfies the error amplifier or until GCH1 is clamped by the internal Zener diode. The time required to close the switch could be quite long (many ms) due to the small currents output by the error amp and depending upon the size of the MOSFET switch.If the voltage at CSN decreases below V BAT1 – 20mV a comparator CC quickly turns off the MOSFETs to preventOPERATIOA status read is illustrated in Figure 2.SSBSCKMOSIMISOBYTE 1BYTE 2131960fa14151617181960faWatchdog TimerCharging will begin when either CHARGE_BAT1 or CHARGE_BAT2 bits are set in the charger register (address: 111). Charging will stop if the charger register is not updated prior to the expiration of the watchdog timer.Simply repeating the same data transmission to the charger register at a rate higher than once per second will ensure that charging will continue uninterrupted.Extending System to More than 2 BatteriesThe LTC1960 can be extended to manage systems with more than 3 sources of power. Contact Linear Technology Applications Engineering for more information.Charging Depleted BatteriesSome batteries contain internal protection switches that disconnect a load if the battery voltage falls below what is considered a reasonable minimum. In this case, the charger may not start because the voltage at the battery terminal is less than 5V. The low current mode of the IDAC must be used in this case to condition the battery. In low current mode, there is no minimum voltage requirement (but dual charging is not allowed). Usually, the battery will detect that it is being charged and then close its protection switch, which will allow the ID AC to switch to normal mode. Smart batteries require that charging current not exceed 100mA until valid charging voltage and charging current parameters are transmitted via the SMBus. The low current IDAC mode is ideal for this purpose.Starting Charge with Dissimilar Batteries in Dual Charge ModeWhen charging batteries of different charger termination voltages, the charger should be started using the following procedure:Step 1. Select only the lowest termination voltage battery for charging, and set the charger to its charging param-eters.Step 2. When the battery current is flowing into that battery, change to dual charging mode (without stopping the charger) and set the appropriate charging parameters for this dual charger condition.APPLICATIO S I FOR ATIO W UU U If this procedure is not followed, and BAT2 is significantly higher voltage than BAT1, the charger could refuse to charge either battery.Charge Termination IssuesBatteries with constant-current charging and voltage-based charger termination might experience problems with reductions of charger current caused by adapter limiting. It is recommended that input limiting feature be defeated in such cases. Consult the battery manufacturer for information on how your battery terminates charging.Setting Output Current LimitThe full scale output current setting of the ID AC will produce V MAX = 102.3mV between CSP and CSN. To set the full scale current of the DAC simply divide V MAX by R SNS .This is expressed by the following equation:R SNS = 0.1023/I MAXTable 1. Recommended R SNS Resistor ValuesI MAX (A)R SNS (Ω) 1%R SNS (W)1.0230.1000.252.0460.0500.254.0920.0250.58.1840.0121Use resistors with low ESL.Inductor SelectionHigher operating frequencies allow the use of smaller inductor and capacitor values. A higher frequency gener-ally results in lower efficiency because of MOSFET gate charge losses. In addition, the effect of inductor value on ripple current and low current operation must also be considered. The inductor ripple current ∆I L decreases with higher frequency and increases with higher V IN .∆I f L V V V L OUT OUT IN =()()−⎛⎝⎜⎞⎠⎟11Accepting larger values of ∆I L allows the use of low inductances, but results in higher output voltage ripple191960faand greater core losses. A reasonable starting point for setting ripple current is ∆I L = 0.4(I MAX ). In no case should ∆I L exceed 0.6(I MAX ) due to limits imposed by IREV and CA1. Remember the maximum ∆I L occurs at the maxi-mum input voltage. In practice 10µH is the lowest value recommended for use.Charger Switching Power MOSFET and Diode SelectionTwo external power MOSFETs must be selected for use with the LTC1960 charger: An N-channel MOSFET for the top (main) switch and an N-channel MOSFET for the bottom (synchronous) switch.The peak-to-peak gate drive levels are set by the V CC voltage. This voltage is typically 5.2V. Consequently, logic-level threshold MOSFETs must be used. Pay close atten-tion to the B VDSS specification for the MOSFETs as well;many of the logic level MOSFETs are limited to 30V or less.Selection criteria for the power MOSFETs include the “ON”resistance R DS(ON), reverse transfer capacitance C RSS ,input voltage and maximum output current. The LTC1960charger is always operating in continuous mode so the duty cycles for the top and bottom MOSFETs are given by:Main Switch Duty Cycle = V OUT /V INSynchronous Switch Duty Cycle = (V IN – V OUT )/V IN The MOSFET power dissipations at maximum output current are given by:P MAIN = V OUT /V IN (I MAX )2(1 + δ∆Τ)R DS(ON) + k(V IN )2(I MAX )(C RSS )(f)P SYNC = (V IN – V OUT )/V IN (I MAX )2(1 + δ∆Τ) R DS(ON)Where δ∆Τ is the temperature dependency of R DS(ON) and k is a constant inversely related to the gate drive current.Both MOSFETs have I 2R losses while the topside N-channel equation includes an additional term for transi-tion losses, which are highest at high input voltages. For V IN < 20V the high current efficiency generally improves with larger MOSFETs, while for V IN > 20V the transition losses rapidly increase to the point that the use of a higher R DS(ON) device with lower C RSS actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage or during a short-circuit whenAPPLICATIO S I FOR ATIO W UU U the duty cycle in this switch is nearly 100%. The term (1 + δ∆Τ) is generally given for a MOSFET in the form of a normalized R DS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs.C RSS is usually specified in the MOSFET characteristics.The constant k = 1.7 can be used to estimate the contribu-tions of the two terms in the main switch dissipation equation.If the LTC1960 charger is to operate in low dropout mode or with a high duty cycle greater than 85%, then the topside N-channel efficiency generally improves with a larger MOSFET. Using asymmetrical MOSFETs may achieve cost savings or efficiency gains.The Schottky diode D1, shown in the Typical Application on the back page, conducts during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on and storing charge during the dead-time, which could cost as much as 1% in efficiency. A 1A Schottky is generally a good size for 4A regulators due to the relatively small average current. Larger diodes can result in additional transition losses due to their larger junction capacitance.The diode may be omitted if the efficiency loss can be tolerated.Calculating I C Power DissipationThe power dissipation of the LTC1960 is dependent upon the gate charge of Q TG and Q BG .(Refer to Typical Application). The gate charge is determined from the manufacturer’s data sheet and is dependent upon both the gate voltage swing and the drain voltage swing of the FET.P D = (V DCIN – V VCC )([f OSC (Q TG + Q BG ) + I VCC ]+ V DCIN • I DCIN )Example: V VCC = 5.2V, V DCIN = 19V, f OSC = 345kHz,Q G2 = Q G3 = 15nC, I VCC = 0mA.P D = 165mW V SET /I SET CapacitorsCapacitor C7 is used to filter the delta-sigma modulation frequency components to a level which is essentially DC.Acceptable voltage ripple at ISET is about 10mV P-P . Since the period of the delta-sigma switch closure, T ∆Σ, is about201960fa10µs and the internal IDAC resistor, R SET , is 18.77k, the ripple voltage can be approximated by:∆∆V V T R C ISET REF SET =∑••7Then the equation to extract C7 is: C V T V R REF ISET SET7=∑••∆∆= 0.8/0.01/18.77k(10µs) ≅ 0.043µFIn order to prevent overshoot during start-up transients the time constant associated with C7 must be shorter than the time constant of C5 at the I TH pin. If C7 is increased to improve ripple rejection, then C5 should be increased proportionally and charger response time to average cur-rent variation will degrade.Capacitor C B1 and C B2 are used to filter the VDAC delta-sigma modulation frequency components to a level which is essentially DC. C B2 is the primary filter capacitor and CB1 is used to provide a zero in the response to cancel the pole associated with C B2. Acceptable voltage ripple at V SET is about 10mV P-P . Since the period of the delta-sigma switch closure, T ∆Σ, is about 11µs and the internal VDAC resistor, R VSET , is 7.2k Ω, the ripple voltage can be ap-proximated by:∆∆V V T R C C VSET REF VSET B B =()∑•||12Then the equation to extract C B1 || C B2 is:C C V T R V B B REF VSET VSET12||•=∑∆∆C B2 should be 10× to 20× C B1 to divide the ripple voltage present at the charger output. Therefore C B1 = 0.01µF and C B2 = 0.1µF are good starting values. In order to prevent overshoot during start-up transients the time constant associated with C B2 must be shorter than the time constant of C5 at the I TH pin. If C B2 is increased to improve ripple rejection, then C5 should be increased proportionally and charger response time to voltage variation will degrade.Input and Output CapacitorsIn the 4A Lithium Battery Charger (Typical Application section), the input capacitor (C IN ) is assumed to absorb all input switching ripple current in the converter, so it must have adequate ripple current rating. Worst-case RMS ripple current will be equal to one half of output charging current. Actual capacitance value is not critical. Solid tantalum low ESR capacitors have high ripple current rating in a relatively small surface mount package, but caution must be used when tantalum capacitors are used for input or output bypass . High input surge currents can be created when the adapter is hot-plugged to the charger or when a battery is connected to the charger. Solid tantalum capacitors have a known failure mechanism when subjected to very high turn-on surge currents. Only Kemet T495 series of “Surge Robust” low ESR tantalums are rated for high surge conditions such as battery to ground.The relatively high ESR of an aluminum electrolytic for C15, located at the AC adapter input terminal, is helpful in reducing ringing during the hot-plug event.Highest possible voltage rating on the capacitor will mini-mize problems. Consult with the manufacturer before use.Alternatives include new high capacity ceramic (at least 20µF) from Tokin, United Chemi-Con/Marcon, et al. Other alternative capacitors include OSCON capacitors from Sanyo.The output capacitor (C OUT ) is also assumed to absorb output switching current ripple. The general formula for capacitor current is:I RMS = (L1)(f)V BATV DCIN()0.29 (V BAT ) 1 –For example:V DCIN = 19V, V BAT = 12.6V, L1 = 10µH, and f = 300kHz, I RMS = 0.41A.EMI considerations usually make it desirable to minimize ripple current in the battery leads, and beads or inductors may be added to increase battery impedance at the 300kHzAPPLICATIO S I FOR ATIO W UU U。

数字电源及其典型器件(六)——LTC系列数字电源管理芯片

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理控制器ＬＴＣ２９７８的结构、特点和功能。
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文章编号：０２１９－２７１３（２０１３）０３－００７２ — ５
１ＬＴＣ系列数字电源管理ＩＣ概述
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ltc3789芯片中文资料及推荐参数

ltc3789芯片中文资料及推荐参数LTC3789 是一款高性能、降压-升压型开关稳压控制器，可以在输入电压高于、低于或等于输出电压的情况下运作。

该器件运用了恒定频率、电流模式架构，故可提供一个高达600kHz 的可锁相频率，而一个输出电流反馈环路则提供了对电池充电的支持。

凭借4V 至38V （最大值为40V）的宽输入和输出范围以及工作区之间的无缝和低噪声转换，LTC3789 成为了汽车、电信和电池供电型系统的理想选择。

在降压或升压模式时，LTC3789 运用专有电流模式控制架构实现恒定频率工作，而且内置了4 个强大的N 沟道MOSFET 栅极驱动器。

LTC3789 还提供一个准确的恒定电流调节环路，用于在宽输入电压范围内调节输入或输出电流。

输入电流限制功能防止输入电源过载，输出电流限制为诸如电池充电器或LED 驱动器等稳定输出电流应用提供了非常容易的解决方案。

该器件在所有工作模式下为过压、过流和短路情况提供了故障保护。

此外，LTC3789 在停机时断开输入电压和输出电压的连接。

用户可以选择连续或脉冲跳跃模式，以最大限度地提高轻负载效率，并允许将IC 同步至一个外部时钟。

脉冲跳跃模式在轻负载条件下可实现最低的纹波，而强制连续模式则工作于一个恒定的频率以满足噪声敏感型应用的需要。

此外，LTC3789 具有可调软启动、电源良好输出，并在-40C 至125C 的工作结温范围内保持1.5% 的基准电压准确度。

当输出位于其设计设定点的10% 以内时，一个电源良好输出引脚将发生指示信号。

LTC3789 采用扁平的28 引脚4mm x 5mm QFN 封装和窄体SSOP 封装。

LTC3789引脚说明LTC3789引脚图1. VFB（PIN1/PIN26）：误差放大器反馈引脚。

LTC3789收到的反馈电压来自于外部电阻分压器输出的电压。

2. SS（PIN2/PIN27）：外部软启动输入引脚。

LTC3789调节VFB电压到较小的0.8V或SS 引脚上的电压。

ltc2954的用法

ltc2954的用法LTC2954是一款高性能电源管理芯片，广泛应用于各种电子设备中。

它具有多种功能，可以实现电源开关、电源监控和电源管理等功能，为电子设备的稳定运行提供了重要保障。

首先，LTC2954可以实现电源开关功能。

在很多电子设备中，为了节省能源和延长电池寿命，需要在设备不使用时自动关闭电源。

LTC2954可以通过控制外部开关管，实现电源的开关功能。

当设备不使用时，LTC2954会自动关闭电源，避免电池能量的浪费。

而当设备需要使用时，LTC2954会自动打开电源，保证设备正常运行。

其次，LTC2954还可以实现电源监控功能。

在电子设备中，电源的稳定性对设备的正常运行至关重要。

LTC2954可以监测电源的电压和电流，并及时反馈给控制系统。

当电源电压或电流异常时，LTC2954会发出警报信号，提醒用户或控制系统进行处理。

这样可以避免因电源问题导致的设备故障或损坏，保证设备的稳定运行。

此外，LTC2954还具有电源管理功能。

在一些特殊应用中，需要对电源进行精确控制，以满足设备的特定需求。

LTC2954可以通过外部控制信号，实现对电源的精确控制。

用户可以根据需要，通过控制LTC2954的工作模式和参数，实现对电源的灵活管理。

这样可以满足不同设备的特定需求，提高设备的性能和可靠性。

总之，LTC2954是一款功能强大的电源管理芯片，广泛应用于各种电子设备中。

它可以实现电源开关、电源监控和电源管理等多种功能，为设备的稳定运行提供了重要保障。

通过LTC2954的应用，可以有效节省能源、延长电池寿命，提高设备的性能和可靠性。

未来，随着科技的不断发展，LTC2954的用途将会更加广泛，为电子设备的发展带来更多的可能性。

LTC电源管理芯片

里被停用。一旦某个系统完成了其断电操作，则其将能够把 KILL 引脚设定为低电平 ( 因而立即关断电源 )，从而终止内部 1024ms 定时器。图 1 示出了采用 EN 引脚来控制用于系统主电源的 DC/ DC 转换器情形。EN 输出为开路漏极；LTC2950-1 为正逻辑器件，而 LTC2950-2 为负逻辑器件。INT 和 KILL 信号利用微处理器来进行通信。 LTC2950-2 的 EN 漏极开路输出是专为 10V 电压而设计的，因此能够对一个外部 P 沟道功率 MOSFET 进行接通 / 关断操作。图 3 示出了在一个两节锂离子电池应用中采用
LTC2951
� 10kV 128ms 32ms � 128ms � 6µA 2mm x 3mm DFN TSOT-8
上电序列。缺省延迟为 32ms，而且，外部 ONT 电容器使得用户能够配置更长的延迟。一旦 EN 引脚电平走高，则起动一个内部 512ms 消隐定时器。该消隐定时器是专为给 DC/DC 转换器达到其最终电压以及微处理器执行上电操任务而提供足够的时间而设计。KILL 引脚必须在 EN 引脚走高的 512ms 时间内被拉至高电平，否则 DC/DC 转换器将被停用。PB 引脚上的下一个变换启动了断电序列。同样，32ms 的缺省延迟以及外部 OFFT 电容器使得用户能够配置更长的延迟。在该时点上将设置一个中断，以表示 DC/ DC 转换器将在 1024ms 的时间
图 2：LTC2950 时序图

图 3：用于电源通路控制的 P 沟道 MOSFET 按钮控制
图 1：DC/DC 降压型转换器的按钮控制表 1：LTC2950/LTC2951 特点对比

ltc2950的原理和应用

LTC2950的原理和应用1. LTC2950简介LTC2950是一款轻型电源管理芯片，适用于各种电池供电系统的自动开关功能。

它基于CMOS工艺，具有低功耗和高稳定性的特点。

本文将介绍LTC2950的原理和应用。

2. LTC2950的工作原理LTC2950采用了一种称为监视器模式的工作原理，其主要功能是监视系统电源电压并控制开关输出。

下面是LTC2950的工作原理：•输入电源电压监测：LTC2950能够监测输入电源的电压，并根据设定的电压阈值来确定系统是否正常工作。

当输入电压低于阈值时，LTC2950将触发系统关闭动作。

•开关输出控制：LTC2950有一个开关输出，用于控制其他外部设备的开关。

当系统运行正常时，开关输出保持开启状态；当输入电压低于阈值时，开关输出将关闭。

•自动开启恢复：LTC2950具有自动开启恢复功能，即当输入电压回复到正常范围内时，开关输出将自动开启，系统将恢复正常工作状态。

3. LTC2950的应用LTC2950的原理和功能使其在电池供电系统中具有广泛的应用。

下面是几个常见的应用场景：3.1 电池供电系统的自动开关LTC2950可以作为电池供电系统的自动开关，当电池电压低于设定阈值时，LTC2950将关闭系统以避免过放电，从而保护电池的寿命。

当电压回复正常时，LTC2950将自动开启系统，实现自动恢复。

3.2 低功耗电源管理由于LTC2950本身具有低功耗特点，可以提供给其他电路使用作为电源管理芯片。

它可以监控输入电压，根据需求控制开关输出，实现低功耗的电源管理功能。

3.3 电池充电系统LTC2950可以作为电池充电系统的控制芯片使用。

它可以监控充电电压和电流，并根据需求控制充电器的开关状态。

当充电电压和电流达到设定值时，LTC2950可以触发停止充电动作，从而保护电池的安全。

3.4 电源切换系统LTC2950还可以应用于电源切换系统中，当主电源发生故障或不稳定时，LTC2950可以自动切换到备用电源以保证系统的正常运行。

ltc2954的用法

LTC2954的用法1. 概述LTC2954是一种低功耗电源管理IC，可用于控制和监测电池供电系统。

它具有多种功能，包括电源开关控制、电池电量检测、系统复位和电源故障检测等。

本文将详细介绍LTC2954的各项功能和使用方法。

2. 功能特点LTC2954具有以下主要功能特点：•电源开关控制：LTC2954可以通过外部触发信号或内部定时器控制电源的开关。

它可以根据需求进行自动开关机控制，提高系统的能效和电池寿命。

•电池电量检测：LTC2954可以监测电池的电量，并提供准确的电量信息。

它采用电流积分技术，可以测量电池的充放电情况，并将电量信息反馈给系统，以便进行电池管理和预警。

•系统复位：LTC2954可以监测系统的电压和电流，并在电源异常或故障时进行系统复位。

它可以检测电压过高、过低、断电等情况，并及时发出复位信号，保护系统的稳定运行。

•电源故障检测：LTC2954可以检测电源故障，如过电流、过温等情况，并及时响应。

它可以通过外部触发信号或内部定时器进行故障检测，并发出警报信号，以便及时处理故障情况。

3. 使用方法LTC2954的使用方法如下：3.1 电源开关控制LTC2954可以通过外部触发信号或内部定时器进行电源开关控制。

通过设置相应的控制寄存器，可以实现自动开关机控制。

例如，设置定时器使LTC2954每隔一定时间检测一次电池电量，当电量低于设定值时，自动关闭电源。

3.2 电池电量检测LTC2954采用电流积分技术进行电池电量检测。

通过测量电池的充放电情况，可以准确计算电池的电量。

可以通过读取相应的寄存器来获取电量信息，并进行电池管理和预警。

3.3 系统复位LTC2954可以监测系统的电压和电流，并在电源异常或故障时进行系统复位。

通过设置复位阈值和延迟时间，可以灵活地配置复位功能。

例如，当电压低于设定值并持续一段时间时，LTC2954会发出复位信号，重启系统。

3.4 电源故障检测LTC2954可以检测电源故障，如过电流、过温等情况，并及时响应。

lt电源芯片

lt电源芯片
LT电源芯片（Linear Technology）是一家全球知名的模拟电
源管理技术领导者，自1971年成立以来，一直致力于提供高
精度、高性能和高可靠性的模拟集成电路产品。

LT电源芯片
在多个领域得到广泛应用，包括航空航天、汽车、工业控制、通信系统等。

LT电源芯片以其独特的技术和创新设计而闻名，其产品具有
以下特点：
1. 高精度：LT电源芯片在输出端提供非常高的精度和稳定性，能够满足各种严苛的应用需求。

无论是电压稳定器、电流源还是放大器，LT电源芯片都能提供卓越的性能。

2. 高效率：LT电源芯片采用高效能的转换器架构，能够将输
入电能高效地转换为输出电能，提高系统的整体效率。

这对于依赖电池供电的应用尤为重要，可以延长电池寿命并减少能量消耗。

3. 宽输入电压范围：LT电源芯片具有宽广的输入电压范围，
可以适应不同电源的输入电压变化。

无论是直流输入还是交流输入，LT电源芯片都能提供稳定的输出电压。

4. 可编程性：LT电源芯片提供了多种可编程功能，可以根据
具体需求进行设置和调整。

这使得设计人员可以根据系统的要求来调整电源输出的电压、电流和保护功能等，提高了系统的灵活性和可扩展性。

5. 高可靠性：LT电源芯片以其高品质和可靠性而备受推崇。

LT电源芯片在设计和制造过程中注重细节和严格的质量控制，以确保产品的可靠性和长寿命。

总之，LT电源芯片作为模拟电源管理领域的领导者，不仅在
性能和可靠性方面表现出众，而且在创新和技术上也保持着领先优势。

无论是用于航空航天、汽车、工业控制还是通信系统，LT电源芯片都能提供满足各种严苛需求的解决方案。

5.2.3 电子设备电源管理芯片_绿色电源——电子设备电源管理技术与解决方案_[共5页]

绿色电源—电子设备电源管理技术与解决方案2145.2.3 电子设备电源管理芯片1．LTC3455电源管理芯片凌特公司的LTC3455是带USB电源管理器和锂离子电池充电器的双DC/DC变换器，它把从前需要5个或更多芯片实现的几个功能集成在一起，能执行电源选择、电池充电、两级DC/DC变换和热变换控制。

此外，它还具有低电池电压指示功能。

LTC3455采用24引脚QFN 封装（4mm×4mm）。

LTC3455可选择和处理来自AC适配器、USB电源及电池的电压。

若AC适配器存在，则芯片从适配器获得电源；否则，它从USB接口获得电源。

若没有连接AC适配器和USB电源，则IC从电池获得电源，并将选择的电源电压输出到芯片中的两个DC/DC变换器和热交换控制器中，产生另外的3个电源电压。

图5-31为LTC3455的功能框图。

图5-31 LTC3455电源管理功能框图LTC3455的电源供给不同于普通的电池和电源管理IC中实现的充电器馈入系统，在这种充电器馈入系统中，外部电源不直接供给用电负载，而是用AC适配器或USB接口来给电池充电，然后由电池给负载供电。

若电池处于深度放电状态，则负载获得电源将有一定的延迟时间。

采用LTC3455可消除这种延迟，LTC3455在接入AC适配器或USB电源之后，即可立即给电子设备供电。

另外，芯片将处理任何不被负载使用的可用电源，并利用其对电池充电。

由于LTC3455消除了充电延迟时间，所以延长了电池的有效运行时间。

每当AC适配器或USB电源有效时，就可去除不必要的电源变换级（即电池充电）。

采用这种电源管理技术提高了系统性能和电源使用效率。

开发LTC3455的关键技术问题是电源启动控制。

在过去的电源系统设计中必须用分立的FET、运放和其他元器件实现这种功能，但在热插拔时会引起大的涌入电流，这会导致系统产生问题。

而LTC3455芯片为USB和AC适配器电源输入提供热插拔保护，它可以独立工作，作为可独立应用的电源管理控制器。

ltc4008笔记本电脑充电控制芯片

TOP VIEWTOP VIEWLTC4008是凌特公司生产的多用途恒流、恒压电池充电控制器,它的输入电压为6-28V,输出电压为3-28V,最小压降为0.5V,电压精度为±0.8%,充电电流达4A。

LTC4008的内部电路框图LTC4008LTC4008的针脚封装图THERMISTOR10kNTCQ3引脚号引脚名称引脚功能1DCIN6-28V直流电源输入端2I CL输入限流指示端3ACP/SHDN关断控制输入端。

该脚为高电平时,用于指示电源适配器电压;该脚被拉低时,充电器关闭4R T定时电阻连接端。

5FAULT充电状态输出端。

6GND接地端7V FB电压反馈误差放大器(EA输入端8NTC电池温度检测输入端9I TH电流模式PWM内部环路控制信号,该引脚电压较高时,充电电流也较大。

在该引脚与GND之间连接一个RC串联网络可对电路提供环路补偿10PROG充电电流编程/监视输入/输出。

在该引脚与地之间接一外部电阻,可与电流牎鞍电阻一起对峰值充电电流进行编程。

该引脚电压可线性指示冲顶啊电流。

11CSP电流放大器CA1输入端12BAT电池电压检测输入端13BATMON电池充电电压指示端。

当未检测到AC适配器时,器件内部开关断开。

该引脚到VFB引脚的外部分压电阻器可用于设定充电器的浮充电压14FLAG充电电流指示输出端。

当充电电流减小到最大编程电流的10%时,该引脚输出低电平15CLN限流放大器CL1正输入端16CLP限流放大器CL1负输入端17TGATE充电开关管高端驱动信号输出端18PGND接地端19BGATE充电开关管低端驱动信号输出端20INFET外部输入P沟道MOSFET驱动信号输出端LTC4008的各引脚功能LTC4008的典型应用电路图。

ltc4412原理

ltc4412原理
LTC4412是一种用于电源管理的芯片，它可以提供电池保护和电源选择功能，以确保系统的稳定工作。

这款芯片采用了先进的技术，能够在多种电源供电情况下实现无缝切换，保护电池免受过充、过放和短路等问题的影响。

在使用电池供电时，LTC4412可以监测电池的电压，并在电压低于安全阈值时自动切断电路，以防止电池过放。

同时，当电池电压恢复正常时，它会重新连接电路，以保证系统正常运行。

LTC4412还具有电源选择功能，可以根据外部电源的优先级选择最佳的供电源。

当外部电源可用时，它会自动切换到外部电源，并将电池断开连接，以延长电池的使用寿命。

而当外部电源不可用时，它会立即切换到电池供电，以保持系统的连续工作。

LTC4412的工作原理非常简单，它通过内部的电源开关和电流限制器来实现电池保护和电源选择功能。

当电池电压低于安全阈值时，电源开关会关闭，以切断电路。

而当外部电源可用时，电源选择电路会自动切换到外部电源。

总的来说，LTC4412是一款功能强大且性能稳定的电源管理芯片，它可以保护电池免受损坏，并确保系统在不同供电情况下的正常运行。

无论是在移动设备、工业控制还是其他领域，LTC4412都能发挥重要的作用，提高系统的可靠性和稳定性。

锂电池线性充电管理芯片LTC4065及其应用

锂电池线性充电管理芯片LTC4065及其应用随着移动计算技术和无线通信技术的发展,微型移动终端设备在移动数据采集、传输、处理及个人信息服务等领域得到越来越多的应用。

锂电池因其体积小、能量密度高、无记忆效应、循环寿命高、高电压电池和自放电率低等优点，近年来已经成为微型移动终端设备的首选电源。

锂电池的特性以及应用环境的需求，对微型移动终端设备充电方案的设计提出了更高的要求。

因此在充电方案的设计中需要综合考虑成本、体积、噪声、效率等因素。

[1] LTC4065是一款用于单节锂电池的完整恒定电流/恒定电压线性充电管理芯片，可提供高达750 mA且准确度为5%的可设置的充电电流，并支持直接使用USB端口对单节锂电池进行充电。

同时其热反馈功能可调节充电电流，以便在大功率工作或高环境温度条件下对芯片温度加以限制，确保安全工作。

由于采用了内部MOSFET架构，因此无需使用外部检测电阻器或隔离二极管。

很少的外部元件数目加上其2 mm×2 mm DFN封装，使得LTC4065尤其适合无线PDA、蜂窝电话、无线传感器终端等应用。

功能齐全的LTC4065还包括自动再充电、低电池电量充电调节、软启动等丰富功能。

图1 LTC4065芯片引脚排列1 LTC4065的引脚功能LTC4065采用了热处理能力较强的6引脚小外形封装(DFN)，且实现产品无铅化，底部采用裸露衬垫，直接焊接至PCB以实现电接触和额定散热性能。

引脚排列如图1所示。

各引脚功能如下：引脚1，GND，接地端。

引脚2，CHRG，漏极开路充电状态输出。

充电状态指示引脚具有三种状态：下拉、2 Hz脉动和高阻抗状态。

该输出可以被用作一个逻辑接口或一个LED驱动器。

对电池进行充电时，有一个内部N沟道MOSFET将GHRG引脚拉至低电平。

当充电电流降至全标度电流的10%时，CHRG 引脚被强制为高阻抗状态。

如果电池电压处于2.9 V以下的持续时间达到充电时间的1/4，则认为电池失效，而且CHRG引脚将以2 Hz的频率脉动。

ltc7001用法

ltc7001用法
LTC7001是一种高性能数字电压控制器芯片，广泛用于工业控制和电源管理应用。

它提供了稳定、可靠的电压控制功能，能够满足复杂的电源管理需求。

以下将详细介绍LTC7001的用法。

1. 电压控制功能：LTC7001具有高精度的电压控制功能，可根据需要调整输出
电压。

它采用了先进的数字控制技术，可以实现具有极低温度漂移和噪声的稳定输出。

2. 稳压保护功能：LTC7001还具备稳压保护功能，可保护电路免受过流、过压、过热等问题的影响。

它内置了多种保护机制，并能够自动检测和应对异常情况，确保电路的安全运行。

3. 通信接口：LTC7001支持多种通信接口，如I2C和SPI，可与其他设备进行
数据交互。

通过这些接口，用户可以方便地对LTC7001进行配置和监控。

4. 可编程控制：LTC7001可以通过编程方式来控制输出电压和其他参数。

用户
可以使用相应的软件或开发工具来编写控制代码，实现复杂的电源管理功能。

5. 可靠性和稳定性：LTC7001经过严格的质量控制和稳定性测试，具有出色的
可靠性和稳定性。

它采用高品质的材料和先进的制造工艺，能够在各种环境条件下正常运行。

总结：
LTC7001是一款功能强大的数字电压控制器芯片，广泛应用于电源管理和工业
控制领域。

通过其高精度的电压控制功能、稳压保护功能、灵活的通信接口和可编程控制特性，LTC7001能够满足复杂的电源管理需求，并提供可靠的电路保护和
稳定性。

ltc2949的典型应用电路的变形

ltc2949的典型应用电路的变形
摘要：
1.LTC2949 简介
2.LTC2949 的典型应用电路
3.LTC2949 应用电路的变形
4.变形电路的优点和应用场景
5.总结
正文：
LTC2949 是一款高度集成的电源管理芯片，它具有低噪声、低失真、低漂移、低静态电流等优点，广泛应用于各种电源管理系统中。

LTC2949 的典型应用电路包括输出电压控制、电池充电管理、电源开关控制等。

其中，输出电压控制是LTC2949 最常见的应用之一，通过外接电阻和电容，可以实现输出电压的精确控制。

电池充电管理是LTC2949 的另一个重要应用，它可以实现电池的智能充电，延长电池寿命。

电源开关控制是
LTC2949 的另一个重要应用，通过控制电源开关的开关时间，可以实现电源的智能管理。

然而，在实际应用中，为了满足不同的应用需求，LTC2949 的应用电路常常需要进行变形。

比如，可以通过调整电阻和电容的值，改变LTC2949 的输出电压范围，以适应不同的应用场景。

还可以通过改变LTC2949 的工作模式，实现不同的电源管理功能。

这些变形电路的优点是，可以灵活地满足不同的应用需求，提高电源管理
的效率和稳定性。

同时，变形电路也可以提高系统的可靠性和安全性，避免电源管理芯片的过热和过压等故障。

LTC3589：电源管理解决方案

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的状态报告和一个中断Fra bibliotek出。ＬＣ５９的单电感器、同步降压一Ｔ３８升压型转换器产生
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具系统复位的按钮ＯＮＯＦ／Ｆ控制；
个２５至５的用户可编程输出电压轨。该降压一．ＶＶ升压
世界电子元器件２．０１９ｇｅｃ．ｃｎｃ０ｅｃ．ｏｍ
钮接口控制器、通过Ｉ接口实现的稳压器控制以及大量Ｃ
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ＰＣ卡、始终接通实时时钟（Ｔ以及多种其它功能供电。ＲＣ）ＬＣ３８Ｔ５９含有３个大电流、高效率降压型稳压器、一个大电流，高效率降压－升压型稳压器、４个低噪声低压差线稳压器（Ｄ。对多稳压器提供支持的是一种可配置度很ＬＯ）高的电源排序能力、动态电压调节输出电压控制、一个按

ltc3588工作原理

ltc3588工作原理
LTC3588是一种单片能量收集系统，具有工作在电量非常低
时（如小于100µW）的能力。

它的工作原理如下：
1. 能量收集：LTC3588通过一个功率跟踪电源，从外部能源
源（如太阳能电池、热电转换器等）收集能量。

该芯片可以调整功率跟踪电阻的阻值来匹配输入能源的特性。

2. 储能：收集到的能量经过一个功率选择器，可以选择将能量储存在电容器或者一个锂离子电池中。

能量储存过程中，
LTC3588对能量进行最大功率点跟踪（MPPT），以确保最高
效率地收集和储存能量。

3. 电量管理：LTC3588还包括一个电池管理器，可以对锂离
子电池进行充电和放电控制，以确保电池的安全和长寿命。

该芯片还可以通过内部可调节的电压稳压器提供固定的输出电压。

4. 增强型能量管理：LTC3588还具有一些增强功能，如可配
置的低功耗模式、电荷泵和电荷比较器，以进一步提高能量管理的效率和灵活性。

总的来说，LTC3588通过能量收集、储能、电量管理和增强
能量管理等功能，实现对非常低功率能量的收集、储存和管理，从而为低功耗电子设备提供可靠的能源供应。