(完整word版)跨阻放大器(TIA )

TIA(互阻放大器设计)

224IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 1, JANUARY 2011A 76 dB 1.7 GHz 0.18 m CMOS Tunable TIA Using Broadband Current Pre-Ampliﬁer for High Frequency Lateral MEMS OscillatorsHossein Miri Lavasani, Member, IEEE, Wanling Pan, Member, IEEE, Brandon Harrington, Member, IEEE, Reza Abdolvand, Member, IEEE, and Farrokh Ayazi, Senior Member, IEEEAbstract—This paper reports on the design and characterization of a high-gain tunable transimpedance ampliﬁer (TIA) suitable for gigahertz oscillators that use high- lateral micromechanical resonators with large motional resistance and large shunt parasitic capacitance. The TIA consists of a low-power broadband current pre-ampliﬁer combined with a current-to-voltage conversion stage to boost the input current before delivering it to feedback voltage ampliﬁers. Using this approach, the TIA achieves a constant gain of 76 dB-Ohm up to 1.7 GHz when connected to a 2 pF load at the input and output with an input-referred noise below 10 Hz in the 100 MHz to 1 GHz range. The TIA is fabricated in a 1P6M 0.18 m CMOS process and consumes 7.2 mW. To demonstrate its performance in high frequency lateral micromechanical oscillator applications, the TIA is wirebonded to a 724 MHz high-motional resistance ( unloaded , , pF) and a 1.006 GHz high-parasitic ( unloaded , p , p pF AlN-on-Silicon resonator. The 724 MHz and 1.006 GHz oscillators achieve phase-noise better than dBc Hz and dBc Hz @ 1 kHz offset, respectively, with a ﬂoor around dBc Hz. The 1.006 GHz oscillator achieves the highest reported ﬁgure of merit (FoM) among lateral piezoelectric micromechanical oscillators and meets the phase-noise requirements for most 2G and 3G cellular standards including GSM 900 MHz, GSM 1800 MHz, and HSDPA.QpA2150 8732 ) 94 1542000750 7100Index Terms—Bandwidth enhancement, broadband ampliﬁer, current ampliﬁer, MEMS, micromechanical resonator, oscillator, phase noise, piezoelectric resonator, sustaining ampliﬁer, transimpedance ampliﬁer.I. INTRODUCTIONAFREQUENCY reference oscillator is a pivotal block of any radio transceiver as it signiﬁcantly affects the performance, size and cost of the transceiver [1]. During the past three decades, reference oscillators have been built based on highquartz crystals [2]. Their superior stability and absolute frequency accuracy has allowed them to be the industry’s preferredManuscript received April 23, 2010; revised June 25, 2010; accepted September 13, 2010. Date of publication November 18, 2010; date of current version December 27, 2010. This paper was approved by Guest Editor Koﬁ Makinwa. H. M. Lavasani, W. Pan, and F. Ayazi are with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, GA 30332 USA (e-mail: hosseinm@; ayazi@). B. Harrington and R. Abdolvand are with the Oklahoma State University, Tulsa, OK 74106 USA. Color versions of one or more of the ﬁgures in this paper are available online at . Digital Object Identiﬁer 10.1109/JSSC.2010.2085890choice for frequency synthesis applications. However, emerging wireless applications have created a number of challenges that can not be addressed by traditional quartz crystal reference oscillators. In multi-gigahertz transceivers [3], the large up-conversion ratio of synthesizers limits the performance of the RF front-end. In addition, higher data-rate transceivers that rely on OFDM-based 3G/4G standards such as HSDPA with stringent error vector magnitude (EVM) speciﬁcations demand very low close-to-carrier [4], [5] and far-from-carrier [6] phase-noise; for low frequency crystals, this is only accommodated by a PLL whose loop ﬁlter’s 3-dB bandwidth (BW) is shrunk to the point that the ﬁlter’s integration on the chip becomes challenging. Another shortcoming is the inherent incompatibility of quartz crystals’ fabrication process with standard CMOS. UHF silicon micromechanical oscillators, on the other hand, offer smaller form factor, higher frequency, and potential integration with ICs while delivering near-crystal phase-noise performance [7]–[9]. Various capacitively-transduced [10], [11] and piezoelectrically-transduced [12] micromechanical resonator technologies have been explored over the past decade to assist in the development of high frequency micromechanical reference oscillators. Unlike high frequency capacitive resonators that suffer k ) from high loss, usually modeled by a large resistance ( called motional resistance [10], and a large DC polarization voltage ( ) [11], piezoelectric resonators exhibit substantially lower motional resistance at comparable frequencies [12], [13]. Recent advances in high-frequency lateral-mode piezoelectric silicon micromechanical resonators have increased their product to a level comparable to quartz crystal [13] but their motional resistance is still signiﬁcantly higher than the thin-ﬁlm bulk acoustic resonator (FBAR). The motional resistance can be reduced at the expense of larger transduction area which increases the parasitic capacitance. This high motional resistance combined with the large parasitic capacitance of the resonator complicates the design of low-power oscillators. Since the most effective way to build oscillators based on lateral-mode micromechanical resonators with high motional resistance is to use a high-gain transimpedance ampliﬁer (TIA) that is placed in the positive feedback loop with the resonators [7]–[9], [14], the development of low-power high-gain sustaining ampliﬁers becomes necessary. Although open-loop architectures have been occasionally used for low phase-noise MEMS oscillators [14], the most0018-9200/$26.00 © 2010 IEEELAVASANI et al.: A 76 dB1.7 GHZ 0.18 m CMOS TUNABLE TIA USING BROADBAND CURRENT PRE-AMPLIFIER225Fig. 1. (a) Block diagram of the LBAR resonator showing the electrode placement for high-order mode excitation. (b) High-order mode shape of the LBAR. (c) Equivalent lumped electrical model of the LBAR.popular TIA topology used in MEMS oscillators remains the shunt-shunt feedback TIA as it improves the BW and reduces the input-referred noise [7], [15]. However, the BW improvement is only made possible by lowering the feedback resistor at the expense of gain [8]. Further increase in the gain requires higher power consumption. A number of advanced BW and gain enhancement techniques have been proposed to improve the performance of broadband TIAs [16]–[18]. A large body of literature has focused around the boosted- regulated cascode (RGC) topology as a reliable approach to lower the input impedance of the TIA and hence increase its BW [16], [19]. Other researchers have explored cascade of ampliﬁers with shunt-shunt local feedback to lessen the impact of inter-stage poles on the frequency response [8], [20]. These techniques, however, are not suitable for low-power low phase-noise micromechanical oscillators; the excessive noise generated by RGC’s input stage deteriorates the phase-noise. In addition, acceptable gain can only be achieved by increasing the number of stages with obvious implication for the power consumption. Using a broadband current ampliﬁer increases the gain while maintaining the BW with little or no extra power. The remainder of this paper is organized as follows. Section II provides an insight into the design and characterization of lateral bulk acoustic resonators (LBAR) that are used throughout this work. The broadband current ampliﬁer topology including the low-power technique used for BW improvement in the ampliﬁer is laid out in Section III. Section IV presents the concept of gain improvement by using the broadband current pre-ampliﬁer introduced in the previous section. In addition, it explains the TIA design methodology and provides theoretical noise analysis. Section IV is followed by experimental results that are partitioned into TIA characterization and interface results with the resonator. Comparisons with state-of-the-art TIAs and high frequency micromechanical oscillators are also listed in this section. Finally, the paper concludes with a brief overview of the TIA performance and its applications in micromechanical oscillators. II. HIGH FREQUENCY LATERAL BULK ACOUSTIC RESONATOR Although still inferior to FBAR in terms of motional resistance, laterally-excited thin-ﬁlm piezoelectric-on-silicon micromechanical resonators have come a long way from lowprototypes in MHz range [21] to state-of-the-art gigahertz batch-fabricated resonators with Q’s in excess of 7,000 [13], 2-3 larger than FBARs that are widely used as ﬁlters in thereceiver front-ends. This section provides a brief overview of the resonator design and characterization. A. Resonator Design The resonators used in this work are from the family of highorder laterally-excited bulk acoustic resonators (LBAR). A simpliﬁed schematic of a third-order thin-ﬁlm piezoelectric-on-silicon LBAR is shown in Fig. 1(a). In these devices the top metal pattern resembles the alternating stress pattern induced in the structure at the targeted high-order resonance mode (Fig. 1(b)). Therefore, the resonance frequency is determined by the distance between two neighboring top electrode ﬁngers (i.e., ﬁnger pitch) [22]. The resonator is a two-terminal device, which consists of an energy-transducing thin-ﬁlm piezoelectric AlN layer sandwiched in between two metal electrodes and stacked on top of a silicon resonant body. The top metal layer is shaped into inter-digitated sense and drive electrodes. The bottom metal is usually considered as the reference and is connected to the global ground. Superior acoustic properties of the silicon layer (e.g., low acoustic loss and high energy density) provide for imand power handling of these resonators [15], [22], proved [23]. The resonance frequency can be approximated by (1) is the ﬁnger pitch, and and are the effective where Young’s Modulus and density of the composite structure, respectively. In order to improve the motional resistance at higher frequencies the mode order can be increased [22], [24]. Utilization of high-order modes also provides for the added advantage of less sensitivity to lithographic error. On the other hand, the multi-ﬁnger pattern designed on a large structure will excite numerous unwanted near-resonance spurious modes some of which are strong in magnitude relative to the targeted resonance mode. The spurious modes’ proximity promotes the energy transfer from the targeted mode of operation to the neighboring unwanted resonance modes, inevitably broadening the resonance peak and diminishing the . The solution is to use additional anchors at displacement nodes along the frequency setting dimension to add constraints to the vibration pattern to better suppress the spurious modes and hence, improve the [13]. B. Resonator Electrical Model An LBAR resonator operating in a higher-order mode can be modeled as a series-RLC with shunt and feedthrough parasitic226IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 1, JANUARY 2011Fig. 2. Frequency response and SEM view of the 724 MHz AlN-on-Si resonator. Fitted RLC model parameters are: C 146:6 aF.R 750 , L 329 7 H and: Fig. 3. Frequency response and SEM view of the 1.006 GHz resonator. Fitted RLC model parameters are: R 150 , L 168 5 H and C 148 6 aF.: :capacitance (Fig. 1(c)) [8]. The shunt parasitic capacitance, , pF) and is caused by the overlap of top and bottom is large ( electrodes in addition to pad parasitics. The feedthrough capac, represents the path between the input and output itance, node and is typically very small ( fF) for laterally-excited micromechanical resonators. The components appearing in the , , and are the main factors in deterseries path, mining the resonance frequency and are referred to as motional capacitance, motional inductance, and motional resistance, respectively. For a homogenous single degree of freedom (SDOF) mechanical system such as a LBAR, these parameters are given by [22](2) where is the electromechanical coupling coefﬁcient, and and are the effective stiffness and mass of the resonator, respectively. Despite the availability of closed form solutions for the aforementioned parameters, the resonator is generally considered as an empirical value and it would be prudent to use the measured frequency of the resonator to accurately determine the ﬁtted model parameters.LAVASANI et al.: A 76 dB1.7 GHZ 0.18 m CMOS TUNABLE TIA USING BROADBAND CURRENT PRE-AMPLIFIER227Fig. 4. (a) Simple CMOS current ampliﬁer; (b) Current ampliﬁer with shunt feedback; (c) Simpliﬁed equivalent circuit for the current ampliﬁer for shunt feedback showing the noise sources.C. Resonator Characterization Two AlN-on-Silicon resonators, one 724 MHz high-motional resistance device (Fig. 2) used to evaluate the TIA performance in the high gain setting, and another 1.006 GHz high-parasitic device (Fig. 3) used to benchmark the TIA performance in the large BW setting, are fabricated and characterized. The resonators are fabricated on silicon-on-insulator (SOI) wafers with 5 m-thick device layer. Both resonators were individually characterized before they are interfaced with the TIA. The frequency response of each resonator is obtained by measuring the calibrated 2-port S-parameters with an Agilent E8364B vector network analyzer (VNA). The ﬁtted model parameters are then extracted from the resonator response. The shunt parasitic capacitance can be approximated by calculating the capacitance from overlap area between top and bottom electrodes. This is usually an underestimate as the fringing ﬁelds, due to the sharp edges in the ﬁngers, increase the capacitance (by as much as 50%). As such, the approximate value of for 724 MHz and 1.006 GHz resonators are 2 pF and 3.2 pF, respectively. Moreover the measured data reveals an unloaded 2, 000 and extracted 750 for the 7 and extracted 724 MHz resonator and 150 for the 1.006 GHz resonator.III. BROADBAND CURRENT AMPLIFIER Despite the availability of the input signal in the form of current, current ampliﬁcation has received little attention as a gain boosting method in traditionally gain-starved low-power CMOS TIAs. This is due to the relatively-high power consumption of most broadband CMOS current ampliﬁers. In many topologies, a portion of the static (DC) current passing through the input stage is ampliﬁed along with the AC content. This complicates the design of low-power high-gain current ampliﬁers. Therefore, a low-power BW enhancement technique becomes necessary. A. Frequency Response and BW Enhancement Most broadband CMOS current ampliﬁers rely on inherently-broadband diode-connected topologies such as the current mirror to maintain high BW [25] (Fig. 4(a)). In addition, the current mirror topology brings an added advantage to the TIA as it can be easily combined with the current-to-voltage conversion stage; the output current can be passed through a resistive load of appropriate size to convert it to voltage. While the input resistance of the current mirror, which is proof the input transistor, is acceptable for most portional to low-frequency applications, even in the presence of large shunt228IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 1, JANUARY 2011parasitic capacitance, the BW requirement imposed by gigahertz resonators puts a strain on the power consumption. To reduce the power consumption, shunt feedback is applied to the input stage resulting in a modiﬁed version of the inherently-low input resistance current-mirror topology (Fig. 4(b)). The equivalent circuit of Fig. 4(c) helps in determining the frequency response of the current ampliﬁer. The body effect is nearly cancelled by shorting the bulk terminal of every MOS device to its source. Neglecting all parasitic capacitance of the , MOS transistors except the larger gate-source capacitance, the transfer function of the current ampliﬁer can be approximated by:(3) where is the drain-source resistance of are transconductances of , , and and are deﬁned as: and , , , respectively.Fig. 5. Comparison of the AC response of the current ampliﬁer with shunt feedback with a simple current ampliﬁer w/o shunt feedback.(4) where , , and are the drain-source resistances of , , and , and , , and are the gate-source , , and , respectively. represents capacitances of the input capacitive load to the current ampliﬁer. From (3), it is clear that the current ampliﬁer is a multi-pole system with the dominant pole associated with the input terminal. In the absence of channel length modulation effects, the midband current gain can be derived by simplifying (3): (5) where is the ratio of to . The current gain is very close to the gain of a current mirror without the feedback. This shows that the shunt feedback does not degrade the gain. Similarly, the input impedance of the current ampliﬁer with shunt feedback is given by:drain-source resistance of the MOS transistor, a pMOS transistor with relatively-low power consumption that yields a small transconductance can substantially lower the input impedance of the current ampliﬁer. Fig. 5 compares the simulated AC current gain and input impedance of the two current ampliﬁers shown in Fig. 4 with similar power consumption; in the current ampliﬁer with shunt feedback, a small portion of the input stage current ﬂows into the feedback path to improve the BW. The BW is extended by as much as 70% with the same input capacitive load while the gain has remained roughly the same (the gain difference is around 0.5 dB); moreover, the in-band ( dB ) input impedance is reduced by more than 2 to less than 50 . Less than expected improvement in the BW can be explained by . the additional parasitic contribution from B. Noise Analysis For large , (3) states that the dominant pole appears at the coming from input of the TIA. In addition, due to the large MHz), the resonator, at sufﬁciently high frequencies ( the loading due to impedance can . With this information, be approximated by the noise analysis can be simpliﬁed by neglecting the effect of and . The simpliﬁed model, (Fig. 4(c)) shows that the input-referred current noise of the TIA can be found in three and are directly added to steps; the noise currents of and its load resistor, , are the input. The noise current of readily available at the output node. They can be referred back to the input when divided by the current gain of the ampliﬁer. Therefore, the problem is reduced to ﬁnding the input-referred and . Considering contribution of the noise currents of Fig. 4(c) and eliminating all noise sources except those of and , the output noise can be approximated by (8), shown and are the at the bottom of the next page, where and and can be approximated drain noise currents of the by(6) , the midband input resistance Neglecting the effect of small where the effect of phase shift due to the parasitic and load capacitances is negligible can be approximated by: (7) where , , and are drain-source conductance of , , and , respectively. Expression (7) shows that the input as resistance is reduced by a factor comparable to the gain of promised by shunt feedback theory. Due to the inherently-large(9)LAVASANI et al.: A 76 dB1.7 GHZ 0.18 m CMOS TUNABLE TIA USING BROADBAND CURRENT PRE-AMPLIFIER229Fig. 6. Schematic diagram of the TIA with current pre-ampliﬁcation stage (Biasing is not shown).where is the noise coefﬁcient and is equal to 2/3 in longchannel regime. Short-channel effects increase by a factor of 2 to 4 [26], [27]. Dividing (8) by the current gain gives the and : input-referred current noise contribution due to(10) Using superposition, it is possible to ﬁnd the noise contribution from other transistors: (11) where and are the input-referred curand , and and , rent noise contributions from respectively. These current noise contributions can be approximated byFrom (11), it is evident that the major contributing factors to and the input-referred noise are the noise current from since their noise current is directly added to the input noise. The combination of these two noise sources, alone, is more than 40% Hz). However, of the theoretical calculated value ( 5 further analysis of (8) determines that the second most imporrather than tant contributing noise source is the noise from . This is due to the larger transconductance, , and extra experiences before ampliﬁcation that the noise current from reaching the output. This provides the designer with a ﬂexibility to minimize the power consumption without signiﬁcant impact can on the input-referred noise. The noise contribution from . Increasing the length of helps be lowered by increasing reduce the effect of channel-length modulation that negatively impacts its drain-source resistance. IV. TIA WITH CURRENT PRE-AMPLIFIER The broadband current ampliﬁer designed in Section III can be used to realize a low-power high-gain TIA. Fig. 6 shows the TIA block diagram and schematic. The TIA consists of 3 sections: a current pre-ampliﬁer that is combined with a current-to-voltage conversion stage, voltage ampliﬁers and a 50 buffer.(12)(8)230IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 1, JANUARY 2011The current ampliﬁer is used as the ﬁrst stage of the TIA to boost the input current signal before converting it to the voltage ratio of to domain. The current gain is set by the ( ). To save power, the current-to-voltage conversion stage is combined with the output stage of the current ampliﬁer. The is mirrored into current ﬂowing through the drain of and passed through a properly-sized resistive load to produce a voltage. Using small resistors ensures minimal effect on BW. mW. The transThe power consumption of the ﬁrst stage is impedance gain in the ﬁrst stage can be written as:, respectively. Replacing the RC “ ” network with a pure , is reduced to: resistive feedback,(17) For small : and large , can be further simpliﬁed to(13) Since the midband current gain is approximated by “ ”, the transimpedance gain is equal to:(18) The output buffer is a broadband 50 -matched CS stage to maximize the power transfer to the 50 input of the measurement equipment. The power matching, however, comes at the expense of larger noise as the current noise of the resistor is inversely proportional to its resistance value. Fortunately, since the drain-source resistance of the nMOS appears in parallel with the load resistance, the 50 constraint can be relaxed. Proper sizing of the nMOS transistor can accommodate a sufﬁciently large resistive load for optimum gain and BW performance. V. EXPERIMENTAL RESULTS Since the TIA is intended for use in micromechanical oscillators, both the standalone and interface test results are necessary. Therefore, in addition to the traditional performance metrics such as the gain, noise and linearity, the TIA should be interfaced with high frequency micromechanical resonators that are representative of those in the intended frequency range. Therefore, this section is partitioned into two parts. A. TIA Standalone Measurement Results(14) The voltage ampliﬁer section consists of two common source (CS) stages that form a modiﬁed Cherry-Hooper with gain tuning in the second stage [28]. The choice of CS over other higher gain topologies such as cascode is to achieve higher output swing, which helps improve the TIA linearity. Shunt-shunt feedback reduces the input and output impedance of the last stage, which in turn, helps increase the BW by pushing the poles to higher frequency. Using a tunable pMOS resistor in the feedback network, improves the linearity, lowers the noise compared to a nMOS resistor, and makes the TIA capable of interfacing with a variety of resonators with different motional resistance. The shunt-shunt resistive feedback is replaced with an RC “ ” network to create a zero whose frequency can be tuned for pole cancellation: (15) where is the equivalent resistance of . Complete pole cancellation without disturbing the gain of the last stage requires the capacitor and at least one of the resistors are made variable; however, since the ﬁrst stage delivers high gain, the focus in the last stage is tilted toward preserving the BW. As such, the widely tunable “ ” network with a pair of variable resistor and capacitor is traded off with a less complex architecture that only uses a variable resistor. Neglecting all capacitances except , the output resistance, , can be approximated by (16) at the bottom of the page, where , , , and are the drain-source resistances , , , and , and is the transconductance of ofThe TIA is fabricated in a 0.18 m 1P6M CMOS process and consumes around 4.8 mA from 1.5 V supply. The fabricated chip measures 650 m 500 m (Fig. 7). To obtain the frequency response of the TIA, two sets of S-parameters, one for maximum and the other for minimum gain, were measured using an RF probe station connected to an Agilent E8564B VNA. The S-parameters are then imported into ADS and interfaced with an ideal AC current source with 2 pF capacitive load that represents the shunt parasitic capacitance of the resonator (Fig. 8(a)). Gain larger than 76 dB with GHz is achieved (Fig. 8(b)). The phase-shift of the TIA is also shown (Fig. 8(c)). The result is in good agreement with the simulation. The gain can be tuned down to 64 dB with BW greater than 2.1 GHz. Replacing the 2 pF load with a(16)LAVASANI et al.: A 76 dB1.7 GHZ 0.18 m CMOS TUNABLE TIA USING BROADBAND CURRENT PRE-AMPLIFIER231Fig. 7. The micrograph of the die fabricated in 0.18 m 1P6M CMOS process and interfaced with the 724 MHz AlN-on-Si resonator. The resonator dimensions are less than 500 m 300 m (including the pads). The absence of on-chip inductors has reduced the IC area to 650 m 500 m.22photo-diode ( 500 fF), the BW can be extended to 2.5 GHz with gain around 76 dB . To measure the input-referred noise of the TIA, the input is connected to ground through a 2 pF capacitor that emulates the shunt parasitic loading of the resonator. The gain is set to maximum and the output signal is recorded from 10 MHz to 5 GHz with an Agilent E4407C spectrum analyzer. Due to the very small BW of the resonators, spot noise at a particular frequency is a more relevant performance metric than the average noise (Fig. 9) [29]. Optimization of input transistor W/L along and current sources with the reduction in the noise of Hz in have yielded a measured input-referred noise 10 the 100 MHz to 1 GHz which is reasonably close to the simulated performance. The excessive noise at higher frequency ( GHz) is due to the gate-induced current noise of MOSFET whose power spectral density (PSD) increases with frequency and is not accurately estimated by BSIM3 model used here. Due to the large power handling of AlN-on-Si resonators [15], the oscillation power is primarily set by the nonlinearity of the TIA. As such, the dynamic range and gain compression of the TIA have direct impact on the oscillator performance. The use of relatively wide-swing CS with feedback in the third stage has pushed the maximum swing, which directly inﬂuences theFig. 8. (a) Simpliﬁed schematic of the setup used to ﬁnd TIA transfer function, (b) transimpedance gain, (c) phase response of the TIA with 2 pF capacitive load.overload current, beyond 0.62 . Considering the input-reHz in the 100 MHz to 1 GHz, the ferred noise of 10 TIA achieves a remarkably high dynamic range of 131 dB in this range. To obtain the 1-dB compression point ( dB ), the S-parameters in the maximum gain setting are measured when to . The the input power is varied from input-referred (after 10 dB adjustment due to the reﬂecdB . The BW is only reduced by 8% to 1.6 GHz tion) is (Fig. 10). The linearity is improved due to the use of a low-gain。

TI跨阻型放大器应用指南

Application ReportZHCA479 – September 2012 1跨阻型放大器应用指南毛华平德州仪器公司 (TI) 高速应用工程师摘要本文简要介绍了Decompensate 型跨阻型放大器的应用常见问题.Abstract: this article simply introduce the normal application of unity gain stable TIA anddecompensated TIA, and the normal issue met in the real application.Key words: GBP (Gain bandwidth product), decompensated, stability, noise,CF(feedback capacitor), overshoot.Contents1 引言 (2)2 TIA 应用概论 (2)3 TIA 应用指标分析 (3)3.1 带宽计算 33.2 噪声计算 64 实际应用中的常见问题 (6)4.1 振荡 64.2 overshoot 95 总结 (11)6. 参考资料 (11)FiguresFigure 1 decomp 和单位增益稳定运放波特图 (2)Figure2 TIA 光电检测电路 (3)Figure3 TIA 用于DAC 输出电流检测电路 (3)Figure4 未补偿时的波特图 (4)Figure5 补偿后的波特图 (5)Figure6 常用TIA 增益与带宽关系 (6)Figure7 未加补偿的20k 放大电路 (7)Figure8 原始输出响应 (7)Figure9 加补偿后的电路 (8)Figure10 加补偿后的脉冲响应 (8)Figure11 原始补偿的频响 (9)Figure12 增大补偿的电路 (10)Figure13 增大补偿后的脉冲响应 (10)Figure14 增大补偿后的频响 (11)ZHCA4792 跨阻型放大器应用指南1 引言 TIA 全称为trans-impedance amplifier. 也就是跨阻型放大器.在需要电流转电压的应用场合, 如检测微弱光电流信号的场合, 通常需要用到跨阻型放大器. TI 有一系列的跨阻放大器,如OPA656,OPA657,OPA843,OPA84,LMH6629等等. Ti 该产品系列主要的优势在于低噪声, 能支持反馈高增益下宽带应用. 这些特点在微弱光检测的场合是非常关键的. 另外TI 的产品是一系列的, 在不同的指标要求如带宽升级时可以很方便地找到pin-pin 兼容的产品.本文介绍了高速TIA 应用中关注的指标及计算过程. 另外介绍了在光检测应用下常见问题的解决.2 TIA 应用概论在TIA 应用时, 由于输入信号是电流, 能够应用于这种场合的跨阻放大通常需要具备较低的电流噪声和电压噪声. 比较典型的两个器件是:OPA657(1.6GHz,输入电流噪声1.8 fA/rtHz, 输入电压噪声4.8nV/rtHz), OPA847(3.9GHz, 输入电流噪声2.5pA/rtHz, 输入电压噪声0.85nV/rtHz). 这两款都是Decompensated 放大器.Decompensated 放大器特点如下:Decompensated 放大器指的是非单位增益稳定的放大器, 如OPA657最小稳定增益是7V/V , OPA847则为12V/V.其波特图和普通放大器比较如下:Figure 1 decomp 和单位增益稳定运放波特图和单位稳定放大器相比, 其特点如下:带宽更宽, 尤其是小信号下的带宽更宽, Slew rate 更快, 以及更大的GBW. 另外一般来讲, decompensated 的放大器能够提供更好的电压噪声.所以在大增益的跨阻放大且要求一定带宽的场合, 使用decompensated 放大器要比单位增益稳定放大器有优势.G AZHCA479跨阻型放大器应用指南 33 TIA 应用指标分析3.1 带宽计算一个用于光电流检测的常规的跨阻型运放的工作电路一般简化如下:Figure2 TIA 光电检测电路或是用于作DAC 的电流转电压的应用场合:Figure3 TIA 用于DAC 输出电流检测电路对一定的运放, 其GBP 是固定的, Cdiff(芯片输入的寄生差分容值), Ccm(芯片输入的寄生共模容值)也是固定的, 选定前面的光检测管APD 或PIN 后,其寄生容值CD 也就是固定了, 当放大倍数RF 固定的时候, 其能达到的-3dB 闭环带宽大约为:ZHCA4794 跨阻型放大器应用指南公式1但是由于前端的寄生电容Cs 和Rf 会在噪声增益曲线上形成一个零点,导致运放的开环增益曲线和噪声增益曲线相交处的逼近速度为-40dB/dec, 这样就会造成运放的不稳定,也就是会引起自激. 其波特图如下:Figure4 未补偿时的波特图所以要达到这样一个稳定工作有一个前提, 需要采用CF 来作补偿, 在该曲线中引入一个极点. 补偿后的曲线如下:ZHCA479跨阻型放大器应用指南 5Figure5 补偿后的波特图所以需要让运放稳定工作, 且达到最宽的2阶butterworth 频响, 其CF 的取值如下:公式2对于decompensated 的运放, 由于其最小增益的要求, 还引来另外一个要求, 就是其增益要大于其最小稳定增益, 由于在高频下, 其增益表达式如下:所以对特定的decompensated 的运放, 这个值要大于其最小增益要求.公式3在一个假定前端的寄生容性为10pF 的场合, 以下是几个运放能达到的带宽和增益的对应关系:ZHCA4796 跨阻型放大器应用指南Figure6 常用TIA 增益与带宽关系3.2 噪声计算在由以上公式算出的带宽后, 运放本身带来的噪声贡献可以由如下公式算出:公式4其中:I EQ = 等效的输入噪声电流, 这个值在带宽 F < 1/(2πR F C F )内有效.I N = 运放本身输入的电流噪声,算inverting 的输入.E N = 运放输入的电压噪声.C D = 前面的光电二极管的寄生电容.F = 带宽,单位为Hz.4kT = 1.6E – 21J at T = 290°K 根据这个公式计算出等效的输入噪声电流后, 就可以算出在TIA 输出后SNR 了.4 实际应用中的常见问题这里整理几个TIA 运放在实际使用中经常遇到的问题:4.1 振荡这个问题在高增益,又有宽带要求的情况下比较常见.比如设计一个20K 增益的放大链路, 假设总的输入的寄生电容很大, 10pF. 根据上面的图可以看出, 采用GBW 最宽的OPA847进行设计, 最宽稳定带宽只能在50M 附近.设计电路如下:ZHCA479跨阻型放大器应用指南7Figure7 未加补偿的20k 放大电路输入20n 的脉宽信号, 10u 的幅度, 得到的波形如下:Figure8 原始输出响应输出有振荡产生.根据公式算出CF 的取值应该为0.24p. 加上后,电路如下:ZHCA4798 跨阻型放大器应用指南Figure9 加补偿后的电路仿真得到: 可以看到, 振荡消失,只剩过冲. 放大倍数也趋向正常.Figure10 加补偿后的脉冲响应在高增益的场合, 有可能反馈电阻自带的电容以及反馈走线带来的寄生电容都可以达到这么微小的电容值. 所以需要依具体的测试结果来确定反馈是否要另外加电容.ZHCA479跨阻型放大器应用指南94.2 overshoot在光时域反射检测光纤状态的场合, 输出上的overshoot 可能会对测量结果产生很大影响,这就需要尽可能地减小TIA 输出的overshoot. 如上图所示的结果, 约有10%的overshoot, 这对实际使用是不利的,需要消除.消除这种过冲最有效的方法是加大反馈电容, 但是这样带来的一个直接后果是带宽减小. 如上面的案例, 在输出有overshoot 的情况下, 原始频响为: -3dB 带宽有40M 左右.Figure11 原始补偿的频响增大反馈到0.45p 时, 过冲消失.ZHCA47910 跨阻型放大器应用指南Figure12 增大补偿的电路Figure13 增大补偿后的脉冲响应但是也可以看到, 20nS的脉冲情况下, 其输出有点被滤除, 增益减小了. 原因就在于输出的带宽变窄, 只剩21MZHCA479跨阻型放大器应用指南 11Figure14 增大补偿后的频响5 总结TIA 运放在作电流放大使用时需要注意带宽和增益的折中, 以及平衡性和带宽的折中. 而同时又得兼顾噪声的贡献, 所以需要综合考虑以上的各项指标.6. 参考资料1. Xavier Ramus “Transimpedance Considerations for High-Speed Amplifiers”2. OPA847,OPA657指标书重要声明德州仪器(TI)及其下属子公司有权在不事先通知的情况下,随时对所提供的产品和服务进行更正、修改、增强、改进或其它更改，并有权随时中止提供任何产品和服务。

tia跨阻放大电路

tia跨阻放大电路摘要：1.TIA 跨阻放大电路的概述2.TIA 跨阻放大电路的工作原理3.TIA 跨阻放大电路的应用领域4.TIA 跨阻放大电路的优点与局限性正文：1.TIA 跨阻放大电路的概述TIA 跨阻放大电路，全称为Transimpedance Amplifier，即跨阻抗放大器，是一种常用于信号处理领域的模拟电路。

它具有很高的输入阻抗和很低的输出阻抗，能够将信号源的电流放大并转换为电压信号，从而实现信号的传输和处理。

2.TIA 跨阻放大电路的工作原理TIA 跨阻放大电路的工作原理主要基于运算放大器的运算原理。

运算放大器具有开环增益无穷大、输入阻抗无穷大、输出阻抗零的特点。

在TIA 电路中，运算放大器的非反相输入端接信号源，反相输入端通过电阻网络形成负反馈，使得电路的输出阻抗接近于零。

这样，信号源的电流被放大并转换为电压信号，从而实现信号的传输和处理。

3.TIA 跨阻放大电路的应用领域TIA 跨阻放大电路广泛应用于各种信号处理系统中，如通信系统、仪器仪表、生物医学工程等。

例如，在光纤通信系统中，TIA 跨阻放大电路用于将光信号转换为电信号；在生物医学工程中，TIA 跨阻放大电路可用于生物传感器的信号处理，实现对生物信号的检测和分析。

4.TIA 跨阻放大电路的优点与局限性TIA 跨阻放大电路具有以下优点：（1）高输入阻抗：TIA 电路的输入阻抗很高，可以最大程度地保留信号源的特性，避免信号衰减或失真。

（2）低输出阻抗：TIA 电路的输出阻抗接近于零，能够驱动较大的负载电阻，实现信号的有效传输。

（3）宽频带：TIA 电路具有较宽的频带，能够处理不同频率范围的信号。

然而，TIA 跨阻放大电路也存在一定的局限性：（1）对运算放大器的依赖性：TIA 电路的工作原理基于运算放大器，因此，运算放大器的性能直接影响TIA 电路的性能。

（2）稳定性：TIA 电路的稳定性受负反馈网络的影响，如果设计不当，可能导致电路不稳定。

TI跨阻型放大器应用指南

Application ReportZHCA479 – September 2012 1跨阻型放大器应用指南毛华平德州仪器公司 (TI) 高速应用工程师摘要本文简要介绍了Decompensate 型跨阻型放大器的应用常见问题.Abstract: this article simply introduce the normal application of unity gain stable TIA anddecompensated TIA, and the normal issue met in the real application.Key words: GBP (Gain bandwidth product), decompensated, stability, noise,CF(feedback capacitor), overshoot.Contents1 引言 (2)2 TIA 应用概论 (2)3 TIA 应用指标分析 (3)3.1 带宽计算 33.2 噪声计算 64 实际应用中的常见问题 (6)4.1 振荡 64.2 overshoot 95 总结 (11)6. 参考资料 (11)FiguresFigure 1 decomp 和单位增益稳定运放波特图 (2)Figure2 TIA 光电检测电路 (3)Figure3 TIA 用于DAC 输出电流检测电路 (3)Figure4 未补偿时的波特图 (4)Figure5 补偿后的波特图 (5)Figure6 常用TIA 增益与带宽关系 (6)Figure7 未加补偿的20k 放大电路 (7)Figure8 原始输出响应 (7)Figure9 加补偿后的电路 (8)Figure10 加补偿后的脉冲响应 (8)Figure11 原始补偿的频响 (9)Figure12 增大补偿的电路 (10)Figure13 增大补偿后的脉冲响应 (10)Figure14 增大补偿后的频响 (11)ZHCA4792 跨阻型放大器应用指南1 引言 TIA 全称为trans-impedance amplifier. 也就是跨阻型放大器.在需要电流转电压的应用场合, 如检测微弱光电流信号的场合, 通常需要用到跨阻型放大器. TI 有一系列的跨阻放大器,如OPA656,OPA657,OPA843,OPA84,LMH6629等等. Ti 该产品系列主要的优势在于低噪声, 能支持反馈高增益下宽带应用. 这些特点在微弱光检测的场合是非常关键的. 另外TI 的产品是一系列的, 在不同的指标要求如带宽升级时可以很方便地找到pin-pin 兼容的产品.本文介绍了高速TIA 应用中关注的指标及计算过程. 另外介绍了在光检测应用下常见问题的解决.2 TIA 应用概论在TIA 应用时, 由于输入信号是电流, 能够应用于这种场合的跨阻放大通常需要具备较低的电流噪声和电压噪声. 比较典型的两个器件是:OPA657(1.6GHz,输入电流噪声1.8 fA/rtHz, 输入电压噪声4.8nV/rtHz), OPA847(3.9GHz, 输入电流噪声2.5pA/rtHz, 输入电压噪声0.85nV/rtHz). 这两款都是Decompensated 放大器.Decompensated 放大器特点如下:Decompensated 放大器指的是非单位增益稳定的放大器, 如OPA657最小稳定增益是7V/V , OPA847则为12V/V.其波特图和普通放大器比较如下:Figure 1 decomp 和单位增益稳定运放波特图和单位稳定放大器相比, 其特点如下:带宽更宽, 尤其是小信号下的带宽更宽, Slew rate 更快, 以及更大的GBW. 另外一般来讲, decompensated 的放大器能够提供更好的电压噪声.所以在大增益的跨阻放大且要求一定带宽的场合, 使用decompensated 放大器要比单位增益稳定放大器有优势.G AZHCA479跨阻型放大器应用指南 33 TIA 应用指标分析3.1 带宽计算一个用于光电流检测的常规的跨阻型运放的工作电路一般简化如下:Figure2 TIA 光电检测电路或是用于作DAC 的电流转电压的应用场合:Figure3 TIA 用于DAC 输出电流检测电路对一定的运放, 其GBP 是固定的, Cdiff(芯片输入的寄生差分容值), Ccm(芯片输入的寄生共模容值)也是固定的, 选定前面的光检测管APD 或PIN 后,其寄生容值CD 也就是固定了, 当放大倍数RF 固定的时候, 其能达到的-3dB 闭环带宽大约为:ZHCA4794 跨阻型放大器应用指南公式1但是由于前端的寄生电容Cs 和Rf 会在噪声增益曲线上形成一个零点,导致运放的开环增益曲线和噪声增益曲线相交处的逼近速度为-40dB/dec, 这样就会造成运放的不稳定,也就是会引起自激. 其波特图如下:Figure4 未补偿时的波特图所以要达到这样一个稳定工作有一个前提, 需要采用CF 来作补偿, 在该曲线中引入一个极点. 补偿后的曲线如下:ZHCA479跨阻型放大器应用指南 5Figure5 补偿后的波特图所以需要让运放稳定工作, 且达到最宽的2阶butterworth 频响, 其CF 的取值如下:公式2对于decompensated 的运放, 由于其最小增益的要求, 还引来另外一个要求, 就是其增益要大于其最小稳定增益, 由于在高频下, 其增益表达式如下:所以对特定的decompensated 的运放, 这个值要大于其最小增益要求.公式3在一个假定前端的寄生容性为10pF 的场合, 以下是几个运放能达到的带宽和增益的对应关系:ZHCA4796 跨阻型放大器应用指南Figure6 常用TIA 增益与带宽关系3.2 噪声计算在由以上公式算出的带宽后, 运放本身带来的噪声贡献可以由如下公式算出:公式4其中:I EQ = 等效的输入噪声电流, 这个值在带宽 F < 1/(2πR F C F )内有效.I N = 运放本身输入的电流噪声,算inverting 的输入.E N = 运放输入的电压噪声.C D = 前面的光电二极管的寄生电容.F = 带宽,单位为Hz.4kT = 1.6E – 21J at T = 290°K 根据这个公式计算出等效的输入噪声电流后, 就可以算出在TIA 输出后SNR 了.4 实际应用中的常见问题这里整理几个TIA 运放在实际使用中经常遇到的问题:4.1 振荡这个问题在高增益,又有宽带要求的情况下比较常见.比如设计一个20K 增益的放大链路, 假设总的输入的寄生电容很大, 10pF. 根据上面的图可以看出, 采用GBW 最宽的OPA847进行设计, 最宽稳定带宽只能在50M 附近.设计电路如下:ZHCA479跨阻型放大器应用指南7Figure7 未加补偿的20k 放大电路输入20n 的脉宽信号, 10u 的幅度, 得到的波形如下:Figure8 原始输出响应输出有振荡产生.根据公式算出CF 的取值应该为0.24p. 加上后,电路如下:ZHCA4798 跨阻型放大器应用指南Figure9 加补偿后的电路仿真得到: 可以看到, 振荡消失,只剩过冲. 放大倍数也趋向正常.Figure10 加补偿后的脉冲响应在高增益的场合, 有可能反馈电阻自带的电容以及反馈走线带来的寄生电容都可以达到这么微小的电容值. 所以需要依具体的测试结果来确定反馈是否要另外加电容.ZHCA479跨阻型放大器应用指南94.2 overshoot在光时域反射检测光纤状态的场合, 输出上的overshoot 可能会对测量结果产生很大影响,这就需要尽可能地减小TIA 输出的overshoot. 如上图所示的结果, 约有10%的overshoot, 这对实际使用是不利的,需要消除.消除这种过冲最有效的方法是加大反馈电容, 但是这样带来的一个直接后果是带宽减小. 如上面的案例, 在输出有overshoot 的情况下, 原始频响为: -3dB 带宽有40M 左右.Figure11 原始补偿的频响增大反馈到0.45p 时, 过冲消失.ZHCA47910 跨阻型放大器应用指南Figure12 增大补偿的电路Figure13 增大补偿后的脉冲响应但是也可以看到, 20nS的脉冲情况下, 其输出有点被滤除, 增益减小了. 原因就在于输出的带宽变窄, 只剩21MZHCA479跨阻型放大器应用指南 11Figure14 增大补偿后的频响5 总结TIA 运放在作电流放大使用时需要注意带宽和增益的折中, 以及平衡性和带宽的折中. 而同时又得兼顾噪声的贡献, 所以需要综合考虑以上的各项指标.6. 参考资料1. Xavier Ramus “Transimpedance Considerations for High-Speed Amplifiers”2. OPA847,OPA657指标书重要声明德州仪器(TI)及其下属子公司有权在不事先通知的情况下,随时对所提供的产品和服务进行更正、修改、增强、改进或其它更改，并有权随时中止提供任何产品和服务。

互阻放大器TIA特性

功能说明
TIA 是由普通 SC/CT 模块构建的。拓扑结构是一种运算放大器，带有可从输出至反相输入间选择的反馈电阻。或者，可选反馈电容也可以在输出到反相输入间保持连接。请参见以下 TIA 配置。
-3 db 频率
该组合框用于显示计算得出的带宽值。该值取决于电阻反馈、电容反馈值和功耗设置。
放置
没有放置特定的选项。
资源
TIA 使用一个 SC/CT 模块。通常，Vref 输入来自电压参考、VDAC 输出或 GPIO 上的外部提供的参考。
文档编号：001-67956 修订版**
第 3 页，共 15 页
PSoC® Creator™ 组件数据手册
互阻放大器 (TIA)
特性
• 可选转换增益 • 可选拐角频率 • 电容式输入源补偿 • 可调功耗设置 • 可选输入参考电压
概述
互阻放大器 (TIA) 组件通过电阻增益和用户选择的带宽向电压转换放大器提供基于运算放大器的电流。其来源为 SC/CT 模块。 TIA 用于将外部电流转换为电压。典型应用包括使用光二极管等电流输出进行的传感器测量。TIA 的转换增益单位为欧姆，其可用范围在 20 K 到 1.0 M 欧姆之间。光二极管等电流输出传感器的输出电容通常较大。这就需要在 TIA 中加入并联反馈电容，以保证稳定性。TIA 具有一个可编程的反馈电容，可以满足这一需要，并提供带宽限制，可降低宽频带噪声。
互阻放大器 (TIA)
PSoC® Creator™ 组件数据手册
模拟模块
1 个 SC/CT 固定 HW 模块
数据路径不适用
数字模块
API 存储器（字节）
宏单元状态寄存器控制寄存器计数器 7 闪存
不适用
不适用

tia跨阻放大电路

tia跨阻放大电路（最新版）目录1.TIA 跨阻放大电路的概述2.TIA 跨阻放大电路的工作原理3.TIA 跨阻放大电路的优点4.TIA 跨阻放大电路的应用领域5.TIA 跨阻放大电路的发展前景正文1.TIA 跨阻放大电路的概述TIA 跨阻放大电路，全称为“传输线输入阻抗匹配跨阻放大电路”，是一种常用于射频通信系统中的放大电路。

其主要作用是在保证输入信号质量的前提下，对信号进行放大，以便在长距离传输过程中降低信号衰减，提高信号传输的稳定性和可靠性。

2.TIA 跨阻放大电路的工作原理TIA 跨阻放大电路通过调整输入阻抗和输出阻抗，实现与传输线的阻抗匹配，从而减小信号在传输过程中的反射和损耗。

具体来说，TIA 跨阻放大电路由两个放大器组成，分别负责电压放大和电流放大。

输入端采用串联电阻的方式，使得输入阻抗与传输线的特性阻抗相匹配。

输出端则通过并联电阻的方式，与负载阻抗相匹配。

3.TIA 跨阻放大电路的优点TIA 跨阻放大电路具有以下优点：a.输入阻抗匹配：通过调整输入端的串联电阻，使得 TIA 跨阻放大电路的输入阻抗与传输线的特性阻抗相匹配，从而减小信号反射，降低信号损耗。

b.电流放大：TIA 跨阻放大电路中的电流放大器可以使得输出电流与输入电流之比保持在一定范围内，从而保证信号在长距离传输过程中的稳定性。

c.电压放大：TIA 跨阻放大电路中的电压放大器可以实现较大的信号放大倍数，以弥补信号在传输过程中的衰减。

4.TIA 跨阻放大电路的应用领域TIA 跨阻放大电路广泛应用于射频通信系统、无线通信设备、卫星通信系统、雷达系统等电子信息领域。

在这些领域中，TIA 跨阻放大电路可以有效提高信号传输的质量和稳定性，保证系统的正常运行。

5.TIA 跨阻放大电路的发展前景随着我国电子信息产业的快速发展，射频通信技术在军事、民用等领域的应用越来越广泛。

作为射频通信系统中的关键部件，TIA 跨阻放大电路在提高信号传输质量和稳定性方面具有重要作用。

tia跨阻放大电路

TIA跨阻放大电路简介TIA跨阻放大电路（Transimpedance Amplifier）是一种常用于光电探测器中的电路，用于将光电器件输出的电流转换为电压信号。

TIA电路的核心是一个跨阻放大器，它能够将输入的电流信号放大，并转换为输出的电压信号。

TIA跨阻放大电路在光通信、光电测量等领域具有广泛的应用。

TIA跨阻放大器原理TIA跨阻放大器的基本原理是利用负反馈放大器的特性，将输入电流转换为输出电压。

它由一个光电二极管作为输入电流源，一个电阻作为负载，以及一个运算放大器（Operational Amplifier）构成。

TIA电路的输入端连接到光电二极管，当光照射到光电二极管时，产生的光电流会通过输入电阻流入电路。

运算放大器的反向输入端连接到电阻的接地端，形成负反馈回路。

运算放大器的输出端与电阻的另一端相连，输出电压通过反馈电阻回到运算放大器的反向输入端。

当输入电流流入电路时，会在电阻上产生一个电压降，根据欧姆定律，电压与电流成正比。

运算放大器通过负反馈作用，将电阻上的电压放大，并输出到负载上。

TIA跨阻放大器设计要点在设计TIA跨阻放大器时，需要考虑以下几个关键要点：1. 输入电阻选择TIA电路的输入电阻需要足够大，以保证光电二极管输出的电流能够流入电路而不被短路。

一般选择高阻值的电阻，如几兆欧姆，以确保输入电流的有效转换。

2. 反馈电阻选择反馈电阻的选择决定了TIA电路的放大倍数。

一般情况下，反馈电阻越大，放大倍数越高。

但是过大的反馈电阻会引入噪声，影响电路的性能。

因此，需要在放大倍数和噪声之间做出权衡。

3. 运算放大器的选择运算放大器是TIA电路的核心部件，需要选择性能优良的运算放大器。

一般选择带宽较宽、噪声较低的运算放大器，以保证电路的放大性能和稳定性。

4. 功耗和速度考虑TIA电路在应用中还需要考虑功耗和速度的问题。

功耗较高的电路会产生较多的热量，影响电路的稳定性；速度较慢的电路会导致信号延迟，影响系统的响应速度。

一文读懂跨阻放大器的工作原理

一文读懂跨阻放大器的工作原理
跨阻放大器（TIA）是光学传感器（如光电二极管）的前端放大器，用于将传感器的输出电流转换为电压。

跨阻放大器的概念很简单，即运算放大器（op amp）两端的反馈电阻（RF）使用欧姆定律VOUT= I RF将电流（I）转换为电压（VOUT）。

在这一系列博文中，我将介绍如何补偿TIA，及如何优化其噪声性能。

对于TIA带宽、稳定性和噪声等关键参数的定量分析，请参见标题为用于高速放大器的跨阻抗注意事项的应用注释。

在实际电路中，寄生电容会与反馈电阻交互，在放大器的回路增益响应中形成不必要的极点和零点。

寄生输入和反馈电容的最常见来源包括光电二极管电容（CD）、运算放大器的共模（CCM）和差分输入电容（CDIFF），以及电路板的电容（CPCB）。

反馈电阻RF并不理想，并且具有可能高达0.2pF的寄生并联电容。

在高速TIA应用中，这些寄生电容相互交互，也与RF交互生成一个不理想的响应。

在本篇博文中，我将阐述如何来补偿TIA。

图1显示了具有寄生输入和反馈电容源的完整TIA电路。

三个关键因素决定TIA的带宽：
总输入电容（CTOT）。

由RF设置理想的跨阻增益。

运算放大器的增益带宽积（GBP）：增益带宽越高，产生的闭环跨阻带宽就越高。

这三个因素相互关联：对特定的运算放大器来说，定位增益将设置最大带宽;反之，定位带宽将设置最大增益。

无寄生的单极放大器
这一分析的第一步假定在AOL响应和表1所示的规格中有一个单极的运算放大器。

DC、AOL（DC）时运算放大器的开环增益
120dB
运算放大器GBP
1GHz。