低速永磁大转矩电动机的数值分析
永磁电机转矩常数深度分析
Abstract —The torque con stan t, together with the back-EMF co n sta n t, was origi n ally used i n perma n e n t mag n et DC commutator motors (PMDC motors) to couple the electric circuit equation s with mechan ical equation s. But it is still an open question whether the con cept of the torque con stan t an d the back-EMF con stan t can be applied to brushless DC (BLDC) motors an d perman en t magn et (PM) AC machin es. This paper presen ts an in -depth study of the two con stan ts un der various real conditions in PM machines. The torque constant at various load con dition s is computed usin g tran sien t 2D fin ite elemen t an alysis (FEA). It is shown that the torque con stan t is n ot a constant for BLDC motors and PM AC machines.Index Terms —Back-EMF constant, brushless DC motors, DC commutator machi n es, fi n ite eleme n t a n alysis, perma n e n t magnets, synchronous machines, torque constant.I. I NTRODUCTIONHE torque constant is defined as the ratio of the torquedelivered by a motor to the current supplied to it, and the back-EMF constant is the ratio of voltage generated in the winding to the speed of the rotor. In PMDC motors, they are almost constant at various load conditions. The torque constant and back-EMF constant couple the electric circuit equations with mechanical equations, and are widely used in motor control.It is of great interest to see whether the concept of the torque constant and the back-EMF constant can be applied to BLDC motors and PM AC motors. Some effort have been made in this regard [1][2]. Reference [2] indicates that an ideal BLDC motor (also called a square-wave motor), under the condition that the line-to-line back EMF waveform is trapezoidal and that the winding current waveform is ideally square, is electrically identical to a PMDC motor. The author also applies the concept of the torque constant and the back-EMF constant to sine-wave PM AC motors under the assumption that the internal power-factor angle between the back EMF and the current is fixed to zero.However, in real cases, the winding current waveform is far from the ideal square-wave in BLDC motors due to current freewheeling. And, in PM synchronous motors, the internal power-factor angle is normally not zero because the torque angle is automatically adjusted according to the changeThe authors are with Ansoft Corporation, Pittsburgh, PA 15219 USA(phone: 412-261-3200; e-mail: dlin@, ping@,zol@). in load. To this end, this paper presents an in-depth study of the torque constant and the back-EMF constant for BLDC motors and PM AC motors. The suitability of the use of the two constants in PM motors is discussed considering the following: current freewheeling, arbitrary back-EMF waveforms, salient pole, variable pulse width and trigger angle, and internal power factor angle.II. R EVIEW OF THE T ORQUE C ONSTANT IN PMDC M OTORS In PMDC motors, the electric circuit equation isb a s V I R E V ++=(1)where V s is the applied DC voltage source, E is the back EMF, V bis the voltage drop of one-pair brushes, I is the input DCcurrent, and R a is the armature resistance. Equation (1) can be coupled with load mechanical equations by introducing⎩⎨⎧==I k T k E T mmE ω (2)where ωm is the angular velocity in mechanical rad/s, T m is theelectromagnetic (air-gap) torque in Nm, k E is the back-EMF constant in Vs/rad, and k T is the torque constant in Nm/A. The torque constant and the back-EMF constant have the following properties:i. k T = k E in the metric unit system; ii. k T and k E are constant; iii. k T and k E are measurable.Property (i) is obvious from the fact that the electric power (EI ) is equal to the mechanical power (T m ωm ) during power conversion.Property (ii) follows since: (1) PMDC motors have large air gaps due to surface mounted magnets, thus the saturation change caused by the armature reaction is negligible; (2) the brush position is mechanically fixed during operation even if it is adjustable; (3) the current in each coil completes commutating within the angle of the brush width, and the commutating duration is independent of the rotor speed; and (4) there is no reluctance torque even if the armature reaction is not aligned with the q-axis.Based on property (ii), the back-EMF constant k E can be measured at no-load condition operating in generator mode. The torque constant k T can be obtained directly from k E , orcan be measured at load operation. It is straightforward to predict the performance of PMDC motors from (1) and (2) in motor control. In-Depth Study of the Torque Constant forPermanent Magnet MachinesD. Lin, P. Zhou and Z. J. CendesT©2008 IEEE.III. T ORQUE C ONSTANT IN BLDC M OTORSEven though the torque constant and the back-EMF constant in BLDC motors are defined in the same way as those in PMDC motors as shown in (2), there are some essential differences regarding the torque constant and the back-EMF constant between BLDC motors and PMDC motors. For the sake of easy discussion, take a Y-connected three phase winding with bridge-type inverter as an example, as shown in Fig. 1. The trigger pulse width for each branch is 120 electrical degrees in turn and the inverter has 6 repeatableoperating states with the state period of 60 electrical degrees.Fig. 1. Y-connected three-phase windings with the bridge-type inverterA . Voltage equation (1) is no longer applicable The voltage equation (1) is no longer applicable in BLDCmotors due to the inductance voltage drop. In PMDC motors, the inductance induced voltage caused by the current commutating will not contribute to the voltage drop across the brush terminals. However, in BLDC motors, the inductance voltage drop becomes comparable with the resistance voltage drop.B. k T and k E are no longer constantIn BLDC motors, E used in (2) is the average back EMF across the DC link, and its value will vary with the current freewheeling duration. In Fig. 1, assume at the previous operating state, the source voltage V s is applied to winding terminals AC via branches 1 and 2, and at the current operating state, V s is applied to winding terminals BC viabranches 3 and 2. When branch 1 is off, the phase-A currentfreewheels through branch 4, which makes winding A to connect in parallel with winding C. If the voltage drop acrossthe conducting transistor in branch 2 is the same as that acrossthe freewheeling diode in branch 4, the average back EMFduring the current operating state is])(21[10∫∫++=sf f T T BC T BC BA s dt e dt e e T E(3) where, e BC and e BA are instantaneous line-to-line inducedvoltages, T s is the state period in second (corresponding to 60electric degrees), and T f is the current freewheeling duration,as shown in Fig. 2. It is obvious from (3) that the average backEMF varies with the current freewheeling duration, andtherefore k Eis not constant for various operations.Fig. 2. Rectified back EMF from trapezoidal line-to-line induced voltagesFor the circuit of Fig. 1, as long as T f < T s , the freewheeling currents always reduce the input DC current and increase the delivered torque, and therefore, k T varies with the current freewheeling duration which in turn varies with the rotor speed.Another case in which k T is not constant is, in interior permanent magnet (IPM) motors, the reluctance torque component also contributes to the air-gap torque due to the salient-pole effects, and the reluctance torque component is not linearly proportional to the DC current. Furthermore, the trigger angle and the pulse width of the controlling signals in BLDC motors are usually controllable. This is also a casewhere k T is not constant.Fig. 3 shows the variation of k T with the speed of a typical surface mounted BLDC motor with fixed trigger angle andpulse width.Fig. 3. Variation of k T with the rotor speed C . k T is no longer equal to k EIn BLDC motors, the back EMF across DC link normally includes ripples associated with arbitrary line-to-line back-EMF waveforms. The ripples become considerable due to thecurrent freewheeling even though the line-to-line induced voltage may have a flat waveform in 60 electric degrees by aspecial design (see the solid lines inside T s in Fig. 2). Theinput current also contains significant ripples because thefreewheeling current is in nature of “generator” current. Byexamining the power conversion, one gets∫⋅⋅=s T s m m dt i e T T 01ω(4)∫⋅∆⋅∆+=sT sdt i e T EI 01where, ∆e and ∆i are the ripples of the DC back EMF and theinput current, respectively. From (4), one concludes that atload conditions k T ≠ k E because T m ωm ≠ EI .D . kE is no longer measurable By measuring the air-gap torque (which is obtained from the load torque and the mechanical loss) and the DC component of the input current at load operation, k T can be determined. However, k E is no longer measurable at load conditions for BLDC motors. It cannot be measured by driving the motor as a generator and rectifying the line voltage with a rectifier as described in [2] because k E at load conditions is different from that at the no-load condition. Also it cannot directly be obtained from k T because k E ≠ k T at load conditions.IV. T ORQUE C ONSTANT IN PM AC M OTORSThe torque constant in PM AC motors can be defined as the ratio of the torque to the peak value of the input AC phasecurrents I peak , and the back-EMF constant is the ratio of thepeak value of the induced phase voltages E peak to the speed of the rotor, as expressed below [2] ⎩⎨⎧==peakT m mE peak I k T k E ω. (5) Most PM AC motors operate as synchronous motors. In PM synchronous motors, the internal power factor angle ϕ i , the angle between the back EMF phasor and the current phasor, is automatically adjusted based on the mechanical load and is normally not zero. In these cases, the delivered mechanical power is E peak peak T m m k E I k T /⋅=ωi rms rms i E T E mI mk k ϕϕcos cos 2⋅= (6) where I rms and E rms denote RMS values of sine-wave phasecurrent and back EMF, and m is the number of phases. For thepower conversion, the mechanical power must be equal to theelectric power, that ism m T ωi rms rms E mI ϕcos =. (7) As a resulti E T k mk ϕcos 2=. (8)One concludes from (8) that k T is not constant for PM synchronous motors even though K E may be constant when the saturation effects can be ignored. It varies with the internal power angle which in turn varies with the mechanical load.Equation (8) is derived under the assumption that the spatial harmonics of the air-gap magnetic fields produced bythe permanent magnets and the phase currents are ignored. Inorder to show the effects of the spatial field harmonics on thetorque constant, a three-phase 4-pole PM synchronousmachine, as show as in Fig. 4, is analyzed using 2D transientfinite element method (FEM). To focus on observing thevariation of the torque constant with the internal power factorangle, the change in saturation caused by armature currents isignored, and thus linear materials are used for all components.Fig. 4. The one-pole geometry layout of the three-phase 4-pole PM synchronousmachine Three-phase windings are applied with DC currents as follows⎪⎩⎪⎨⎧−=−==IAm I IAm I IAmI CB A *5.0*5.0 (9) where IAm is set to be 0 and 1A via parametric analysis. Therotor speed is set to be 1500rpm, and the rotor initial position is set to such a position that the phase-A winding has positive maximum induced voltage at time = 0. The computed torques at IAm = 0 and 1A are shown in Fig. 5. It can be seen from Fig. 5 that the torque at IAm = 1A consists of two components: one is the component producedby the phase currents, and the other is the cogging torque component which is produced by the permanent magnets at 0phase currents. Because linear materials are used, the torque component produced by the phase currents can be directlyderived from the result of the torque at IAm = 1A minus thetorque at IAm = 0, as shown in Fig. 6. By definition, the curvein Fig. 6 shows the torque constant because the torque isproduced by unit phase currents. One notes that the torqueconstant is not a constant as had been anticipated and istherefore not suitable for use with PM AC machines.Fig. 5. Torques at different phase currents varying with the internal power factor angle ϕ i (time=20ms corresponds to ϕ i =360 electric degrees)Fig. 6. Torque produced by unit phase current varying with the internal power factor angle ϕ i (time=20ms corresponds to ϕ i =360 electric degrees)V. C ONCLUSIONThe torque constant and the back-EMF constant which were originally used in PMDC motors are generally not suitable for BLDC motors and PM synchronous motor analysis. Detailed computations of both constants with real motors reveal that they are no longer constant but, instead, vary significantly with load conditions.R EFERENCES[1]Electro-Craft Handbook, Fifth Edition, August 1980, ISBN 0-960-1914-0-2.[2]J.R. Hendershot Jr, and T. J. E. Miller, Design of Brushless PermanentMagnet Motors, Magna Physics Publishing and Clarendon Press, Oxford, 1994.Din gshen g Lin received his B.S. and M.S. degrees in Electrical Engineering from Shanghai University, Shanghai, China, in 1982 and 1987, respectively. He is currently a Senior Research and Development Engineer at Ansoft Corporation, Pittsburgh, PA. Before he joined Ansoft in 1999, he was an Associate Professor of electrical engineering at Shanghai University. His research interests include design and optimization techniques of electrical machines and electromagnetic field computation. He received the third prize of the Chinese National Award of Science and Technology, in 1987, and two second prizes of the Shanghai City Award of Science and Technology, in 1986 and 1989.Ping Zhou received his M.S. degree from Shanghai University, China in 1987 and his Ph.D. degree from Memorial University of Newfoundland, Canada in 1994. He was with Shanghai University as a lecturer after his undergraduate study in the same university in 1977. He was a Visiting Scholar of Memory University of Newfoundland from 1989 to 1991. Since 1994, he jointed Ansoft Corporation in the R&D department. Currently, he is the manager of Electromechanical R&D group at Ansoft. His research interests include finite element numerical field computation, circuit coupling, multi-physics coupling and electrical machine modeling.Zoltan Cendes is Founder and Chairman of Ansoft Corporation, Pittsburgh, PA, and is an Adjunct Professor at Carnegie Mellon University, Pittsburgh, PA. In addition to his role at Ansoft, Dr. Cendes has served as a Professor of Electrical and Computer Engineering at Carnegie Mellon University, as an Associate Professor of Electrical Engineering at McGill University, Montreal, Canada, and as an Engineer with the Corporate Research and Development Center of the General Electric Company in Schenectady, NY. Dr. Cendes received his M.S. and Ph.D. degrees in Electrical Engineering from McGill University and his B.S.E. degree from the University of Michigan.。
永磁电机转矩常数的深度
In Depth Study of the Torque Constant for Permanent-Magnet Machines
指導老師：黃昌圳 學生：陳育俊
摘要
介紹 PMDC馬達的轉矩常數回顧 BLDC馬達的轉矩常數 PM交流馬達的轉矩常數 轉矩常數的其他定義方式 結論
輸入電流因為也包含顯著的漣波。藉由功率轉 換的定理，可以得到
1 Tmm Ts
0
Ts
1 e i dt EI Ts
0
Ts
e i dt
(4)
Δe和Δi是直流反電動勢和輸入電流的漣波，相 對地，從由上式，得知因為Tmωm≠EI所以在負 載情況kT≠kE。
D. kE在負載下不再被精確測量
交流永磁馬達的轉矩常數被定義為轉矩與交流 輸入相電流Ipeak的峰值的比例，並且反電動勢 常數為感應相電壓Epeak與動子轉速的比例
E peak k E m T k I m T peak
(5)
大部分交流永磁馬達操作如同同步馬達。在永 磁同步馬達，內部的功率因素角ψi是反電動勢 相量和電流相量之夾角，從機械負載的觀點， 夾角會隨著負載自動被調整，因此一般不是零。 在這些情況，傳送的機械功率為
(13)
R1是相繞組電阻，Ld和Lq是在dq軸上的繞組同步 電感，p代表d/dt，並且 . eq -md -(nppmd) m (14)
λmd是由永磁轉換成d軸的繞組磁通交鏈，ωm是 rad/s的機械角速度，npp是馬達的極對數。 以N· m的轉矩為 Tm n pp (Lq - Ld )idiq - mdiq (15) 大部分的表面型永磁馬達 Lq=Ld，因此轉矩為
永磁同步电机直接转矩控制系统低速性能研究
科技创新导报 ２００８ ＮＯ．１７ Ｓｃｉｅｎｃｅ ａｎｄ Ｔｅｃｈｎｏｌｏｇｙ Ｉｎｎｏｖａｔｉｏｎ Ｈｅｒａｌｄ
n
rN
圆周方向的分布剩磁密度，θr 为定子齿中心线
与转子磁极中心线之间的夹角。
由（１）式可知，齿槽转矩是周期性脉动的，
其基波周期为 ｚ，并且齿槽转矩是电机固有特
性，与转矩和转速大小无关。
１．２ 纹波转矩的影响
在磁路不饱和、忽略电枢反应的影响
及定子绕组 Ｙ 型连接无中线且三相对称的
前提下，得到总的电磁转矩为［１］：
图 ２ 为直接转矩控制系统主程序流程图。
３ 结语
本文详细分析了直接转矩控制下永磁同 步电动机在低速运行时转矩脉动产生的原因， 并给出了硬件原理图与主程序流程图。随着 直接转矩控制技术的实用化、全数字化、智 能化与集成化的发展，永磁同步电动机直接转 矩控制系统在低速运行时的性能将不断得到 改善，其应用必更加广泛。
s
当系统运行在低速时，定子电阻 Rs 的影响加
大，磁链的观测值将会出现很大误差，严重影
响着整个系统的控制性能。
２ 直接转矩控制的硬件实现
在交流调速领域，尤其在对电机控制时， 由于系统运算量大，电机的电气时间常数小等 特点，使得一般的微处理器很难胜任。通常在 交流调速领域里选 ＤＳＰ 作为其控制器处理器。
（３）
上式表明在大惯性、高次谐波频率及 电机在高速运行时，转速的脉动将会很小， 即大的转动惯性起到了滤波器的效应，纹 波转矩可被转子惯性所吸收，但当转速很 低时，纹波转矩影响加大，转速脉动也会相 应变大。 １．３ 死区效应的影响
低速大转矩永磁同步电机及其控制系统共3篇
低速大转矩永磁同步电机及其控制系统共3篇低速大转矩永磁同步电机及其控制系统1低速大转矩永磁同步电机及其控制系统永磁同步电机是一种磁铁固定的电机，在工业生产中应用广泛。
低速大转矩永磁同步电机是其中一种，在许多应用场合广受欢迎。
本文将介绍低速大转矩永磁同步电机及其控制系统的工作原理、特点以及在不同领域的应用。
一、低速大转矩永磁同步电机的工作原理低速大转矩永磁同步电机是一种基于磁场共振原理来实现转矩输出的电机，其结构包括永磁体、定子和转子。
永磁体固定在定子上，输送直流电流产生轴向磁场，而定子上的绕组产生旋转磁场。
转子上的磁场与旋转磁场相互合作，使得转子受到的转矩最大化。
由于磁场共振效应，使得低速大转矩永磁同步电机在稳态运行时，能够产生更大的转矩输出，同时保持较高的效率。
二、低速大转矩永磁同步电机的特点1.具有高效率和高功率因数。
低速大转矩永磁同步电机的效率可以达到80%以上，功率因数可以接近1。
2.具有高精度和高性能。
低速大转矩永磁同步电机的转矩输出和转速能够实时控制，可以满足不同领域下的高性能和高精度要求。
3.工作稳定、可靠性高。
低速大转矩永磁同步电机适用于长期持续运转，并且不需要额外的机械结构来保证稳定性。
三、低速大转矩永磁同步电机的控制系统低速大转矩永磁同步电机的控制系统需要实现对转速、转矩和位置等参数的控制。
传统的控制方法包括PID控制、模型预测控制等，但是由于低速大转矩永磁同步电机的特殊性质，需要采用更加先进的控制方法。
现在广泛使用的控制方法有：磁场定向控制和磁场调制控制。
磁场定向控制是通过控制不同轴的磁场来实现对电机的转速和位置的控制。
磁场调制控制则是通过在电机不同部分施加不同频率的磁场以达到控制转速和转矩的效果。
四、低速大转矩永磁同步电机的应用由于其高效率、精度和稳定性，低速大转矩永磁同步电机在很多领域都得到了广泛应用。
在机床上，低速大转矩永磁同步电机可以带动机床的主轴，实现高精度和高速度的金属加工。
永磁电机转矩常数的深度课件
電樞反應影響的飽和度變化可以被忽略 。
b)在運作時電刷位置被機械固定即使他可以被調整 。 c)在電刷寬度的角度內每個線圈的電流完成換相，並
且換流持續時間不受轉子速度的影響。
d)即使電樞阻抗並沒有排列在q軸，也沒有磁阻轉矩。
由特性3)的觀點，反電動勢kE可以在發電機模 式無載情況下被測量，轉矩常數kT可以直接從 kE獲得，或者從負載操作下獲得。
A. 固定電樞磁場旋轉轉子
三相繞組是用於直流由下表示
IIba
Im -0.5
Im
(9)
Ic -0.5 Im
Im經過參數分析設定為0和1A。轉子轉速設定 為1500rpm，轉子的起初位置設定在a相繞組在
t=0時有正的最大感應電壓的位置。
在Im為0和1A時，轉矩顯示下圖。
我們可以看到Im=1A的轉矩包括兩個成分： 其中一個成分由相電流產生，另一個頓轉轉矩
永磁电机转矩常数的深度
摘要
介紹 PMDC馬達的轉矩常數回顧 BLDC馬達的轉矩常數 PM交流馬達的轉矩常數 轉矩常數的其他定義方式 結論
介紹
Tm=kTI。
E=kEωm。 kT和kE 將電路方程式與機械方程式結合一起，
並且廣泛使用在馬達運動控制。 兩個常數使用在PM馬達必須討論以下：
馬達控制可以很簡單的從上面兩式預測永磁直
流馬達的特性。
BLDC馬達的轉矩常數
低速永磁风力发电机起动阻力矩的分析计算
样机
额定容量 / kVA
额定转速 / r. m in - 1
极数
起动 计算值 /N. m
阻力 实测值 /N. m
矩
偏差 /
1# 3 750 8 5. 5 4. 8 14. 6
2# 3 200 12 7. 8 7. 2 8. 33
3# 5 150 16 9. 5 8. 8 7. 95
4# 7. 5 150 16 10. 4 9. 8 6. 12
(3)
Vδ = Aδδ
(4)
对于气隙磁场
Bδ
=
μ 0
Hδ
(5)
式中 μ0 ———真空磁导率
由式 ( 1) 、式 ( 2)得
Bm Hm Am hm =σkrBδHδAδδ 考虑式 ( 3) 、式 ( 4) 、式 ( 5) ,并整理得
(Bm Hm ) Vm = σμkrBδ2 Vδ 0
自起动低速大转矩永磁同步电动机的设计分析
自起动低速大转矩永磁同步电动机的设计分析摘要:现在有很多场合需要用到低速大转矩的驱动电机,直驱式电机相比传统驱动电机具有显著的优点。
依照低速大转矩自起动永磁同步电动机的技术要求初步设计了一种电机,介绍了该电机的基本结构,计算了电机的主要参数,并利用ANSYS、MATLAB软件,采用时步有限元法进行了仿真计算,并对仿真结果进行了一定的分析。
关键词:低速自起动永磁同步电动机设计分析随着科学技术的发展,越来越多的场合需要用到低速大转矩的驱动装置,普通电机转速较高,在日常应用中需辅助一定的减速机构,这既降低了效率,又造成设备上的浪费。
文献[5]提出根据pn=60f,在频率确定情况下,增加电机的极对数可大幅度地降低转速,同时输出较大转矩,这种电机可用于低速直接传动,能够省齿轮箱等笨重的减速机构,因此具有很好的应用前景。
本文提出的多极永磁同步电动机,在极对数数倍于普通电机的情况下,铁芯槽数并不提高太多,与极数接近,提高了电机的单位体积出力。
从文献[1]可知本电机的结构和设计方法均与传统电机有很多不同之处,与传统的永磁同步电动机相比,其显著的特点有:多极的磁路安排,绕组分配特殊;电机重量减轻,电机体积小,具有高功率密度(单位体积所产生的转矩大);具有自起动能力。
文章给出了设计方案,介绍了该电机的结构,然后给出了电机时步有限元仿真结果,并对仿真结果进行一定的分析研究,最后提出了设计的不足之处和需要改进的地方。
1 电机的基本设计方案1.1 模型机规格此电机的极数为30,定子槽数为36,由于极槽数接近,与传统交流电机的一个极下有3相绕组的结构形式有较大差别,每极每相槽数为分数,即2/5。
电机永磁体嵌放于转子侧,采用内置切向式结构。
电机的主要尺寸是依照Y400-6系列电机的规格作为参考确定的。
永磁同步电动机为减小过大的杂散损耗,降低电动机的振动与噪声和便于电动机的装配,其气隙长度?一般要比同规格的感应电动机的气隙大。
永磁无刷力矩电动机峰值转矩能力的研究
本文1996年4月22日收到 设计分析 永磁无刷力矩电动机峰值转矩能力的研究孙立志 王 强 陆永平(哈尔滨工业大学 哈尔滨150001)Study on Peak Torque Capability of Permanent Magnet Brushless Torque MotorSun Lizhi Wang Qiang Lu Yongping(Harbin Institute o f Technolog y,Ha rbin 150001) 【摘 要】 在考虑了永磁无刷力矩电动机峰值极限转矩主要制约因素的基础上,分别讨论了正弦波及方波驱动方式下的力矩电动机峰值极限电流及转矩,并针对正弦波驱动方式下的该类电机分析了影响峰值转矩能力的主要因素,文中还进行了数值计算并加以实验验证。
【关键词】 永磁无刷力矩电动机 峰值转矩能力【Abstract 】 This paper considers th e majo r con-straints to thc peak to rque capa bility o f permanent mag ne t brushless to rque mo to r (PM BL T M ),discusses pea k cur-re nt limit and pea k to rque limit o f sine wav e PM BL TMand squa rewav e o ne respec tiv ely ,and ana ly zes the influ-ence o f som e factor s on the peak to rque capa bility of PM -BL T M.N umc rical calculation a nd ex periments a re made accor ding to a sa mple mo tor.【Keywords 】 pe rmanent mag net br ushless to rque mo tor pea k tor que ca pability1前 言在一些控制系统中,常常需要所使用的力矩电动机产生瞬时峰值转矩,从而以较大的加速度来驱动负载,由于使用NdFeB 表面磁钢的无刷力矩电动机具有高转矩惯量比、高过载能力[1],所以非常适合此种运行状态。
低速大转矩多极永磁电机齿槽转矩削弱的有效方式
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699 轮毂式永磁同步电动机电磁转矩的数值计算与误差分析
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由虚位移法计算所得的电磁转矩特性 ./01234!5,602+1 2437809 15/18/520: ); <+3285/ 0603,; !02=4: 图> *+,- > 气隙单元的剖分示意图 G09= ,060352+46 +6 5+3 ,5H 30,+46
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! 网格剖分对电磁转矩计算误差的影响
波永磁同步电机的电磁转矩，均可获得较高的计算 精度。但麦克斯韦应力法具有计算量小和计算精度 高的特点，更适合于电磁转矩的数值计算。 （#）单元剖分的疏密度和形状将直接影响电磁 转矩的计算精度。当运用麦克斯韦应力法计算转矩 时，气隙单元的剖分尤为重要。此此，积分路径所 经的三角元须对称分布（见图 =） ，方可保证转矩 数值解的计算精度。 （>）应用斜槽结构或减小槽口宽度，可以有效 地抑制表面磁极式永磁电机的定位转矩。 参考文献
低速大转矩永磁同步电动机的转子结构及永磁体设计策略
低速大转矩永磁同步电动机的转子结构及永磁体设计策略摘要：本文在探讨永磁同步电机与低速大转矩永磁同步电机概念后，分析转子机构的设计策略以及永磁体的优化设计。
仅以本文设计成果，为我国电机企业借鉴参考，形成永磁同步电机开发的全新思路。
关键词：永磁同步电机；永磁体；转子结构；转子支架中图分类号：TM341 文献标识码：ARotor Structure and Permanent Magnet Design Strategy of Low Speed High Torque Permanent Magnet Synchronous MotorHao Shuangge, Hongyan, Yan Shuqing, Wang ShengGuizhou Aerospace Linquan Motor Co., Ltd. Guizhou Guiyang 550000Abstract: After discussing the concepts of permanent magnet synchronous motor and low-speed high torque permanent magnet synchronous motor, this paper analyzes the design strategy of rotor mechanism and the optimization design of permanent magnet. Based solely on the design results of this article, it is intended to serve as a reference for Chinese motor enterprises and form a new approach for the development of permanent magnet synchronous motors.Keywords: Permanent magnet synchronous motor; Permanent magnet; Rotor structure; Rotor bracket在国家环保政策不断深入以及永磁材料价格逐渐区域稳定的环境之下，我国永磁同步电机的应用范围越发广泛，且应用经验不断丰富、积累，大量企业均以永磁同步电机取代了以往的异步电机，从而基于低速大转矩永磁同步电机的优势提升企业生产效率。
低速永磁风力发电机的参数分析及优化设计
第34卷第9期 2011年9月合肥工业大学学报(自然科学版)JO U RN AL O F H EFEI U N IV ERSIT Y OF T ECH N OL O GYVol.34No.9 Sept.2011收稿日期:2010-12-22基金项目:安徽省科技攻关计划资助项目(06012179)作者简介:何庆领(1969-),男,安徽肥东人,博士生,合肥工业大学副研究员;王群京(1960-),男,安徽蚌埠人,博士,合肥工业大学教授,博士生导师.Doi:10.3969/j.issn.1003-5060.2011.09.010低速永磁风力发电机的参数分析及优化设计何庆领, 王群京(合肥工业大学电气与自动化工程学院,安徽合肥 230009)摘 要:文章讨论了低速永磁同步风力发电机的设计特点,为了有效地减少阻力矩,采用分数槽绕组,为减少漏磁通,采用瓦片型和放射状的永磁体安装结构,并重点对结构参数与运行性能之间的内在关系进行了参数分析,为风力发电机本体的优化设计打下基础。
在一定安装尺寸的限制下,以电机效率作为优化目标,采用基于混沌理论的最优化算法获取风力发电机的最大输出效率。
关键词:风力发电机;参数分析;优化设计中图分类号:T M 315 文献标识码:A 文章编号:1003-5060(2011)09-1317-04Parameter analysis and optimal design for low -speedpermanent magnet wind turbine generatorsH E Qing -ling , WANG Qun -jing(School of E lectric E ngineering an d Automation,H efei U nivers ity of T echnology,Hefei 230009,Chin a)Abstract:This paper discusses the design character istics of low -speed per manent m ag net sy nchr ono us w ind tur bine generator,including the use of fr actio nal slot w indings to effectively reduce the r esist -ance m oment,the use of tiles and reflective -like structure to reduce leakage flux,and the installationof perm anent mag net.T he intrinsic r elationship betw een structur al parameters and o peratio nal per -form ance is also analyzed for the optimal design o f w ind turbine foundation.Aim ing at optimizing the motor efficiency,the optim ization algor ithm based on chaos theo ry can be used to obtain the max im um output efficiency o f wind turbine generator under a certain restriction of installatio n size.Key words:wind turbine generator;parameter analy sis;o ptimal design0 引 言风力资源是一种清洁、安全和可再生的绿色能源。
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低速永磁大转矩电动机的数值分析
根据电磁场理论和传热学知识，分析了电机在堵转运行的特性。
以一台5kW、10极切向式永磁大转矩电动机为例，利用场路耦合有限元法，对电机进行二维场、路及运动的耦合分析，得出堵转运行时的电磁场分布。
标签：大转矩永磁电机；堵转；场路耦合
随着大转矩电机的广泛应用，低速大转矩永磁电动机也逐渐成为该行业的发展的必然趋势。
当发生堵转时，电机内部温度高的问题也是需要分析解决的必然问题。
因此分析电机在堵转运行时温度分布情况是十分重要的。
要了解这种状态下的温度分布首先要分析此时的电磁场分布情况。
本文以一台3相10极低速大转矩电动机机为例，分析该电机堵转时的电磁场及温度场。
为了更全面的考虑电机非线性及结构的复杂性，本文首先采用场路耦合的方法计算电机的二维电磁场，只有在准确电机电磁场的基础上，才能准确计算电机的各种损耗，进而求解此时电机三维温度分布。
1 堵转时的电磁场分布
1.1 场路耦合分析
本文采用场路耦合法。
电机的直线部分和端部分开研究。
在永磁电机等效电路中阻抗不改变的前提下，来模拟电机端部的影响。
同时，为了计及谐波的影响，采用电感模拟定子谐波漏抗。
其中定子线圈端部相电阻计算公式为[2]：
其中，？籽W-导线电阻率，N-每相串联匝数，lc-线圈半匝平均长度，Nt-导线并绕根数，A’c-每根导线截面积，a-定子相绕组并联支路数。
定子线圈端部漏抗及谐波漏抗等效电感的计算公式为：
L=（Xd1+Xe1）/？棕
式中，Xd和Xe分别是定子端部漏抗及谐波漏抗，？棕是电机角速度。
为了简化分析，在电磁场求解过程中，假设：
（1）电磁场是似稳场。
（2）材料为各向同性，忽略铁磁材料的磁滯效应。
（3）忽略电导率？滓和磁导率？滋的温度效应，仅为空间函数。
（4）电磁场为二维分布。
（5）永磁材料用等效面电流模拟。
（6）场区中各场量随时间按正弦变化
其表达式为
式中-求解区域1-电机定子外圆和转子内圆边界；？祝2-永磁体边界；Js-永磁体边界等效面电流密度；A-磁矢量；？滋-磁导率；Jz-外加轴向电流密度；-？滓dA/dt-涡流密度。
通过采用有限元法求解。
图1单个定子内外绕组电流自由度和电动势自由度耦合图，图2为电机的场路耦合模型。
1.2 仿真结果与分析
通过公式法计算的结果可以看出，端部电阻为0.40425Ω，端部电感为0.049135H。
本文仿真低速大转矩电机的主要参数为：额定功率5kW，额定电压220V，定子槽数48，极数10，定子铁心长度225 mm。
利用Ansys软件对所建场路耦合模型进行模拟分析，得出电机在堵转和负载时的磁力线分布图。
图3和图4分别为负载和堵转运行时的磁力线分布图。
由图3、图4可以看出，电机负载运行时磁力线分布规律，磁力线垂直通过气隙，当堵转运行时磁力线扭曲相当严重。
这种现象产生的主要原因是电机绕组电流过大，导致铁心饱和严重，磁力线发生扭曲变形。
由于堵转时电机定子电流比额定电流大很多。
只有在准确电磁场的情况下，才能进一步准确分析出电机的损耗分布。
2 结束语
本文运用传热学知识，通过电机在堵转状态时的特性，运用ANSYS软件对低速大转矩电动机的定、转子整体进行电磁场分析，全面细致的了解电机故障堵转时电磁场分布状况。
从电机定、转子电磁场可以看出，由于堵转时电机的定子电流不断上升，电机磁通密度饱和不断加深。
这说明所建立的模型和使用方法是正确可行的。
参考文献
[1]王延觉，杨凯.低速大转矩直接驱动电动机研究与应用进展.微特电机，2007（5）：46-49.
[2]夏正泽，刘慧娟.基于场路耦合法德异步牵引电机电磁场分析.微电机，2009，3（42）：21-23.
[3]Hannalla A Y，Macdonald D C. Numerical analysis of transient field problems in electrical machines. Proceedings of the Institution of Electrical Engineers，1976，
123（9）：893-898.
[4]付敏，孔祥春.水轮发电机定子三维温度场的有限元计算.电机与控制学报，2000（4），193-197.
[5].孙立柱.低速大转矩永磁电动机电磁场与温度场数值分析.沈阳工业大学硕士论文，2011.
作者简介：隋勇（1976-），男，辽宁丹东人，工程师，学士，主要从事技术招标相关工作。